Specification
The invention relates to Code Division Multiple Access communication systems and, more particularly, to methods and apparatus for providing downlink diversity in such systems employing Walsh codes.
Background of the Invention
Consider a Code Division Multiple Access (CDMA) system that uses orthogonal Walsh functions to separate users on the downlink along with a base station specific covering code. It is well known that "downlink" refers to the communication path or paths from a base station to a mobile terminal or station (hereinafter referred to as a "mobile"), as compared to the "uplink" which is the communication path or paths from a mobile to a base station. For a system with K mobiles receiving signals from a common base station, the transmitted signal on a single antenna may be represented as:
where P, is the power transmitted to the /ith mobile, s,(t) and w,(t) are the data signal and unique Walsh function intended for the ith mobile, respectively, Pp is the power of the pilot signal which uses Walsh function 0, and p(t) is the covering code for the base station of interest. Further, the Walsh functions are orthogonal and repeat every symbol time TSJ i.e.:
where h(t) is the complex multiplicative distortion caused by the wireless channel and n(t) is thermal noise. Mobile / correlates the received signal with the ith Walsh function during the kl symbol interval after uncovering to achieve the decision statistic zf[k]:
where h[k] represents the cumulative effect of the channel h(t) over the kth symbol interval, and s,[k] is the kth transmitted symbol for the ith mobile.
The transmitted symbol can be recovered by using an estimate of the channel, h[k], obtainable from the pilot channel, i.e.:
wheref(•) is an appropriate decision function. Assuming a flat, Rayleigh faded channel, in the absence of fast, accurate power control, the resulting performance of the link will be rather poor due to the lack of diversity. As a result, it is desirable to have a second antenna at the receiver to allow diversity reception, improving performance considerably. However, mobile handsets do not easily allow a second antenna to be added. Thus, methods of achieving diversity performance from the transmitter have been proposed.
One method of achieving diversity performance is to transmit the same signals on multiple carriers. However, this is wasteful of the one resource that cannot afford to be wasted in mobile communications, namely, bandwidth. A second more reasonable alternative is delay diversity. This method purposely causes multipath by transmitting the signal twice from the base station with the second transmission delayed in time by several chips and occurring on a separate antenna. By dividing power over two transmissions transmit power is not increased, but time diversity is provided which can easily be exploited by the mobile's Rake receiver with no required changes. However, the scheme is ultimately limited by the self-interference caused by this intentional multipath. Since
all user signals are transmitted synchronously, this multipath interference can be quite large, especially for a moderate to heavily loaded system.
Other techniques have been proposed for attempting to improve diversity in the area of space-time coding. For example, a simple two-branch transmission diversity technique is described in Siavash M. Alamouti, "A Simple Transmit Diversity Technique for Wireless Communications," IEEE Journal On Select Areas In Communications, Vol. 16, No. 8, October 1998, the disclosure of which is incorporated herein by reference. Further, the use of channel codes for improving the data rate and the reliability of communications over fading channels using multiple transmit antennas is described in Vahid Tarokh et al., "Space-Time Codes for High Data Rate Wireless Communication: Performance Criterion and Code Construction," IEEE Transactions On Information Theory, Vol. 44, No. 2, March 1998, the disclosure of which is incorporated herein by reference. Still further, a technique for coding messages for transmission on the downlink so as to use multiple transmitting antennas for improved reception in fading environments is described in U.S. provisional application Serial No. 60/114,621, filed on January 4, 1999, entitled: "Space-Time Spreading Method of CDMA Wireless Communication," which is the basis for a U.S. non-provisional application filed on April 2, 1999 having the same title.
Summary of the Invention
The present invention provides methods and apparatus for improving downlink diversity in a CDMA system. This is advantageously achieved by employing Walsh codes (also known as Walsh functions) in conjunction with M transmit antennas at a common base station serving K mobiles, where M may or may not be equal to K. It is to be appreciated that such form of downlink diversity is referred to herein as Walsh diversity or space-time code diversity.
In any case, each transmit antenna transmits a signal representing the result of the modulation of Walsh codes by data signals for each of the K mobiles, assuming there is data pertaining to a particular mobile. For example, a first transmit antenna may transmit
a signal that may include: a component that is the result of a first Walsh code modulated by a data signal associated with a first mobile; a component that is the result of a second Walsh code modulated by a data signal associated with a second mobile; up to and including a component that is the result of a kh Walsh code modulated by a data signal associated with a kth mobile. Then, a second transmit antenna may transmit a signal that may include; a component that is the result of the Kth Walsh code modulated by a data signal associated with a first mobile; a component that is the result of the (K-l)th Walsh code modulated by a data signal associated with a second mobile; up to and including a component that is the result of the first Walsh code modulated by a data signal associated with a f£h mobile. As is evident, in this example, the same number of Walsh codes are used in conjunction with the second antenna as are used at the first antenna, but simply redistributed with respect to the data signals that respectively modulate them. The ordinals assignment of users may be arbitrary, that is, any user can be specified as user K. However, the relative assignment of Walsh codes is accomplished in accordance with a transmission matrix T. The transmission matrix maps the user data symbols onto the Walsh codes for each antenna and is preferably designed such that its columns are representative of the transmit antennas and are orthogonal. The rows of the matrix are representative of orthogonal channels such as the Walsh codes and frequency bands. Similar redistribution occurs for each of the M transmit antennas such that the respective component in each of the M transmit signals associated with a given mobile's data signal modulates a unique Walsh code. Thus, at a given mobile, the mobile need only correlate with M different Walsh codes since each transmit signal received will contain a component that is the result of M different Walsh codes having been modulated by a data signal associated with that particular mobile. However, it should be noted that in specific embodiments, less than M different Walsh codes may be used. It is also to be appreciated that the M antennas at the base station are preferably independent or uncorrelated with respect to one another.
As will be explained below, the set of Walsh codes to select from includes 2 log2:(k) walsh codes. However, with respect to any given mobile, there is needed only as many different Walsh codes as there are transmit antennas serving the particular
mobile. The results of the respective M correlations are combined and, with obtained channel estimates, the receiver estimates the particular data pertaining to that particular mobile so that it can be provided to the user. It is to be appreciated that K can be less than the total number of mobiles with which the base station is capable of establishing communications. Also, the present invention provides for various ways to estimate the respective channels between the M transmit antennas and a mobile. Accordingly, the present invention provides improved downlink diversity using Walsh codes in such a manner that: (i) does not require additional receive antennas at a mobile, (ii) does not waste bandwidth; and (iii) does not cause self-interference.
It is to be appreciated that the downlink diversity techniques of the invention may be applied to the physical layer proposed in the IS-95C physical layer draft proposal. This form of diversity advantageously extends the orthogonal transmit diversity (OTD) option of the draft proposal, which obtains only minimal coding gain, and provides true diversity gain. The present invention provides various IS-95C direct-spread and multi-carrier embodiments. However, it is to be understood that the techniques of the invention are more widely applicable to any CDMA communication systems or the like.
Brief Description of the Drawings
•>. FIG. 1 is a block diagram illustrating a CDMA system for implementing downlink
diversity according to the invention;
FIG. 2 is a flow chart illustrating a method for providing downlink diversity according to the invention;
FIG. 3 is a block diagram illustrating a Walsh spreading portion of a IS-95C CDMA transmitter;
FIG. 4 is a block diagram illustrating a Walsh spreading portion of a CDMA transmitter according to an exemplary embodiment of the invention;
FIG. 5 is a block diagram illustrating a CDMA receiver according to an exemplary embodiment of the invention;
FIG. 6 is a graphical representation illustrating a comparison of BER performance with respect to various forms of diversity including downlink diversity according to the invention;
FIG. 7 is a diagram illustrating transmission frequencies in IS-95C multi-carrier implementation;
FIG. 8 is a tabular representation illustrating transmission characteristics for a single user using multi-carrier according to an exemplary embodiment of the invention;
FIGs. 9A, 9B and 9C are block diagrams illustrating portions of a CDMA transmitter according to a first exemplary IS-95C direct spread embodiment of the invention;
FIGs. 10A, 10B and IOC are block diagrams illustrating portions of a CDMA transmitter according to a second exemplary IS-95C direct spread embodiment of the invention,
FIGs. 11 A, I IB and 11C are block diagrams illustrating portions of a CDMA transmitter according to a first exemplary IS-95C multi-carrier embodiment of the invention;
FIGs. 12A, I2B and 12C are block diagrams illustrating portions of a CDMA transmitter according to a second exemplary IS-95C multi-carrier embodiment of the invention;
FIG. 13 is a block diagram illustrating a portion of a CDMA receiver according to a first exemplary IS-95C multi-carrier embodiment of the invention; and
FIG. 14 is a block diagram illustrating a portion of a CDMA receiver according to a second exemplary IS-95C multi-carrier embodiment of the invention.
Detailed Description of Preferred Embodiments
As will be explained in detail, the present invention provides methods and apparatus for achieving improved diversity on the downlink path between a base station and mobiles in a CDMA system. This is achieved by employing multiple transmit
antennas at a common base station and a set of 2[log2(k) Walsh codes in accordance with a transmission matrix T, as will be explained, where there are K mobiles associated with the base station. In this manner, downlink diversity is advantageously achieved without requiring an additional receive antenna at any mobile, without wasting bandwidth, and without causing self-interference.
Consider the system described above with respect to equations (1) through (5), but now let us consider two transmit antennas and, for the sake of simplicity, two users. For the moment, we shall ignore the pilot. On the first antenna we transmit:
It is to be appreciated that sx(t) represents a data signal associated with the first user and .v2(t) represents a data signal associated with the second user. Further, s1(t)* represents the complex conjugate of the data signal associated with the first user and s2(t)* represents the complex conjugate of the data signal associated with the second user. It is to be understood that the use of the asterisk (*) in connection with a variable or function is intended to denote the complex conjugate of that variable or function.
At the receiver, we uncover and correlate with the two Walsh codes. At the output of the two Walsh correlations we obtain (dropping the dependence on symbol interval):
which introduces interference terms in the decision statistics. However, assuming we have estimates of the channel distortions /^and h2, we can obtain a signal estimate for the first mobile as follows:
It is to be appreciated that this is identical to the decision statistic for two-antenna diversity reception (without the 3dB aperture gain). Accordingly, we advantageously achieve diversity gain without employing additional antennas at the receiver, without requiring additional bandwidth, and without causing self-interference. It is to be appreciated that "diversity gain" refers to improved reception due to improvement in the statistical distribution of the signal-to-noise ratio at the mobile.
The techniques of the present invention can also be extended to an arbitrary number of antennas. Some important aspects of the techniques of the invention may be represented in the context of a transmission matrix T. If we let rows represent the Walsh codes and the columns represent the transmit antennas at a base station, the two-antenna two-user technique can be represented by:
In order to obtain the desired signals at the receiver without interference, the columns of T need to be orthogonal. For example, in the above formulation:
where ti is the ith column of t and ()l represents the transpose conjugate.
To increase the number of users, we simply increase the matrix wherein the columns are orthogonal. One other requirement is that the matrix have 2" rows for some integer n. In other words, for K users we would require 2 loh2-(K)l codes where | • | is the
ceiling operator. That is, n = flog2(AT)"|. The number of antennas M however is
arbitrary provided that M < n. It is likely that M < K , for example, M = 2 or 4. As a further example, if we assume BPSK (binary phase shift keying) modulation with four users and two antennas, we can use the transmission matrix:
This transmission matrix is also key to decoding the received signal. We can represent the vector of Walsh outputs z as:
Walsh diversity as described herein relies on estimates of the channel coefficients to obtain diversity performance and remove interference terms due to other users. The following is an explanation of preferred methods of performing the channel estimation process in a CDMA system employing Walsh diversity.
In one embodiment, the channel estimation process for acquiring the required channel coefficients is to assign a single Walsh code as a pilot channel on each antenna. That Walsh code would be used only for that antenna and we would thus need to use M Walsh codes for channel estimation where M is the number of transmit antennas.
As an example, consider a system with two transmit antennas labeled antenna 0 and antenna 1 and two users (mobiles). We reserve Walsh code 0 for the pilot associated with antenna 0 and Walsh code 1 for the pilot associated with antenna I.
Thus, at the receiver, we obtain after Rf filtering, down-conversion, and uncovering (i.e., removing the covering code):
where Pp is the power associated with either pilot, P, and s, are the power and data signal associated with mobile i, and h1 is the channel seen at the mobile from antenna /. Now, at the output of the Walsh correlator in the receiver for mobiles 1 and 2 (using Walsh codes 2 and 3), we have;
To obtain estimates of S1 and .s'2 we first need estimates of h0 and h1. This is accomplished in a straightforward manner by simply correlating with Walsh codes 0 and 1. Thus, we estimate h0 as:
It is clear that for M transmit antennas, we simply reserve M Walsh codes, one for each of the M pilot channels.
In a system that does not employ pilot channels, but rather uses pilot symbols, we can also acquire channel estimates in a straightforward manner. We simply transmit N known data symbols synchronously on each of the 2walsh Walsh codes. This is illustrated by the following example. Consider a system with two transmit antennas and four users. The received signal after down-conversion is:
where we have assumed real symbols. At the output of the four Walsh correlators in the receiver, we have:
where n1 is the AWGN (additive white gaussian noise) component of correlator output /'. If all the training symbols are 1, i.e., sj= 1, then we simply have:
If all of the training symbols are not equal to unity but are known, we can still obtain channel coefficients. In general, the outputs of the Walsh correlators z can be expressed as:
z = H s + n (27)
where H is the channel matrix which describes the mapping of transmit symbols to Walsh correlators and H is a vector of noise samples. As explained above, the system of the
premultiplying the vector z by the conjugate transpose of the channel matrix which must be estimated. However, it can be shown that equation (27) can be expressed as:
where T is the transmission matrix and h is the channel vector. Now, it can be shown
that T is also orthogonal, i.e., TlT = ml, where m is the number of Walsh codes. Thus,
to obtain channel coefficients we simply transmit a training sequence for all users simultaneously so that T is known. Thus, at each training symbol interval we can estimate the channel as:
The number and frequency of the training symbols are determined by the SNR (signal-to-noise ratio) required in the channel estimate and the rate of the channel. The training
symbols must be inserted often enough to track the changing channel but as seldom as possible so that data capacity is not wasted. Also, by increasing the length N of the training sequence the SNR of the channel estimate is improved since we can average out the effect of noise on each individual sample, but again data capacity is wasted. The size and frequency of the training sequence is thus a design parameter.
Referring now to FIGs. 1 and 2, a generalized block diagram illustrating a CDMA system 100 and a flow chart for implementing a downlink diversity method 200 according to the invention are shown. A base station 102 in the CDMA system 100 is equipped with M transmit antennas 104-1 through 104-M. According to the invention, the M transmit antennas respectively transmit M data signals to K mobiles 106-1 through 106-K (step 202 of FIG. 2). It is to be appreciated that, similar to equations (6) and (7) above, each of the M data signals contain a data signal portion that pertains to each of the K mobiles. Given K mobiles, each transmit signal requires 2 l0S:CM Walsh codes, in accordance with the invention. The relative assignment of Walsh codes is accomplished in accordance with a transmission matrix 7', as explained above. However, each mobile needs only M Walsh codes.
Each mobile contains a single antenna which receives the respective data signals transmitted by the M base station antennas. As shown in FIG. 1, each data channel between a base station transmit antenna and a mobile has some unique channel characteristics (e.g., signal distortion, etc.) associated therewith. As explained above, proper estimation of the channels is important to the estimation of the data signal at the receiver. Thus, the estimate of the channel formed between the transmit antenna 104-1
and the mobile 106-1 is represented as h\, while the estimate of the channel formed
between transmit antenna 104-M and mobile 106-1 is represented by h\f. This notation is extended to the data channels between the AY transmit antennas and the mobile 106-K.
Next, the following steps occur at each of the mobiles, however, such steps are generally explained with respect to the fC mobile. In step 204, the mobile obtains channel estimates h1k through hMk, for example, in one of the ways described above.
Then, in step 206, the mobile uncovers and correlates the received data signals using the M Walsh codes. This step is equivalent to the computation of equations (8) and (9) above with respect to the two antenna/two user scenario. Lastly, in step 208, the estimate of the data intended for the fCh mobile, sK(t) is found. Again, this corresponds to equation (10) or equation (17) above.
IS-95C Direct-Spread Embodiment
Currently, the physical layer draft proposal for IS-95C supports as an option a form of transmit diversity which is known as orthogonal transmit diversity (OTD). This is described in J. Losh, "TR45 Mobile Station-Base Station Compatibility Standard For Dual-Mode Wideband Spread Spectrum Systems (Physical Layer)," IS95C Draft Proposal (Revision 7), February 16, 1999, the disclosure of which is incorporated herein by reference.
In the optional OTD mode, users are assigned two Walsh codes. The data is split into even and odd streams, and coded independently. The two codes are closely related.
If, for example, a usery is assigned a Walsh code, vrv(/), of length N in the normal
mode, then the usery would be assigned two codes which are formed from w;v (/) in the optional OTD mode. These two codes are formed as follows:
In addition, a separate pilot channel is available for each antenna. Currently, OTD mode only supports two antennas, but it is easily extended to four antennas. The physical layer to support OTD for two antennas is described in the above-referenced IS-
95C Physical Layer Draft Proposal (J. Losh at pg. 3-27). The relevant Walsh spreading portions of the physical layer are shown in FIG. 3.
As shown in FIG. 3, an input sequence is multiplexed amongst the four inputs Yn, YQI, YH and YQ2. Starting with an input sequence s(n), the inputs may be represented as: Yn = {*(.v(0), .v(4), 5(8)....}; YQ, = {.v(2), s(6), s(10) •..}; YI2 = {.v(l), A-(5), s(9)....}- and YQ2 = {s(3), .v(7), s(\ 1)....}. The inputs are respectively held in symbol repetition units 402 through 416 for two symbol intervals and multiplied by the indicated signs. As shown, the signs of the symbol repetition units that receive an input are different with respect to Antenna 1 and Antenna 2. For example, input symbol Yn is multiplied by (+ +) signs for Antenna 1 but by (+ -) signs for Antenna 2. The signs incorporate the conjugation and change of sign operation required for Walsh diversity. Similarly, the signs used to multiply the other symbols of the input sequence are different between the two antennas.
In multipliers 418 through 424, a pre-assigned Walsh function is multiplied by a quasi-orthogonal function (QOF, as described in the IS-95C standard). The output of the multiplier 418 is then respectively multiplied with the outputs of the symbol repetition units 402 and 404 in multipliers 426 and 428, while the output of the multiplier 420 is respectively multiplied with the outputs of the symbol repetition units 406 and 408 in multipliers 430 and 432. Also, the output of the multiplier 422 is respectively multiplied with the outputs of the symbol repetition units 410 and 412 in multipliers 434 and 436, while the output of the multiplier 424 is respectively multiplied with the outputs of the symbol repetition units 414 and 416 in multipliers 438 and 440. Again, this is the spreading operation. The in-phase and quadrature portions of the signal are labeled Iin and Qjm respectively. The in-phase portion Iin(,(o) associated with Antenna 1 is summed with the in-phase portion ljn(io) associated with Antenna 1 in summer 442, while the quadrature portion Qin(011) associated with Antenna 1 is summed with the quadrature portion Qin(11) associated with Antenna 1 in summer 444. The outputs of summers 442
and 444 are then mixed with the carrier components for Antenna 1 and transmitted, imilarly, the in-phase portion linm) associated with Antenna 2 is summed with the in-phase portion Im(io) associated with .Antenna 2 in summer 446, while the quadrature portion Qin(oi) associated with Antenna 2 is summed with the quadrature portion Qin
