The field of the invention is that of power electronics. It relates more particularly to the conversion of electronic switching power.

The invention relates to AC-DC converters, that is to say to AC-DC converters or rectifiers, switching type suitable for use as power factor correctors or PFCs with reference to the English expression. Saxon "Power Factor Correction". This type of converters makes it possible to guarantee a level of harmonics, a mass and a lower volume than the autotransformers-rectifiers known as “ATRU” (in reference to the English expression “AutoTransformer Rectifier Unit”) or than the transformers- "TRU" rectifiers (in reference to the English expression "Transformer Rectifier Unit"). Switching converters used as PFCs make it possible to guarantee a high power factor and a very low harmonic rate.

The solutions currently implemented are most often made up of two conversion stages, a PFC stage proper, non-isolated and step-up, and a step-down DC-DC conversion stage, most often isolated. Generally, in three-phase, the system is composed of 3 isolated single-phase conversion chains wired in "delta" or "star" on the three-phase AC bus and interconnected in parallel on the output DC bus. Naturally three-phase conversion systems (full Boost type bridge) do not offer a very high level of efficiency.

An example of an AC-DC converter without an input diode bridge and which can be used as a PFC is disclosed in the patent application US 20100259240. This converter is intended for single-phase applications.

As can be seen in FIG. 1, this converter comprises, for an input voltage V A c applied between an input terminal and a common terminal:

an input inductor L1 comprising a first terminal connected to the input terminal and a second terminal connected to a controllable input switch, bidirectional in voltage S VB , also connected to the common terminal,

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- a branch connected to the second terminal of the input inductor L1 and comprising, in series, an input capacitor Cr and a resonance inductance L r whose value is much lower than that of the input inductance L1,

- a first output diode C R1 comprising an anode connected to the common terminal and a cathode connected to the resonance inductance Lr,

a second output diode C R2 comprising a cathode connected to an output terminal of the converter and an anode connected to the resonance inductor Lr.

The converter is able to deliver a direct voltage V s between the common terminal and the output terminal of the converter connected to an output capacitor C 0 . The output voltage V s supplies a load R. This voltage V s is adjusted by adjusting the duty cycle of the control of the input switch S V B- This converter without a diode bridge at the input, makes it possible to deliver at output, a voltage of the same sign, both for a positive input voltage and for a negative input voltage.

One problem with this topology is the volume of the output capacitor.

An aim of the invention is to provide a topology making it possible to limit the aforementioned problem.

To this end, the invention relates to a switching AC-DC converter intended to deliver a continuous output voltage V out between a first output terminal B1 and a second output terminal B2, said converter comprises at least one string of conversion intended to convert an input alternating voltage applied between an input terminal E and a neutral point N, the conversion chain comprising:

an input inductor L comprising a first terminal of the input inductor connected to the input terminal E and a second terminal of the input inductor,

an input switch S comprising a first terminal connected to the second terminal of the input inductor L, the input switch S being a switch which can be controlled bidirectionally in voltage and in current,

AMENDED SHEET (ARTICLE 19)

an LC circuit comprising a first terminal connected to the second terminal of the input inductor L and a second terminal connected to an intermediate point PI, the LC circuit comprising between its first terminal and its second terminal, a resonance inductance L c of lower value than that of the input inductance L, and a link capacitor C connected in series with the resonance inductance L c ,

a first output switch D1 comprising a first terminal connected to the first output terminal B1 and a second terminal connected to the intermediate point PI, allowing current to flow only from the intermediate point PI to the first output terminal B1,

a second output switch D2 comprising a first terminal connected to the second output terminal B2 and a second terminal connected to the intermediate point PI, the second switch allowing current to flow only from the second terminal B2 to the intermediate point,

the converter further comprising:

a first output capacitor Ci comprising a terminal connected to the first output terminal B1 and another terminal connected to a second terminal of the input switch S,

- a second output capacitor C 2 with the same capacity as the first output capacitor C 1 and with a capacity greater than the capacity of the link capacitor C, the second output capacitor C 2 comprising a terminal connected to the second output terminal B2 and another terminal connected to the second terminal of the input switch S.

Advantageously, the converter according to the invention comprises at least one of the following characteristics taken alone or in combination:

the converter is intended to transform a polyphase voltage comprising several phases into a direct voltage, said converter comprising several identical conversion chains, each conversion chain receiving one of the phases on its input terminal,

- the second terminal of the input switch is not connected to the neutral point,

- the converter comprises control means configured to control the input switch S so that the chain of

AMENDED SHEET (ARTICLE 19)

conversion operates as a step-down during a converter start-up phase,

- the converter comprises control means configured to control the input switch S so that, during a starting phase of the converter, the output voltage Vout is a monotonically increasing function of time and reaches a setpoint voltage on passing by a non-zero intermediate output voltage lower than the setpoint voltage then to control the converter, during a steady state phase, starting when the output voltage reaches the setpoint voltage, so as to keep the output voltage substantially fixed at the setpoint voltage for a non-zero period,

- the control means are configured so that the converter start-up phase includes a phase of increasing the output voltage V out of the converter with a DC derivative from an initial voltage to the setpoint voltage,

- the input inductor is connected to the input terminal via a low pass filter,

- the resonance inductor has a magnetic circuit with a cut-off frequency of less than 10 MHz,

- the output capacitors have a capacitance at least one hundred times greater than the capacitance of the link capacitor and the input inductance has an inductance value at least a thousand times greater than the inductance value of the inductor resonance.

The invention also relates to a method for controlling an AC-DC converter according to the invention. This method comprises a step of starting the converter during which the input switch S is controlled so that the conversion chain operates in step-down mode.

The method advantageously comprises at least one of the following characteristics taken alone or in combination:

- The method comprises a starting phase of the converter starting at the start of the converter, during which the input switch S is controlled so that the output voltage V out is a monotonically increasing function of time and reaches a reference voltage in passing

AMENDED SHEET (ARTICLE 19)

by a non-zero intermediate output voltage lower than the setpoint voltage, the method comprising a steady state phase, starting when the output voltage reaches the setpoint voltage, during which the switch is controlled so as to maintain the voltage output substantially fixed at the setpoint voltage for a non-zero period,

- the starting phase comprises a phase of increasing the output voltage Vout of the converter with a DC derivative from an initial voltage to the setpoint voltage,

- the conversion chain operates as an elevator during a steady state phase subsequent to the start-up phase.

Other characteristics and advantages of the invention will become apparent on reading the detailed description which follows, given by way of non-limiting example and with reference to the appended drawings in which:

- Figure 1 already described shows an electrical diagram of a converter of the prior art,

- Figure 2 schematically shows an electrical network comprising a single-phase AC-DC converter,

- Figure 3 shows schematically the conversion chain of the converter of Figure 2,

FIG. 4 diagrammatically represents a conversion assembly of a converter intended to convert a three-phase voltage into a direct voltage,

- Figures 5a and 5b represent the shapes of the currents and voltage in the conversion chain of Figure 3 as a function of time,

FIG. 6 represents an example of the control means in the case of a converter intended to convert a polyphase voltage into a direct voltage,

FIG. 7 represents another example of the control means in the case of a converter intended to convert a polyphase voltage into a direct voltage.

From one figure to another, the same elements are identified by the same references.

AMENDED SHEET (ARTICLE 19)

The invention relates to a switching AC-DC converter of the type without diode bridge at input and which can be used as a power factor correction circuit or RFC converter with reference to the English expression "Power factor. corrector ”. As represented in FIG. 2, the converter 100 comprises an AC-DC conversion chain intended to be connected directly at the input to a line delivering an alternating voltage V in and to delivering a direct voltage V out between two output terminals B1 and B2 of the converter 100. The current t delivered by the converter 100 supplies a load Z which may or may not be purely resistive.

In FIG. 3, there is shown an example of a conversion chain CH intended to convert the AC input voltage V in applied between an input terminal E and a neutral point N into a DC voltage V out delivered between the first terminal output B1 and the second output terminal B2 of the converter. The CH conversion chain includes:

an input inductor L comprising a first terminal of the input inductor connected to the input terminal E and a second terminal of the input inductor,

an input switch S comprising a first terminal connected to the second terminal of the input inductor L, the input switch S being a switch which can be controlled bidirectionally in voltage and in current,

an LC circuit comprising a first terminal connected to the second terminal of the input inductor L and a second terminal connected to an intermediate point PI, the LC circuit comprising between its first terminal and its second terminal, a resonance inductance L c of lower value than that of the input inductance L, and a capacitor C, called a link capacitor, connected in series with the resonance inductor L c ,

a first output switch Di, comprising a first terminal connected to the first output terminal B1 and a second terminal connected to the intermediate point PI, configured to allow current to flow only from the intermediate point PI to the first output terminal B1, the first output switch D 1 is, in the example of FIG. 3, a diode comprising a cathode connected to the first output terminal B1 and an anode connected to the intermediate point PI,

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- a second output switch D 2 comprising a first terminal connected to the second output terminal B2 and a second terminal connected to the intermediate point PI, the second switch being configured to allow current to flow only from the second terminal B2 to the intermediate point , the second output switch D 2 is, in the example of FIG. 3, a diode comprising an anode connected to the second output terminal B2 and a cathode connected to the intermediate point PI.

According to the invention, the converter 100 further comprises:

a first output capacitor Ci comprising a terminal connected to the first output terminal B1 and another terminal connected to a second terminal of the input switch S,

- a second output capacitor C 2 of the same capacity c2 as the first output capacitor Ci (of capacity c1) and of capacity c2 greater than the capacity c of the link capacitor C, the second output capacitor C 2 comprising a connected terminal to the second output terminal B2 and another terminal connected to the second terminal of the input switch S.

The output capacitors Gi, C 2 can be polarized, that is to say each comprise a positive terminal and a negative terminal, but this is not compulsory. In the case of FIG. 3 where the capacitors are polarized, the positive terminal of the first output capacitor Ci is connected to the first output terminal B1 and its negative terminal is connected to the positive terminal of the second output capacitor C 2 and to the second terminal of the input switch S, the positive terminal of the second output capacitor C 2 is connected to the second output terminal B2.

The output capacitors have a capacity at least one hundred times higher than the capacity of the link capacitor and preferably at least one thousand times higher (cl> 1000 * c and c2 = cl).

Likewise the input inductance L has an inductance value I at least a thousand times higher than the value le of the inductance Le. In other words, l ³ 1000 * le.

The input switch S is a bidirectional switch that can be controlled in terms of voltage and current. In other words, this input switch S is able, in an open state, to block a current whatever the polarity of the voltage at its terminals and, in a closed state, to conduct the

AMENDED SHEET (ARTICLE 19)

current in both directions from its first terminal to its second terminal and from its second terminal to its first terminal. Such a switch comprises, for example, two transistors connected in series with a common source. These transistors can be insulated gate bipolar transistors or IGBT, from the English Insulated Gâte Bipolar transistor or field effect transistors or F ET with reference to the English expression "field effect transistor" such as for example transistors N type insulated gate field effect or MOSFET (acronym for "Metal Oxide Semiconductor Field Effect Transistor".

As a variant to the embodiment of FIG. 3, at least one of the output switches Di , D 2 is a switch controlled so as to obtain letting the current flow only in one direction.

Advantageously, at least one output switch comprises a controlled switch connected to B1 or B2 and to PI. This controlled switch is controlled so as to allow current to flow only in the desired direction. To this end, the switch comprises control means configured to control the output switch so that the latter allows current to flow only in the desired direction. This solution is more complex than the use of diodes but has an interest in the field of high powers. The switch may for example comprise a thyristor or an insulated gate field effect transistor more commonly known as MOSFET (acronym for the expression “Metal Oxide Semiconductor Field Effect Transistor”) with synchronous rectification.

The converter 100 also comprises, as represented in FIG. 2, COM control means capable of controlling the input switch S visible in FIG. 3. These COM control means comprise ECOM control establishment means for generating a duty cycle control d and actuating means ACT comprising at least one actuator for actuating the input switch S. In other words, the actuator is actuated by the ECOM command establishment means so as to actuate the 'input switch S so that at each switching period T, the input switch S is closed from time 0 to time d * T and open from time d * T to time T. The input switch S is then open, at each switching period, for a period of* T, where d '= 1 -d.

AMENDED SHEET (ARTICLE 19)

The open or closed state of the input switch S has an effect on the blocking or conduction of the current through each of the two output switches, D 1 and D 2 , of the conversion chain. Di, D 2 are configured and arranged (in the case of diodes) or able to be controlled to block or conduct the current according to the polarity of the voltage at the input of the conversion chain so that a direct current of the same polarity is obtained at the output of the conversion chain whatever the polarity of the voltage at the input of the conversion chain.

The ratio between the maximum input voltage (peak of the voltage in AC) and the output voltage V out depends on the duty cycle of the input switch S.

The resonant inductance L c and the link capacitor C form a resonant circuit LC when the input switch S is closed, regardless of the polarity of the input current. This resonant circuit is active only during a resonance half-period, when switch S is closed. The resonance period Tr of the LC circuit is given by the following formula:

Tr = 2 * p *] Le * C

The topology of FIG. 3 makes it possible to output a voltage of the same sign, and adjustable regardless of the sign of the input voltage V in.

Due to the arrangement of the output capacitors according to the invention, the conversion chains, when they are controlled to be elevating, are slightly elevating. They are two times less elevating than Vienna bridge topologies of the type described in the article “3-Phase Power Factor Correction, Using Vienna Rectifier Approach and Modular Construction for Improved Overall Performance, Efficiency and Reliability”, Mr. Abhijit D. Pathak, et Al. Indeed, as we will see hereafter, V out = V inc / (1 -d) where V inc is the peak value of the AC input voltage V in . The link capacitor C of the LC circuit charges to ± V 0UÎ / 2 (according to the sign of V in). This low-rise solution can make it possible to dispense with a voltage step-down stage depending on the desired output voltage.

The presence of a single bidirectional input switch for voltage and current, the absence of an input rectifier bridge (full bridge of

AMENDED SHEET (ARTICLE 19)

input diodes) and the simplicity of the topology make it possible to achieve a high level of integration and a high conversion efficiency.

When the converter is used in RFC, the input switch is controlled so that the current drawn l L at the input is proportional and in phase with the input voltage V in . This makes it possible to obtain a low level of harmonics and therefore a high power factor.

In a polyphase system, as the capacities of the output capacitors Ci and C 2 are identical or substantially identical, the point M, that is to say the point of connection between the two output capacitors Ci and C 2 or the midpoint , is a fictitious neutral with the same potential as the neutral point N of the input network.

The invention therefore finds particular utility for vehicles (large aircraft, weapon planes, surface vessels, submarines, surface or submarine drones, etc.), including the neutral of the on-board generation (or on-board network) is not distributed or in which it is not possible to draw current from the neutral. Another advantage concerns security. Indeed, a fault between a phase and the neutral results in an overcurrent which it is easy to detect in order to protect the network. The proposed topology makes it possible to avoid a connection to the mechanical ground. An advantage of this configuration is to avoid the circulation of certain harmonics on the network. For example, in the case of a three-phase network, the 3rd harmonic easily transits on the aircraft neutral.

The M point and the neutral point may or may not be linked according to the desired advantages.

Another object is to provide an AC / DC converter intended to receive a polyphase voltage as input, allowing its use in higher power applications.

A variant is therefore proposed in which the converter is intended to convert a polyphase voltage conveyed by a polyphase line into a direct voltage delivered between the two output terminals of the converter. The converter then comprises a conversion assembly comprising several conversion chains transforming

AMENDED SHEET (ARTICLE 19)

each an alternating voltage, corresponding to one of the phases of the polyphase line, applied between the two input terminals of this conversion chain, in a direct voltage delivered between the two output terminals B1 and B2 of the converter.

Such an ENS conversion assembly is represented in FIG. 4. This ENS conversion assembly comprises several identical conversion chains CH1, CH2, CH3. Each of these chains is identical to the conversion chain of FIG. 3. These chains are connected to the output capacitors Ci and C 2 of the ENS conversion assembly, ie of the converter, in the same way as the output capacitor. CH chain. The different conversion chains are connected to different phases of the polyphase line (phases 1, 2 and 3 in figure 4) and subjected to their respective input terminals Ei, E 2 , E 3at respective voltages V1, V2, V3 corresponding to the voltages of the respective phases of the polyphase line. The number of conversion chains is equal to the number of phases of the input voltage network. FIG. 4 illustrates a non-limiting example of a power converter comprising three conversion chains intended to be connected to a three-phase input network, but the number of phases of the input polyphase line and therefore the number of conversion chains could be different. Each of these chains comprises an intermediate point PU, PI2, PI3 of the chain.

As the phases are balanced (the network delivering phases of the same amplitude and the same frequency but out of phase) and as the conversion chains are all identical, the current absorbed on the different phases is the same (except for the phase shift between the phases). The point M is therefore a fictitious neutral with the same potential as the neutral N of the input network, even if the latter is not distributed, that is to say even if these points M and N are not connected to one another.

We will now describe the operation of the CH conversion chains with reference to FIG. 3 and to FIGS. 5a and 5b.

The value of the input inductance L is much greater than that of the resonant inductance L c and the capacitance of the input capacitor is low compared to the capacitance of each of the capacitors Ci and C 2 , the capacitors Ci and C with identical capacity.

AMENDED SHEET (ARTICLE 19)

The switching period T is much less than the period of the AC input voltage V in .

The analysis consists of dividing the breakdown period into three operating phases. The input switch S, the inductive and capacitive components are considered here as perfect.

Operation is described during a positive half-wave of the input voltage. During a negative half-wave, the currents are reversed and the roles of diodes Di and D 2 are also exchanged (D 1 is used to reverse the charge of C and D 2 conducted for a period of (1 -d) * T).

We place ourselves in a so-called steady state phase during which the switch S is controlled so that V out is fixed (output voltage in steady state) and we consider a chopping period during which the input voltage V in can be considered constant (because the evolution of the voltage V in is slow compared to a switching period).

The operation is the same for each of the chains in figure 4.

Phase 1

The initial time t = 0 corresponds to the start of the chopping period T.

The input switch S is closed from 0 to t1 which blocks the diode D 2 . As can be seen in FIG. 5b, the voltage V c at the terminals of C decreases by resonance between L c and C until the current in the diode D 2 is canceled, which is blocked naturally while S remains closed.

The current I s is the current flowing in the input switch S towards the midpoint M. The current I LC is the current absorbed by the LC circuit. The current I L is the current absorbed by the conversion chain:

At any time I s (t) = - c (+ 4 (4

4 (t) = Vin * t / L + 4 (t = 0)

4 (t) believes as shown in Figure 5a in dotted lines.

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with w

! l c * c

D2 hangs at time t1:

tl = p * jL c * C

which is only valid if

V c (t = 0) is greater than v ° ut / 2- Thus the absolute value of 4 c (t) increases

until the diode is blocked, then decreases to t1 as can be seen in FIG. 5a in solid lines.

Phase 2

The input switch S is always closed but D 2 is blocked from t1 to d * T with T the switching period (T = 1 / F where F is the switching frequency).

As visible in figure 5a:

At t1, l L c = 0 as visible in figure 3, l out = 0 and l s (t) = l L (t) l L (t) = V in * t / L +! I (t = t1 ) the load of the inductance L is linear as visible in figure 5a.

Phase 3

The switch S is open at the instant t2 = d * T and the remainder until the end of the chopping period T.

During this third phase, part of the energy stored in L is returned to C and to C1. This duration (Td * T) is very short compared to the period of the resonant frequency of the LC circuit. We find ourselves working around w * ΐ = p / 2 with w = V, - - -. The shape of the current L

/ (L c + L) * C

is quasi-linear as can be seen in FIG. 5a.

During the third phase, we can therefore write:

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(î) = Le (t),

The current discharge of the input inductance L is quasi-linear and the current i [_ (t) in L decreases, while C is charged in voltage and V c (t) increases in a quasi-linear way as visible in figure 5b.

It is noted that for the average charge of the connection capacitor C, that is to say V c average, to be constant, there must be equality between the energy released during phase 1 and that released during phase 1. phase 3; this corresponds to an equality between the positive and negative areas of the current e shown in hatched lines in FIG. 5a.

It can be seen that at each chopping period T, the same energy is transferred to the two output capacitors Ci and C2 of the same capacity, to C2 from t = 0 to t1 and to Ci from t = d *T to T. Thus the capacitors are charged at the switching frequency and not at the network frequency much lower than the switching frequency. This configuration makes it possible to limit the storage of energy and therefore the size of the capacitors, which makes it possible to achieve a high level of integration and / or to deliver a stable output voltage, by limiting the ripples of the output voltage ( the average voltage is the same at the terminals of the two capacitors during a chopping period). It should be noted that in the case of an assembly of the Vienna bridge type as in the article "3-Phase Power Factor Correction, Using Vienna Rectifier Approach and Modular Construction for Improved Overall Performance, Efficiency and Reliability", Mr. Abhijit D. Pathak, et al, one of the output capacitors is charged on the positive half-wave of the network and the other output capacitor is charged on the negative half-wave of the network. In the assembly according to the invention, the charging of C1 and C2 is therefore carried out at the switching frequency instead of the network frequency.

Determination of the output voltage V t:

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The amplitude of the variation in current l L is the same between 0 and d * T as between d * T and T in a regime during which the current l L at the start of the period is identical to that at the end of the period.

From t = 0 to d * T: AI L = V in * d * T / L

From t = d * T to T: AI L = (V out / 2 + V c - V in ) * (1 - d) * T / L = (V out -V in ) * (1 - d) * T / L with V c = V out / 2

We deduce: (V out - V in ) * (1 - d) = V in * d that is to say:

V out = V in / (1 - d)

This is the equation of a step-up converter (or "Boost" in Anglo-Saxon terminology).

This condition is verified if the product d * T is greater than or equal to Tr / 2 (Tr being the resonance period of the LC circuit formed by L c and C), otherwise the discharge of the capacitor C is not possible during the period cutting, that is to say between 0 and d * T, and the elevator operation described above cannot be installed.

When d * T is less than Tr / 2, then the discharge of the capacitor C is not possible between 0 and d * T. The converter then operates in step-down.

Thus, the step-up operation is obtained when the ECOM control establishment means control the input switch S so that the duty cycle d is greater than or equal to Tr / 2. The ECOM command establishment means can also be configured so that V out is equal to a predetermined reference value V ¥ ns .

In other words, there can be no resonance of L c and C and therefore step-up operation as long as the load of C (V c ) has not reached the value of the voltage V out / 2 (i.e. the charge of C 2 ).

In regulated step-up operation, that is to say when the output voltage is fixed, the average voltage Vc at the terminals of C is equal to V out / 2.

In single-phase or polyphase alternating current, the voltage V in used in the equations given above is the peak value of the phase-to-neutral voltage of the network (between phase and neutral). In step-up operation, the voltage

AMENDED SHEET (ARTICLE 19)

output voltage V out is therefore always greater than the maximum peak value of the phase-to-neutral voltage between phase and neutral V in .

Commutations:

When the input switch S closes, the inductors L and Le oppose the establishment of the current and switching is carried out by zero current also called Zero Current Switching or ZCS in English terminology.

The input switch S is not perfect, it has a parasitic capacitance, the opening of this switch then loads its parasitic capacitance. The inductance Le opposes the establishment of the current to the diode D 1 . This results in an oscillating regime or switching noise due to the resonance of the parasitic capacitance of the input switch S with the inductance Le, which can lead to overvoltages at the terminals of the switch. input S. The frequency of the oscillations can be very high (> 1 ÛMHz)

In order to limit the amplitude and duration of the switching noise due to the blocking of the input switch, the inductor Le can be configured so that its magnetic circuit at least partially dissipates the energy of the switching noise so to limit and preferably eliminate the overvoltages when the output switch S is blocked. For this purpose, the magnetic circuit of the inductor Le has a cut-off frequency of less than 10 MHz. The frequency of the switching noise is typically greater than 10 MHz.

The resonance inductance Le, which contributes to the switching noise, comprises for example a magnetic circuit with a distributed air gap. This type of inductance is, for example, made of a material with low magnetic permeability, for example less than 100 n H / turn 2 . The low energy of the switching noise, at a frequency greater than a few MHz, will be dissipated in the magnetic material.

Use in RFC

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When the voltage V c crosses zero , the charge on C must be reversed, which cannot be done instantly and may affect the shape of the current absorbed.

The values ​​of the components Le and C are advantageously chosen to ensure the reversal of the voltage Vc on the capacitor C in a sufficiently short time to allow a range of variation of the duty cycle large enough to maintain the dynamic of the input voltage and to limit the peak current l s flowing through the input switch. This makes it possible to limit a consequent oversizing of the input switch.

Start-up

When the converter starts up, that is to say when the input voltage V in is established on the input terminal, when the output switch S is not controlled, a call for current inherent in switches D 1 and D 2(the converter behaves like a rectifier). This current draw is traditionally limited by providing a pre-charge device downstream or upstream of the converter of the type comprising a switch or a contactor short-circuiting, in steady state, resistors connected in series with the phases or a switch. (transistor) power on the DC bus, which short-circuits, in steady state, a resistor connected in series, upstream of the output capacitors. The pre-charge device allows the charging of the output capacitors at the start of the converter. However, this type of pre-charge device is bulky.

An advantage of the converter according to the invention is to make it possible to adjust the voltage V out delivered to a predetermined value between zero and a value greater than the peak value of the phase-to-neutral voltage, which can make it possible to limit the current inrush. starting by integrating the rise of the output voltage over a long time of several AC periods of V in . This applies equally well in the single phase as in the polyphase case.

The inrush current limitation at start-up is achieved by a judicious control of the input switch.

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Indeed, in the absence of chopping, the capacitors C, Ci and C 2 charge to voltage values ​​inversely proportional to their capacities. Consequently, V c is almost equal to V in because C is of very low value compared to Ci and C 2 . The output voltage V out then remains minimal (in the inverse ratio of the values ​​of the capacitances of the capacitors C and Ci or C) according to the considered half-wave.

When the converter cuts out, the link capacitor C can remain charged at an average voltage greater than V out / 2, if the duty cycle d is adjusted so as to prevent its complete discharge during the start of the switching period T, the voltage output V out is then equal to V in - V c and Vou t is all the lower as V c is large.

This is only possible if the duty cycle d is such that d * T is of a duration less than the half-period of resonance of the LC circuit (couple C and Le), that is to say * T