1
FORM 2
THE PATENTS ACT, 1970
(39 of 1970)
&
THE PATENTS RULES, 2003
COMPLETE SPECIFICATION
[See section 10, Rule 13]
MOTOR DRIVE APPARATUS;
MITSUBISHI ELECTRIC CORPORATION, A CORPORATION ORGANISED
AND EXISTING UNDER THE LAWS OF JAPAN, WHOSE ADDRESS IS 7-3,
MARUNOUCHI 2-CHOME, CHIYODA-KU, TOKYO 1008310, JAPAN
THE FOLLOWING SPECIFICATION PARTICULARLY DESCRIBES THE
INVENTION AND THE MANNER IN WHICH IT IS TO BE PERFORMED
2
DESCRIPTION
Field
[0001] The present invention relates 5 to a motor drive
apparatus that drives a synchronous motor.
Background
[0002] A servo system using a synchronous motor is used
10 as a power source for various mechanical devices. In a
general servo system, a speed controller and a current
controller are connected in series. A limiter is provided
at the output of each controller to protect the synchronous
motor and the mechanical device. Moreover, in general, a
15 power converter that outputs an alternating current voltage
to the synchronous motor has a limit on the maximum voltage
that can be output or a limit on the maximum current that
can be output. Such a limit of the power converter also
functions similarly to the limiter.
20 [0003] Each controller is provided with an integrator to
control the output so as to eliminate a steady-state error.
When the output of each controller is saturated by the
limiter, and the integrated value becomes excessive by
continuing the integration, a windup phenomenon is known to
25 occur in which the output value does not change from the
limit value even if a command value changes. The windup
phenomenon may excite sustained oscillation. The windup
phenomenon causes a reduction in the stability of control
by the servo system. As one method of preventing the
30 windup phenomenon, when it is detected that the output of
each controller is saturated, the command value input to
each controller can be lowered such that the saturation is
released.
3
[0004] Patent Literature 1 discloses a control method
that is related to a speed control device of a motor and
that reduces a speed command value when an output voltage
of a power converter reaches an upper limit to result in
saturation of the output voltage. When 5 the output voltage
of the power converter is saturated, the speed control
device according to Patent Literature 1 performs voltage
phase control such that the phase of a voltage command
value becomes a lead phase with respect to dq rotating
10 coordinates, and reduces the speed command value by
performing calculation that corrects the speed command
value when the phase angle of the voltage command value is
determined to have exceeded a threshold.
15 Citation List
Patent Literature
[0005] Patent Literature 1: Japanese Patent No. 5256009
Summary
20 Technical Problem
[0006] When the control method described in Patent
Literature 1 is applied, a motor drive apparatus can
prevent the occurrence of the windup phenomenon by
appropriately adjusting control parameters. However,
25 according to the control method described in Patent
Literature 1, many control parameters need to be adjusted.
In addition, a servo system or plant model includes many
non-linear elements that exhibit complex characteristics.
Thus, according to the control method described in Patent
30 Literature 1, the motor drive apparatus adjusts the control
parameters by trial and error and thus is problematic in
terms of a heavy workload required for the adjustment to
perform stable control on the motor.
4
[0007] The present invention has been made in view of
the above, and an object of the present invention is to
provide a motor drive apparatus capable of reducing a
workload required for an adjustment to perform stable
control 5 on a motor.
Solution to Problem
[0008] In order to solve the above-described problem and
achieve the object, a motor drive apparatus according to
10 the present invention includes: a current controller to
convert a value of a phase current flowing through a motor
into values of a d-axis current and a q-axis current that
are currents in a dq coordinate system, and control the
phase current by determining a voltage command on the basis
15 of the d-axis current and a d-axis current command as well
as the q-axis current and a q-axis current command; a
voltage amplitude calculating unit to obtain a voltage
amplitude that is an amplitude of the voltage command; a
speed controller to control a rotational speed of the motor
20 by determining the q-axis current command on the basis of a
speed command, the rotational speed, and a speed droop
amount that reduces the speed command; a flux weakening
controller to perform flux control to limit an amplitude of
a voltage output to the motor, by determining the d-axis
25 current command on the basis of the voltage amplitude and a
first voltage limit value; and a speed droop controller to
control the speed droop amount on the basis of the voltage
amplitude and a second voltage limit value. The speed
droop controller determines the speed droop amount that
30 makes the voltage amplitude smaller than the second voltage
limit value.
Advantageous Effects of Invention
5
[0009] According to the present invention, the motor
drive apparatus has an effect that the workload required
for the adjustment to perform stable control on the motor
can be reduced.
5
Brief Description of Drawings
[0010] FIG. 1 is a block diagram illustrating an example
of a configuration of a motor drive apparatus according to
a first embodiment of the present invention.
10 FIG. 2 is a block diagram illustrating an example of a
configuration of a speed controller included in the motor
drive apparatus according to the first embodiment.
FIG. 3 is a diagram for explaining a voltage vector
representing a voltage state of a synchronous motor to be
15 controlled by the motor drive apparatus according to the
first embodiment.
FIG. 4 is a block diagram illustrating an example of a
configuration of a flux weakening controller included in
the motor drive apparatus according to the first embodiment.
20 FIG. 5 is a block diagram illustrating an example of a
configuration of a speed droop controller included in the
motor drive apparatus according to the first embodiment.
FIG. 6 is a diagram illustrating an example of a
control model of the motor drive apparatus and the
25 synchronous motor according to the first embodiment.
FIG. 7 is a diagram illustrating a model that
approximates the control model illustrated in FIG. 6 in the
vicinity of an operating point in a high-speed region.
FIG. 8 is a first diagram for explaining the design of
30 a flux weakening control gain of the flux weakening
controller illustrated in FIG. 4.
FIG. 9 is a second diagram for explaining the design
of the flux weakening control gain of the flux weakening
6
controller illustrated in FIG. 4.
FIG. 10 is a third diagram for explaining the design
of the flux weakening control gain of the flux weakening
controller illustrated in FIG. 4.
FIG. 11 is a fourth diagram for explaining 5 the design
of the flux weakening control gain of the flux weakening
controller illustrated in FIG. 4.
FIG. 12 is a first diagram illustrating the design of
a speed droop control gain of the speed droop controller
10 illustrated in FIG. 5.
FIG. 13 is a second diagram illustrating the design of
the speed droop control gain of the speed droop controller
illustrated in FIG. 5.
FIG. 14 is a third diagram illustrating the design of
15 the speed droop control gain of the speed droop controller
illustrated in FIG. 5.
FIG. 15 is a fourth diagram illustrating the design of
the speed droop control gain of the speed droop controller
illustrated in FIG. 5.
20 FIG. 16 is a graph illustrating an example of
operating waveforms when the motor drive apparatus
according to the first embodiment is used.
FIG. 17 is a diagram illustrating an example of a
hardware configuration of a motor drive apparatus according
25 to a second embodiment of the present invention.
Description of Embodiments
[0011] A motor drive apparatus according to embodiments
of the present invention will now be described in detail
30 with reference to the drawings. Note that the present
invention is not limited to the embodiments.
[0012] First Embodiment.
FIG. 1 is a block diagram illustrating an example of a
7
configuration of a motor drive apparatus according to a
first embodiment of the present invention. A motor drive
apparatus 100 according to the first embodiment drives a
synchronous motor 1. The motor drive apparatus 100 is
connected to a power converter 3. The synchronous 5 motor 1
is mechanically connected to a mechanical device 2. The
synchronous motor 1 is a power source for the mechanical
device 2. The mechanical device 2 operates when the power
converter 3 outputs an alternating current voltage to the
10 synchronous motor 1. The synchronous motor 1, the power
converter 3, and the motor drive apparatus 100 make up a
motor system that drives the synchronous motor 1.
[0013] In the first embodiment, the synchronous motor 1
is a permanent magnet synchronous motor in which a
15 permanent magnet is provided on a rotor. The synchronous
motor 1 may be a wound field synchronous motor in which a
field winding is wound around a rotor, or may be a
reluctance synchronous motor that obtains rotational torque
by utilizing saliency of a rotor. The arrangement of the
20 permanent magnet in the synchronous motor 1 may be of an
embedded type or a surface type. In the first embodiment,
the synchronous motor 1 is assumed to be a three-phase
synchronous motor. The synchronous motor 1 may be a
synchronous motor other than the three-phase synchronous
25 motor. For example, the synchronous motor 1 may be a twophase
synchronous motor or a five-phase synchronous motor.
[0014] The mechanical device 2 need only be a device
that operates when the synchronous motor 1 is driven. In
the first embodiment, the mechanical device 2 is assumed to
30 be a refrigerant compressor that is a typical example of an
application whose control adjustment tends to take time.
The refrigerant compressor is incorporated into an
appliance such as an air conditioner, a chiller, or a
8
refrigerator. Many refrigerant compressors include an
integrated structure in which a motor is incorporated in
order to reduce the number of parts. Thus, in many
refrigerant compressors, it is difficult to perform the
control adjustment by the motor alone. 5 Moreover, the
refrigerant compressor has a pressure condition that
changes gradually with respect to time, and thus takes time
for the pressure to stabilize. Since it takes time for the
pressure to stabilize, the control adjustment of the
10 refrigerant compressor tends to take a long time.
[0015] The refrigerant compressor includes various types
of compressors such as a rotary compressor, a scroll
compressor, a screw compressor, a reciprocal compressor,
and a turbo compressor. It is common for any type of
15 refrigerant compressor to require complicated control
adjustment. The refrigerant compressor as the mechanical
device 2 may be any of the various types of compressors.
The mechanical device 2 may be a device other than the
refrigerant compressor.
20 [0016] The power converter 3 converts power input from a
power source (not illustrated) into power of a prescribed
form, and outputs the power. In the first embodiment, the
power converter 3 is assumed to be a general-purpose
voltage source inverter. The voltage source inverter is a
25 device that switches and converts a direct current voltage
supplied from a direct current voltage source into a
desired alternating current voltage. The power converter 3
converts the direct current voltage into the alternating
current voltage on the basis of a voltage command 12 output
30 from the motor drive apparatus 100, and outputs the
alternating current voltage obtained by the conversion to
the synchronous motor 1. Note that the power converter 3
may be another type of circuit such as a current source
9
inverter or a matrix converter, or may be a multi-level
converter as long as desired alternating current power can
be supplied to the synchronous motor 1.
[0017] A current detection unit 4 detects a phase
current flowing through the synchronous motor 5 1. The type,
arrangement, and the like of the current detection unit 4
are not particularly limited. The current detection unit 4
may be a current sensor of a type using a transformer
called a current transformer (CT), or may be a current
10 sensor of a type using a shunt resistor. The current
detection unit 4 may use a combination of the CT and the
shunt resistor. The current detection unit 4 illustrated
in FIG. 1 is disposed at a wiring between the synchronous
motor 1 and the power converter 3, and measures the phase
15 current flowing through the synchronous motor 1. The
current detection unit 4 outputs a signal 11 indicating the
value of the phase current. Note that the current
detection unit 4 may be disposed at a position other than
the position illustrated in FIG. 1. For example, the
20 current detection unit 4 may be disposed inside the power
converter 3.
[0018] In a case where the current detection unit 4 is
disposed inside the power converter 3, the current
detection method that can be used includes a one-shunt
25 current detection method in which a shunt resistor is
disposed on an N side of a direct-current bus of the power
converter 3, a lower-arm shunt current detection method in
which a shunt resistor is inserted in series with a lower
arm of the power converter 3, or the like. As compared to
30 the case of using the CT, the one-shunt current detection
method and the lower-arm shunt current detection method
have a limit on the timing at which the current can be
detected, but can reduce the component cost.
10
[0019] In a case where the synchronous motor 1 is a
three-phase synchronous motor, on the basis of Kirchhoff's
current law, the motor drive apparatus 100 uses the value
of the phase current of any two of the three phases to be
able to calculate the value of the phase 5 current of the
other one phase. Therefore, the current sensor need only
be disposed for any two of the three phases, and need not
be disposed for the other one phase.
[0020] The motor drive apparatus 100 controls the
10 synchronous motor 1 by vector control. The motor drive
apparatus 100 includes a position/speed specifying unit 5,
a speed controller 6, a dq-axis current controller 7, a
voltage amplitude calculating unit 8, a flux weakening
controller 9, and a speed droop controller 10.
15 [0021] In order to perform vector control on the
synchronous motor 1, a magnetic pole position θe and a
rotational speed ωe of the synchronous motor 1 need to be
detected or estimated. The position/speed specifying unit
5 specifies the magnetic pole position θe and the
20 rotational speed ωe of the synchronous motor 1.
Specifically, the position/speed specifying unit 5
estimates the magnetic pole position θe and the rotational
speed ωe on the basis of the voltage command 12 output from
the dq-axis current controller 7 and the value of the phase
25 current detected by the current detection unit 4. The
position/speed specifying unit 5 outputs the specified
magnetic pole position θe and the specified rotational
speed ωe.
[0022] A position sensor that detects the magnetic pole
30 position θe may be attached to the synchronous motor 1. A
rotary encoder or resolver is used as the position sensor.
Instead of the position sensor, a speed sensor such as a
tachogenerator may be attached to the synchronous motor 1.
11
Note that the use of the position sensor or the speed
sensor may not be suitable for the synchronous motor 1 due
to restrictions such as use environment and cost. In the
first embodiment, the motor drive apparatus 100 is assumed
to perform position sensorless control. 5 The motor drive
apparatus 100 is not limited to the one in which the
position sensor or the speed sensor is not used, and may be
one in which the position sensor or the speed sensor is
used. Note that the refrigerant compressor described above
10 is a typical example of an application in which the
position sensor or the speed sensor is not readily used.
[0023] Various methods have been proposed regarding the
position sensorless control of the synchronous motor 1, and
the first embodiment may basically use any method. As a
15 known method, for example, a speed estimation method is
available in which a state quantity of the synchronous
motor 1 is estimated by a state observer, and the
rotational speed ωe is adaptively identified using an
estimation error of the state quantity. This method is a
20 method called an adaptive observer, and has an advantage in
that speed estimation robust to a change in an induced
voltage constant can be performed. When the adaptive
observer is not used, the magnetic pole position θe may be
estimated simply from an arctangent of a speed
25 electromotive force. This method is called an arctangent
method. The arctangent method has a disadvantage in that
an error occurs in speed estimation when the induced
voltage constant has an error, but involves simpler
calculations than the adaptive observer. Many other
30 position sensorless control methods have been proposed, and
any method may be used as long as the magnetic pole
position θe and the rotational speed ωe can be estimated.
[0024] The speed controller 6 controls the rotational
12
speed ωe of the synchronous motor 1 by determining a q-axis
current command iq
* on the basis of a speed command ω1
* that
is a first speed command, a speed droop amount Δω, and the
specified rotational speed ωe.
[0025] FIG. 2 is a block diagram illustrating 5 an example
of a configuration of the speed controller included in the
motor drive apparatus according to the first embodiment.
The speed controller 6 includes adders 21 and 25, a
subtractor 22, a speed feedback (FB) controller 23, and a
10 speed feedforward (FF) controller 24.
[0026] The speed command ω1
* is input to the speed
controller 6 from outside the motor drive apparatus 100.
The speed command ω1
* may be obtained by calculation in the
motor drive apparatus 100. The speed command ω1
* and the
15 speed droop amount Δω are input to the adder 21. The adder
21 adds up the speed command ω1
* and the speed droop amount
Δω, and outputs a second speed command ω2
* that is an added
result. The speed droop amount Δω will be described later.
The second speed command ω2
* and the rotational speed ωe are
20 input to the subtractor 22. The subtractor 22 outputs a
difference between the second speed command ω2
* and the
rotational speed ωe. The speed FB controller 23 performs
FB control such that the difference input from the
subtractor 22 equals zero.
25 [0027] As the speed FB controller 23, a proportional
integral (PI) controller is used. It is known in the PI
controller that a steady-state error with respect to a step
response equals zero. The use of the PI controller
facilitates the gain design. As the speed FB controller 23,
30 a controller based on a control rule other than PI control
may be used. In order to achieve zero steady-state error,
a controller having an integrator is used for the speed FB
controller 23. The speed FF controller 24 is connected in
13
parallel to the speed FB controller 23. The second speed
command ω2
* is input to the speed FF controller 24. The
speed FF controller 24 performs FF control on the
rotational speed ωe. With the speed FF controller 24
provided, the speed controller 6 can accelerate 5 a control
response. The adder 25 generates the q-axis current
command iq
* by adding up an output value of the speed FB
controller 23 and an output value of the speed FF
controller 24.
[0028] A d-axis current command id
10 * is determined by the
flux weakening controller 9. The speed controller 6 may
determine the d-axis current command id
* by “maximum torque
per ampere control (MTPA)”. The d-axis current command id
*
will be described later.
15 [0029] The dq-axis current controller 7 as a current
controller controls the phase current flowing through the
synchronous motor 1. As the dq-axis current controller 7,
a vector controller that performs vector control on dq
rotating coordinates is used. A typical vector controller
20 performs current control on the dq rotating coordinates
with respect to the magnetic pole position θe. When the
phase current is converted into a value on the dq rotating
coordinates, an alternating current value is converted into
a direct current value that makes the control easy, whereby
25 the motor drive apparatus 100 performs the current control
on the dq rotating coordinates. Since the coordinate
transform requires information on the magnetic pole
position θe, the magnetic pole position θe specified by the
position/speed specifying unit 5 is input to the dq-axis
30 current controller 7.
[0030] The dq-axis current controller 7 performs the
coordinate transform to convert the value of the phase
current into a value of a d-axis current and a value of a
14
q-axis current that are currents in a dq coordinate system.
The dq-axis current controller 7 also determines the
voltage command 12 on the basis of the d-axis current and
the d-axis current command id
* as well as the q-axis
current and the q-axis current command iq
5 *. The dq-axis
current controller 7 adjusts a d-axis voltage command such
that the d-axis current matches the d-axis current command
id
*. The dq-axis current controller 7 adjusts a q-axis
voltage command such that the q-axis current matches the qaxis
current command iq
10 *. The dq-axis current controller 7
thus determines the voltage command on the dq rotating
coordinates.
[0031] The dq-axis current controller 7 includes a PI
controller (not illustrated) that performs FB control on
15 the d-axis current, a PI controller (not illustrated) that
performs FB control on the q-axis current, and a noninteracting
controller (not illustrated) that performs FF
compensation on an interacting component of the dq axis.
If the d-axis current can properly follow the d-axis
current command id
20 * and the q-axis current can properly
follow the q-axis current command iq
*, a method other than
the method described above may be used as the control
method of the dq-axis current controller 7.
[0032] The dq-axis current controller 7 performs the
25 coordinate transform from the voltage command on the dq
rotating coordinates to a value of three-phase stationary
coordinates on the basis of the magnetic pole position θe.
The dq-axis current controller 7 outputs the voltage
command 12 on the three-phase stationary coordinates to the
30 power converter 3.
[0033] The voltage amplitude calculating unit 8 obtains
a voltage amplitude that is the amplitude of the voltage
command. The amplitude of the voltage command is also
15
referred to as the norm of a voltage command vector or the
absolute value of the voltage command vector. Various
methods can be considered as a method of calculating the
amplitude of the voltage command. The voltage amplitude
calculating unit 8 calculates the amplitude 5 of the voltage
command by, for example, the calculation expressed in the
following expression (1). The voltage amplitude
calculating unit 8 outputs a result of the calculation of
the voltage amplitude.
10 [0034] [Expression 1]
... (1)
[0035] In the expression, |νdq
*| represents the voltage
amplitude, “νd
*” represents the d-axis voltage command, and
“νq
*” represents the q-axis voltage command. When the
15 voltage amplitude calculating unit 8 performs the
calculation of expression (1), the voltage commands νd
* and
νq
* on the dq rotating coordinates are input to the voltage
amplitude calculating unit 8 from the dq-axis current
controller 7.
20 [0036] Note that the voltage amplitude calculating unit
8 may calculate a modulation factor instead of the voltage
amplitude |νdq
*|. The modulation factor is a
standardization of the voltage amplitude |νdq
*| in order to
evaluate how large the voltage amplitude |νdq
*| is with
25 respect to the output limit of the power converter 3. The
voltage amplitude calculating unit 8 calculates the
modulation factor “M” by calculation expressed in the
following expression (2).
[0037] [Expression 2]
30 ... (2)
[0038] In the expression, “VDC” represents a direct
16
current bus voltage of the voltage source inverter as the
power converter 3. The direct current bus voltage is
detected by a direct current bus voltage detector. The
direct current bus voltage detector is not illustrated. A
voltage region in which the modulation 5 factor obtained by
expression (2) is smaller than one is called an inverter
linear region. A voltage region in which the modulation
factor obtained by expression (2) is larger than one is
called an overmodulation region or a voltage saturation
10 region.
[0039] The flux weakening controller 9 performs flux
control for controlling the amplitude of the voltage output
to the synchronous motor 1 by determining the d-axis
current command id
* on the basis of the voltage amplitude
|νdq
15 *| and a first voltage limit value Vlim1. The speed
droop controller 10 controls the speed droop amount Δω on
the basis of the voltage amplitude |νdq
*| and a second
voltage limit value Vlim2. Here, the details of the flux
weakening controller 9 and the speed droop controller 10
20 will be described.
[0040] FIG. 3 is a diagram for explaining a voltage
vector representing a voltage state of the synchronous
motor to be controlled by the motor drive apparatus
according to the first embodiment. FIG. 3 illustrates the
25 voltage vector when the synchronous motor 1 as an embedded
permanent magnet synchronous motor rotates in the highspeed
region. In the high-speed region, a voltage drop due
to coil resistance of the synchronous motor 1 is often
negligible, so that the voltage drop due to the coil
30 resistance is omitted in FIG. 3. FIG. 3 illustrates the
voltage vector in a steady state and omits a transient term.
[0041] In the synchronous motor 1, as the rotational
speed ωe increases, a speed electromotive force ωeΦa
17
increases. Here, “Φa” represents a dq-axis flux linkage
and is a value unique to the motor. The speed
electromotive force ωeΦa is generated in the direction of
the q-axis. In the permanent magnet synchronous motor, the
q-axis current and magnet torque 5 of the motor are
proportional to each other. The synchronous motor 1
normally outputs torque to cause the mechanical device 2 to
perform some mechanical work. The q-axis current iq flows
through the synchronous motor 1, and a voltage ωeLqiq is
10 generated in the direction of the d-axis by armature
reaction of the q-axis current iq. Here, “Lq” represents a
q-axis inductance.
[0042] On the other hand, the d-axis current id
contributes to a small extent to the torque, and thus is
15 controlled to a smaller value in a low-middle speed region,
in which the rotational speed is slower than that in the
high-speed region, than in the high-speed region. As a
known method of determining the d-axis current command id
*
in the low-middle speed region, a method such as id=0
20 control or MTPA is available.
[0043] Generally, there is a limit to the maximum
alternating current voltage that the power converter 3 can
output to the synchronous motor 1. In the high-speed
region, a vector sum of the speed electromotive force ωeΦa
25 and the voltage ωeLqiq may exceed the maximum output voltage
of the power converter 3, and a method called flux
weakening control needs to be used.
[0044] When the dq-axis voltage has a limit value of
“Vom”, the limit value Vom satisfies a relationship of the
30 following expression (3), which is an approximate equation,
in the high-speed region. Note that strictly speaking, the
output limit range of the power converter 3 has a hexagonal
shape, but is approximated to a circle here. Although the
18
discussion in the first embodiment assumes the
approximation to a circle, it is needless to say that the
discussion may be made by assuming exactly a hexagon.
[0045] [Expression 3]
5 ... (3)
[0046] In the first embodiment, a circle whose radius
centered on the origin is the limit value Vom is referred
to as a voltage limit circle 30. Note that the limit value
Vom is known to vary depending on the value of the direct
10 current bus voltage in a case where the power converter 3
is a pulse width modulation (PWM) inverter.
[0047] The speed electromotive force ωeΦa is very large
in the high-speed region; therefore, in order to increase
the q-axis current iq, it is necessary to pass the d-axis
15 current id in a negative direction and to keep the
amplitude of a voltage command vector ν* within the range
of the voltage limit circle 30. As described above, the
method of control that reduces the voltage amplitude by
generating a d-axis stator flux Ldid in the direction
20 opposite to the dq-axis flux linkage Φa is generally called
flux weakening control. Here, “Ld” represents a d-axis
inductance.
[0048] The simplest method of flux weakening control is
a method of determining the d-axis current command id
* on
25 the basis of a voltage equation. By solving the above
expression (3) for the d-axis current id, the following
expression (4) can be obtained.
[0049] [Expression 4]
... (4)
30 [0050] However, the flux weakening control that obtains
19
the d-axis current id on the basis of the above expression
(4) has a disadvantage in that it is sensitive to a change,
variation, or the like of a motor constant, and is not used
often in the industry.
[0051] Integral flux weakening control 5 is known as one
method used instead of the flux weakening control based on
the above expression (4). For example, a method is known
in which the d-axis current command id
* is determined by
performing integral control on a difference between the
voltage amplitude |νdq
10 *| and the first voltage limit value
Vlim1. In the following description, such a method may be
referred to as “d-axis current command-manipulating flux
weakening control”.
[0052] FIG. 4 is a block diagram illustrating an example
15 of a configuration of the flux weakening controller
included in the motor drive apparatus according to the
first embodiment. The flux weakening controller 9 includes
a subtractor 41 and an integrator 42 with a limiter. The
subtractor 41 outputs a difference obtained by subtracting
the voltage amplitude |νdq
20 *| from the first voltage limit
value Vlim1. The integrator 42 obtains the d-axis current
command id
* by integrating a result of multiplying the
difference by a control gain (not illustrated). Since the
flux weakening controller 9 is a controller that integrates
25 the difference between the first voltage limit value Vlim1
and the voltage amplitude |νdq
*|, the motor drive apparatus
100 can automatically adjust the d-axis current command id
*
to an appropriate value that is neither too large nor too
small.
[0053] In a case where the voltage amplitude |νdq
30 *| is
larger than the first voltage limit value Vlim1, the
difference therebetween is negative, so that the d-axis
current command id
* changes in the negative direction. On
20
the contrary, in a case where the voltage amplitude |νdq
*|
is smaller than the first voltage limit value Vlim1, the
difference therebetween is positive, so that the d-axis
current command id
* changes in the positive direction. In
general, a limiter is appropriately provided 5 for the d-axis
current command id
*. The provision of the limiter prevents
the divergence of the integral operation in the integrator
42. The provision of the limiter also prevents the
demagnetization of the synchronous motor 1 due to the daxis
current command id
10 * being excessive. Moreover, a
limiter in the positive direction may be provided in order
to prevent the passage of the positive d-axis current id
when the synchronous motor 1 rotates in the low-middle
speed region. The limit value in the positive direction is
15 usually set to zero or a “current command value by maximum
torque per ampere control”.
[0054] In order to explain the usefulness of the motor
drive apparatus 100 according to the first embodiment,
another method widely known as a flux weakening control
20 method will be described. The “positional error command
calculation”, which is the method described in Patent
Literature 1 above, is considered to be a kind of integral
flux weakening control. According to the method of flux
weakening control described above, the phase angle of the
25 voltage command advances as a result of manipulating the daxis
current command, but a similar effect can be obtained
when the phase of the voltage command is directly
manipulated. The method of directly manipulating the phase
of the voltage command is referred to as “voltage phase
30 control” or the like. It is presumed that the “voltage
phase control” is also used in the “positional error
command calculation”. Another method is known in which,
instead of the phase of the voltage command, the phase of
21
control coordinates is shifted in the advancing direction
with respect to the magnetic pole position. In the
following description, these methods using phase
manipulation may be referred to as “phase-manipulating flux
weakening control”. Every phase-5 manipulating flux
weakening control has a disadvantage in that the
mathematical perspective is poor and the calculation for
determining the control gain is complicated.
[0055] In general, the poor mathematical perspective
10 greatly affects the difficulty of control adjustment. The
classical control engineering approach is powerful means
for the gain design, but does not work when a plant model
or controller includes non-linear elements. A
trigonometric function is required for a phase rotating
15 manipulation, but many differential equations including a
trigonometric function are non-linear elements. The
trigonometric function can be linearly approximated if the
amount of phase manipulation is small, but the amount of
phase manipulation in flux weakening control changes
20 greatly in the range of zero to 90 degrees, so that it is
difficult to perform the linear approximation. It is
generally recognized that the discussion of non-linear
control is difficult, and the control adjustment is not
easy. When an appropriate gain cannot be theoretically
25 found, trial and error experiments are to be repeated to
adjust the control gain, which requires a great deal of
effort. In that respect, it can be said that the phasemanipulating
flux weakening control is an unfavorable
method.
30 [0056] In the motor drive apparatus 100 according to the
first embodiment, the “d-axis current command-manipulating
flux weakening control” enables the gain design to be
performed easily as compared to the “phase-manipulating
22
flux weakening control”. The gain design in the “d-axis
current command-manipulating flux weakening control” will
be described later.
[0057] FIG. 5 is a block diagram illustrating an example
of a configuration of the speed droop controller 5 included
in the motor drive apparatus according to the first
embodiment. Here, a description will be made of a
configuration assumed to be applied to an application that
performs only power running operation in forward rotation
10 such as the refrigerant compressor. It is of course
possible that the speed droop controller 10 has a
configuration in consideration of reverse rotation or
regenerative operation.
[0058] The speed droop controller 10 includes a
15 subtractor 51 and an integrator 52 with a limiter. The
subtractor 51 outputs a difference obtained by subtracting
the voltage amplitude |νdq
*| from the second voltage limit
value Vlim2. The integrator 52 obtains the speed droop
amount Δω by integrating a result of multiplying the
20 difference by a control gain (not illustrated). Since the
speed droop controller 10 is a controller that integrates
the difference between the second voltage limit value Vlim2
and the voltage amplitude |νdq
*|, the motor drive apparatus
100 can automatically adjust the speed droop amount Δω to
25 an appropriate value that is neither too large nor too
small.
[0059] In a case where the voltage amplitude |νdq
*| is
larger than the second voltage limit value Vlim2, the
difference therebetween is negative, so that the speed
30 droop amount Δω changes in the negative direction. On the
contrary, in a case where the voltage amplitude |νdq
*| is
smaller than the second voltage limit value Vlim2, the
difference therebetween is positive, so that the speed
23
droop amount Δω changes in the positive direction. The
integrator 52 limits the range that the speed droop amount
Δω can take by a limiter such that the integration
operation does not diverge. By setting an upper limit
value of the speed droop amount Δω to zero, 5 the motor drive
apparatus 100 can prevent the synchronous motor 1 from
decelerating under a condition that voltage saturation does
not occur. That is, the speed droop controller 10 adjusts
the speed droop amount Δω such that the voltage amplitude
|νdq
10 *| does not exceed the second voltage limit value Vlim2.
The speed droop controller 10 thus determines the speed
droop amount Δω that causes the voltage amplitude |νdq
*| to
be smaller than the second voltage limit value Vlim2.
[0060] An appropriate value need only be set as a lower
15 limit value of the speed droop amount Δω. The description
here assumes the case where voltage saturation occurs in
the high-speed region, so that it is sufficient in many
cases if, for example, the lower limit value of the speed
droop amount Δω is set to a value that is about -10% to -
20 20% of the maximum speed ωMax of the synchronous motor 1.
As described above, in the power running operation in the
forward rotation, the range that the speed droop amount Δω
can take is 0≥Δω≥-0.2ωMax.
[0061] On the basis of the speed droop amount Δω thus
25 obtained, the speed controller 6 reduces the speed command
ω1
* and determines the second speed command ω2
*. In a case
where serious voltage saturation occurs such as when a load
torque larger than the maximum torque that the synchronous
motor 1 can output is applied to the synchronous motor 1,
30 the motor drive apparatus 100 eases the voltage saturation
by reducing the speed command ω1
*. By configuring the flux
weakening controller 9 and the speed droop controller 10 as
described above, the gain design of the flux weakening
24
controller 9 and the speed droop controller 10 can be
performed very easily.
[0062] Next, the design of the gain in the motor drive
apparatus 100 will be described with reference to FIGS. 6
to 15. FIG. 6 is a diagram illustrating 5 an example of a
control model of the motor drive apparatus and the
synchronous motor according to the first embodiment. FIG.
6 illustrates details of a controller model of the motor
drive apparatus 100 and an electric plant model of the
10 synchronous motor 1. Here, the control design for
specifically determining a flux weakening control gain KIfw
of the flux weakening controller 9 and a speed droop gain
KIst of the speed droop controller 10 will be described.
[0063] FIG. 7 is a diagram illustrating a model that
15 approximates the control model illustrated in FIG. 6 in the
vicinity of an operating point in the high-speed region.
In a case where the control response of the dq-axis current
controller 7 is determined to be sufficiently high as
compared to the control response of the speed controller 6,
20 the flux weakening controller 9, and the speed droop
controller 10, it can be considered that the d-axis current
command id
* substantially matches the d-axis current id and
that the q-axis current command iq
* substantially matches
the q-axis current iq. It is also assumed that the
25 rotational speed ωe changes gradually near the operating
point. Furthermore, it is assumed that the rotational
speed ωe is sufficiently high, and a voltage drop due to
armature resistance R is very small and negligible. Under
these conditions, the control model illustrated in FIG. 6
30 can be simplified and expressed as in FIG. 7.
[0064] Here, the design of the flux weakening control
gain KIfw, which is the control gain of the flux weakening
controller, will be described. FIG. 8 is a first diagram
25
for explaining the design of the flux weakening control
gain of the flux weakening controller illustrated in FIG. 4.
FIG. 9 is a second diagram for explaining the design of the
flux weakening control gain of the flux weakening
controller illustrated in FIG. 4. FIG. 5 10 is a third
diagram for explaining the design of the flux weakening
control gain of the flux weakening controller illustrated
in FIG. 4. FIG. 11 is a fourth diagram for explaining the
design of the flux weakening control gain of the flux
10 weakening controller illustrated in FIG. 4.
[0065] The block diagram illustrated in FIG. 8 can be
obtained by omitting the speed droop controller 10 and the
speed controller 6 from the model illustrated in FIG. 7.
Here, a transfer function for obtaining the voltage
amplitude |νdq
15 *| on the basis of the first voltage limit
value Vlim1 will be considered. Since the transfer function
is a function expressed as single input and single output,
input elements other than the first voltage limit value
Vlim1 are considered to be constant near the operating point.
That is, the q-axis current command iq
20 * and the dq-axis
flux linkage Φa are ignored. Under such conditions, the
block diagram illustrated in FIG. 9 can be obtained from
the block diagram illustrated in FIG. 8.
[0066] The block diagram illustrated in FIG. 10
25 represents a normative model of the flux weakening
controller 9 based on the block diagram illustrated in FIG.
9. It is desirable that the flux weakening controller 9 be
designed such that the voltage amplitude |νdq
*|
appropriately follows a change in the first voltage limit
30 value Vlim1. It is also desirable that the speed until the
response converges be specified using a freely selected
time constant. Therefore, the normative model of the flux
weakening controller 9 should be a first-order low-pass
26
filter 60. The low-pass filter 60 has a cutoff angular
frequency ωfw. The cutoff angular frequency is a reciprocal
of the time constant.
[0067] It is clear that the low-pass filter 60
illustrated in FIG. 10 is equivalent to 5 the configuration
illustrated in FIG. 11 by a simple modification. The lowpass
filter 60 illustrated in FIG. 11 includes a subtractor
61 and an integrator 62. The flux weakening controller 9
can obtain desired response characteristics by designing
10 the flux weakening control gain KIfw such that the open-loop
transfer function in the block diagram illustrated in FIG.
9 matches the open-loop transfer function in the block
diagram illustrated in FIG. 11. Therefore, the flux
weakening control gain KIfw is determined by the following
15 expression (5).
[0068] [Expression 5]
... (5)
[0069] Next, the design of the speed droop control gain,
which is the control gain of the speed droop controller 10,
20 will be described. FIG. 12 is a first diagram illustrating
the design of the speed droop control gain of the speed
droop controller illustrated in FIG. 5. FIG. 13 is a
second diagram illustrating the design of the speed droop
control gain of the speed droop controller illustrated in
25 FIG. 5. FIG. 14 is a third diagram illustrating the design
of the speed droop control gain of the speed droop
controller illustrated in FIG. 5. FIG. 15 is a fourth
diagram illustrating the design of the speed droop control
gain of the speed droop controller illustrated in FIG. 5.
30 The speed droop control gain is determined on the basis of
a transfer function of the speed controller 6 and a
transfer function of the synchronous motor 1.
27
[0070] The block diagram illustrated in FIG. 12 can be
obtained by omitting the flux weakening controller 9 from
the model illustrated in FIG. 7. Here, a transfer function
for obtaining the voltage amplitude |νdq
*| on the basis of
the second voltage limit value Vlim2 will 5 be considered.
Since the transfer function is a function expressed as
single input and single output, input elements other than
the second voltage limit value Vlim2 are considered to be
constant near the operating point. That is, the d-axis
current command id
10 * and the dq-axis flux linkage Φa are
ignored. Under such conditions, the block diagram
illustrated in FIG. 13 can be obtained from the block
diagram illustrated in FIG. 12.
[0071] Moreover, the block diagram illustrated in FIG.
15 14 can be obtained by modifying the block diagram
illustrated in FIG. 13. The block diagram illustrated in
FIG. 14 includes a transfer function of the speed FB
controller 23. Here, the gain design of the speed FB
controller 23 will be described first.
20 [0072] As a method of designing a proportional gain KPS
of the speed FB controller 23, for example, a method using
the following expression (6) is known. As a method of
designing an integral gain KIS of the speed FB controller
23, for example, a method using the following expression
25 (7) is known.
[0073] [Expression 6]
... (6)
[Expression 7]
... (7)
30 [0074] In the expressions, “J” represents inertia, “Pm”
represents a pole logarithm, “ωSC” represents a speed
control band, and “ωPI” represents a PI breakpoint angular
28
frequency. When “ωPI” and “ωSC” are determined with a
policy that a target value response is determined on the
proportional control side and the integral control is
operated only to make the steady-state error zero, “ωPI” is
preferably set to one fifth 5 or less of “ωSC”.
[0075] An open-loop transfer function GO (s) in the
block diagram illustrated in FIG. 14 is expressed by the
following expression (8). Therefore, the block diagram
illustrated in FIG. 14 can be modified as the block diagram
10 illustrated in FIG. 15.
[0076] [Expression 8]
... (8)
[0077] A closed-loop transfer function GC (s) in the
block diagram illustrated in FIG. 15 is expressed by the
15 following expression (9). In expression (9), the degree of
a complex number “s” of a transfer function parameter is
two.
[0078] [Expression 9]
... (9)
20 [0079] A general expression for a transfer function of a
second-order lag system is expressed by the following
expression (10). In the expression, “ζ” represents a
damping coefficient, and “ωn” represents a natural angular
frequency.
25 [0080] [Expression 10]
... (10)
[0081] By comparing the coefficient of the denominator
in expression (9) and the coefficient of the denominator in
expression (10), the speed droop gain KIst that allows the
30 natural angular velocity ωn of the speed droop controller
29
10 to take a freely selected value can be determined by the
following expression (11).
[0082] [Expression 11]
... (11)
[0083] Note that the damping coefficient 5 ζ of the speed
droop controller 10 is expressed by the following
expression (12).
[0084] [Expression 12]
... (12)
10 [0085] In a case where the damping coefficient ζ is not
appropriate, the speed droop control by the speed droop
controller 10 becomes unstable. When the damping
coefficient ζ is less than 0.5, the fluctuation of the
speed droop amount Δω until the speed droop amount Δω
15 converges is noticeable. It is thus desirable that the
damping coefficient ζ be at least 0.5 or more. Note that
since it is clear that ωn>0 and ωPI>0, the damping
coefficient ζ is always a positive value. Therefore, it
can be said that the transfer function expressed in the
20 above expression (9) is stable.
[0086] It is difficult to perform the gain design in the
case of the control configuration that performs a flag
determination as to whether or not the voltage is saturated
and performs processing of dropping the speed command as in
25 Patent Literature 1 described above. On the other hand,
according to the first embodiment, the clear gain design as
described above is possible by configuring the control
system as illustrated in FIG. 1.
[0087] FIG. 16 is a graph illustrating an example of
30 operating waveforms when the motor drive apparatus
according to the first embodiment is used. FIG. 16
30
graphically illustrates an example of the relationship
between each of the rotational speed ωe, a load torque T,
the d-axis current command id
*, the voltage amplitude |νdq
*|,
and the speed droop amount Δω and time.
[0088] It is assumed that the load torque 5 T is gradually
increased from time t1 to time t5 as illustrated in FIG. 16
when the synchronous motor 1 rotates at a constant speed.
In the period up to the time t2, the voltage amplitude
|νdq
*| is smaller than the first voltage limit value Vlim1,
and thus the d-axis current command id
10 * is zero. At time t2,
the voltage amplitude |νdq
*| exceeds the first voltage limit
value Vlim1. Then, the integral flux weakening controller 9
increases the d-axis current command id
* in the negative
direction so that the voltage amplitude |νdq
*| does not
15 increase any more.
[0089] It is assumed that the d-axis current command id
*
is increased in the negative direction to reach a lower
limit value IdLimL at time t3. The lower limit value IdLimL
is set to protect the synchronous motor 1 from
20 demagnetization, heat generation, and the like. The d-axis
current id exceeding the lower limit value IdLimL cannot be
passed through the synchronous motor 1. Therefore, after
time t3, the speed command ω1
* needs to be lowered in order
to ease the voltage saturation.
25 [0090] In FIG. 16, the second voltage limit value Vlim2
is set to a value higher than the first voltage limit value
Vlim1. The voltage amplitude |νdq
*| increases during the
period from time t3 to time t4 and reaches the second
voltage limit value Vlim2 at time t4, and the speed droop
30 amount Δω starts to be generated. During the period from
time t4 to time t5, the rotational speed ωe decreases due
to the generation of the speed droop amount Δω, and the
voltage amplitude |νdq
*| stops increasing. After time t5,
31
the load torque T is constant so that the rotational speed
ωe stops decreasing.
[0091] In the first embodiment, the first voltage limit
value Vlim1 and the second voltage limit value Vlim2 are set
separately and the second voltage limit 5 value Vlim2 is
higher than the first voltage limit value Vlim1, whereby the
motor drive apparatus 100 shifts the operation timing of
the flux weakening control and the operation timing of the
speed droop control from each other. As a result, the
10 motor drive apparatus 100 can increase the output torque of
the synchronous motor 1 by making the best use of the flux
weakening control.
[0092] Note that in the case of increasing the maximum
torque and reducing copper loss by utilizing an
15 overmodulation region of the power converter 3, the first
voltage limit value Vlim1 and the second voltage limit value
Vlim2 can be set within a range expressed by the following
expression (13). As a result, the motor drive apparatus
100 can use the output limit range of the synchronous motor
20 1 to the fullest by controlling the speed droop amount Δω
after the limitation of the voltage amplitude by the flux
weakening control is no longer effective.
[0093] [Expression 13]
... (13)
25 [0094] According to the first embodiment, the motor
drive apparatus 100 can prevent a phenomenon in which the
control of the synchronous motor 1 becomes unstable at the
time of voltage saturation without performing complicated
work for the control adjustment. In applications to a
30 refrigerant compressor or the like, labor saving in the
control adjustment is a great advantage. Furthermore, the
motor drive apparatus 100 can increase the maximum torque
32
and reduce the copper loss by utilizing the overmodulation
region of the power converter 3. As described above, the
motor drive apparatus 100 has the effect that the workload
required for the adjustment to perform stable control on
the motor 5 can be reduced.
[0095] Second Embodiment.
In a second embodiment, a hardware configuration of
the motor drive apparatus 100 will be described. FIG. 17
is a diagram illustrating an example of the hardware
10 configuration of the motor drive apparatus according to the
second embodiment of the present invention. In the second
embodiment, components identical to those in the first
embodiment are denoted by the same reference numerals as
those assigned to the corresponding components in the first
15 embodiment. FIG. 17 illustrates the synchronous motor 1,
the power converter 3, and the current detection unit 4
that are included in the motor system and the mechanical
device 2 that operates with the synchronous motor 1 as the
power source, together with the motor drive apparatus 100.
20 [0096] The motor drive apparatus 100 includes a
processor 101 and a memory 102 as the hardware
configuration. The functions of the position/speed
specifying unit 5, the speed controller 6, the dq-axis
current controller 7, the voltage amplitude calculating
25 unit 8, the flux weakening controller 9, and the speed
droop controller 10 illustrated in FIG. 1 are implemented
by the processor 101 executing a program stored in the
memory 102.
[0097] The processor 101 is a central processing unit
30 (CPU), a processing unit, an arithmetic unit, a
microprocessor, a microcomputer, or a digital signal
processor (DSP). The memory 102 includes a volatile
storage device such as a random access memory and a
33
nonvolatile auxiliary storage device such as a flash memory.
The memory 102 may include an auxiliary storage device such
as a hard disk instead of the nonvolatile auxiliary storage
device. The illustration of the volatile storage device
and the auxiliary storage device is omitted. 5 The processor
101 reads the program stored in the auxiliary storage
device via the volatile storage device. The processor 101
outputs data such as a calculation result to the volatile
storage device. The processor 101 may save the data in the
10 auxiliary storage device via the volatile storage device.
[0098] Various modes have been studied for the power
converter 3 and the current detection unit 4, but basically
any mode may be used therefor. The motor system may be
provided with voltage detecting means that detects an input
15 voltage or output voltage of the power converter 3 or
voltage detecting means that detects a direct current bus
voltage.
[0099] Basically, any method may be used as a method of
transmitting and receiving data between the components.
20 Each component may transmit and receive a digital signal,
or may transmit and receive an analog signal. The digital
signal may be communicated by parallel communication or
serial communication. The analog signal and the digital
signal may be converted as appropriate by a converter (not
25 illustrated). For example, in a case where the phase
current detected by the current detection unit 4 is
expressed by an analog signal, the analog signal is
converted into a digital signal by a digital to analog
(D/A) converter (not illustrated), and data is transmitted
30 to the processor 101. The D/A converter (not illustrated)
may be provided inside the motor drive apparatus 100 or
inside the current detection unit 4.
[0100] The signal of the voltage command transmitted
34
from the processor 101 to the power converter 3 may be
either an analog signal or a digital signal. Moreover, the
processor 101 may include a modulation unit such as a
carrier comparison modulation unit or a space vector
modulation unit. The processor 101 5 may transmit the
voltage command, which is a pulse train obtained after
modulation, to the power converter 3. In a case where the
voltage detecting means that detects the input voltage or
the output voltage of the power converter 3 or the voltage
10 detecting means that detects the direct current bus voltage
is provided, basically, any method may be used as a method
of transmission and reception between the voltage detecting
means and the motor drive apparatus 100. In a case where
the position sensor is attached to the synchronous motor 1,
15 basically, any method may be used as a method of
transmission and reception between the position sensor and
the motor drive apparatus 100.
[0101] The processor 101 determines the voltage command
12 by performing speed control calculation and current
control calculation on the basis of the speed command ω1
20 *.
When the amplitude of the voltage command 12 exceeds the
first voltage limit value Vlim1, the flux weakening control
operates, and when the amplitude of the voltage command 12
exceeds the second voltage limit value Vlim2, the speed
25 droop control operates.
[0102] The speed command ω1
*, the first voltage limit
value Vlim1, and the second voltage limit value Vlim2 are
given to the motor drive apparatus 100 from a computer
outside the motor drive apparatus 100. The illustration of
the computer that gives the speed command ω1
30 *, the first
voltage limit value Vlim1, and the second voltage limit
value Vlim2 to the motor drive apparatus 100 is omitted.
The speed command ω1
*, the first voltage limit value Vlim1,
and the second voltage limit value Vlim2 may be calculated
inside the processor 101. Depending on the computing power
of the processor 101, the processor 101 may perform
calculation processing other than the calculation of the
speed command ω1
*, the first voltage limit 5 value Vlim1, and
the second voltage limit value Vlim2.
[0103] The configuration illustrated in the above
embodiment merely illustrates an example of the content of
the present invention, and can thus be combined with
10 another known technique or partially omitted and/or
modified without departing from the scope of the present
invention.
Reference Signs List
15 [0104] 1 synchronous motor; 2 mechanical device; 3
power converter; 4 current detection unit; 5
position/speed specifying unit; 6 speed controller; 7 dqaxis
current controller; 8 voltage amplitude calculating
unit; 9 flux weakening controller; 10 speed droop
20 controller; 11 signal; 12 voltage command; 21, 25 adder;
22, 41, 51, 61 subtractor; 23 speed FB controller; 24
speed FF controller; 30 voltage limit circle; 42, 52, 62
integrator; 60 low-pass filter; 100 motor drive
apparatus; 101 processor; 102 memory.
WE CLAIM:
1. A motor drive apparatus comprising:
a current controller to convert a value of a phase
current flowing through a motor into values of a d-axis
current and a q-axis current that are 5 currents in a dq
coordinate system, and control the phase current by
determining a voltage command on the basis of the d-axis
current and a d-axis current command as well as the q-axis
current and a q-axis current command;
10 a voltage amplitude calculating unit to obtain a
voltage amplitude that is an amplitude of the voltage
command;
a speed controller to control a rotational speed of
the motor by determining the q-axis current command on the
15 basis of a speed command, the rotational speed, and a speed
droop amount that reduces the speed command;
a flux weakening controller to perform flux control to
limit an amplitude of a voltage output to the motor, by
determining the d-axis current command on the basis of the
20 voltage amplitude and a first voltage limit value; and
a speed droop controller to control the speed droop
amount on the basis of the voltage amplitude and a second
voltage limit value, wherein
the speed droop controller determines the speed droop
25 amount that makes the voltage amplitude smaller than the
second voltage limit value.
2. The motor drive apparatus according to claim 1,
wherein the second voltage limit value is a value larger
30 than the first voltage limit value.
3. The motor drive apparatus according to claim 1 or 2,
wherein the speed droop controller is a controller to
integrate a difference between the second voltage limit
value and the voltage amplitude.
4. The motor drive apparatus according to any one of
claims 1 to 3, wherein the flux weakening 5 controller is a
controller to integrate a difference between the first
voltage limit value and the voltage amplitude.
5. The motor drive apparatus according to any one of
claims 1 to 4, wherein a control gain of the speed droop
controller is determined on the basis of a transfer
function of the speed controller and a transfer function of
the motor.