Method of Controlling of a Brushless Permanent-Magnet Motor
The present invention relates to a method of controlling a brushless permanent-magnet
motor.
Efforts are continually being made to improve the efficiency of brushless permanentmagnet
motors.
The present invention provides a method of controlling a brushless permanent-magnet
motor, the method comprising: dividing each half of an electrical cycle into a
conduction period followed by a primary freewheel period; dividing the conduction
period into a first excitation period, a secondary freewheel period, and a second
excitation period; exciting a winding of the motor during each excitation period; and
freewheeling the winding during each freewheel period, wherein the secondary
freewheel period has a position and length within the conduction period that reduces the
harmonic content of current in the winding relative to back EMF in the winding.
For a permanent-magnet motor, the torque-to-current ratio during excitation is at a
maximum when the waveform of the phase current matches that of the back EMF.
Accordingly, by employing a secondary freewheel period that reduces the harmonic
content of the phase current waveform relative to the back EMF waveform, the
efficiency of the motor is improved.
The secondary freewheel period may occur at a time when the back EMF in the winding
is rising, and the primary freewheel period may occur at a time when back EMF is
principally falling. Upon exciting the phase winding, the phase current may rise at a
faster rate than that of the back EMF. As a result, the phase current may lead the back
EMF. The secondary freewheel period serves to check momentarily the rise in the
phase current. Consequently, the phase current more closely follows the rise of the back
EMF during the conduction period. The primary freewheel period makes use of the
inductance of the winding such that torque continues to be generated by the phase
current without any additional power being drawn from the power supply. As the back
EMF falls, less torque is generated for a given phase current. Accordingly, by
freewheeling the winding during the period of falling back EMF, the efficiency of the
motor may be improved without adversely affecting the torque.
The length of the secondary freewheel period may be less than each of the primary
freewheel period, the first excitation period and the second excitation period.
Consequently, the secondary freewheel period acts to check momentarily the rise of the
phase current without adversely affecting the power of the motor.
The method may comprise exciting the winding with a supply voltage, and varying the
length of the conduction period in response to changes in the supply voltage and/or the
speed of the motor. As a result, better control may be achieved over the power of the
motor.
As the supply voltage decreases, less current and thus less power are driven into the
motor over the same conduction period. Equally, as the speed of the motor increases,
the magnitude of the back EMF induced in the winding increases. Less current and thus
less power are then driven into the motor over the same conduction period.
Accordingly, in order to compensate for this, the method may comprise increasing the
conduction period in response to a decrease in the supply voltage and/or an increase in
the speed of the motor.
The first excitation period and the second excitation period may have the same length.
This then has at least two advantages. First, the harmonic content of the phase current
is better balanced over the two excitation periods. As a result, the total harmonic
content of the phase current during the conduction period is likely to be lower than if
the two excitation periods were of different lengths. Second, when implementing the
method in hardware, the hardware need only store a single excitation period, which can
then be used to define both excitation periods. Alternatively, where the method
comprises varying the length of the conduction period in response to changes in the
supply voltage and/or the motor speed, the hardware need only store a single excitation
period for each voltage and/or speed point. As a result, less memory is required to store
the excitation period(s).
The length of the secondary freewheel period may be fixed. This then has the
advantage that, when implementing the method in hardware, the hardware need only
store a single secondary freewheel period. In spite of this advantage, the method may
nevertheless comprise exciting the winding with a supply voltage, and varying the
secondary freewheel period in response to changes in the supply voltage and/or the
speed of the motor. In particular, the method may comprise increasing the length of the
secondary freewheel period in response to an increase in the supply voltage or a
decrease in the speed of the motor. As the supply voltage increases, current in the phase
winding rises at a faster rate during excitation. As a result, the harmonic content of the
phase current waveform relative to the back EMF waveform is likely to increase. By
increasing the length of the secondary freewheel period in response to an increase in the
supply voltage, the rise in the phase current is checked for a longer period and thus the
harmonic content of the phase current waveform may be reduced. As the speed of the
motor decreases, the back EMF rises at a slower rate. Additionally, the magnitude of
the back EMF decreases and thus current in the phase winding rises at a faster rate
during excitation. The back EMF therefore rises at a slower rate whilst the phase
current rises at a faster rate. As a result, the harmonic content of the phase current
waveform relative to the back EMF waveform is likely to increase. By increasing the
secondary freewheel period in response to a decrease in the speed of the motor, the rise
in the phase current is checked for a longer period and thus the harmonic content of the
phase current waveform may be reduced. Accordingly, increasing the secondary
freewheel period in response to an increase in the supply voltage and/or a decrease in
the speed of the motor may result in further improvements in efficiency.
The present invention also provides a method of controlling a brushless permanentmagnet
motor, the method comprising operating in dual-switch mode over a first speed
range and operating in single-switch mode over a second speed range, wherein the
second speed range is higher than the first speed range, each mode comprises dividing
each half of an electrical cycle into a conduction period followed by a primary
freewheel period, the single-switch mode comprises exciting a winding of the motor
during the conduction period and freewheeling the winding during the freewheel period,
and dual-switch mode comprises: dividing the conduction period into a first excitation
period, a secondary freewheel period and a second excitation period; exciting the
winding during each excitation period; and freewheeling the winding during each
freewheel period.
When operating over the first speed range, the length of each electrical half-cycle is
longer and thus the back EMF rises at a slower rate. Additionally, the magnitude of the
back EMF is lower and thus the phase current rises at a faster rate. The back EMF
therefore rises at a slower rate but the phase current rises at a faster rate. As a result, the
phase current may rise at a faster rate than that of the back EMF during excitation. By
introducing the secondary freewheel period, the rise in the phase current is checked
momentarily such that the rise in the phase current more closely follows the rise in the
back EMF. As a result, the harmonic content of the phase current waveform relative to
the back EMF waveform is reduced and thus the efficiency of the motor is increased.
When operating over the second speed range, the length of each electrical half-cycle is
shorter and thus the back EMF rises at a faster rate. Additionally, the magnitude of the
back EMF is higher and thus the phase current rises at a slower rate. The back EMF
therefore rises at a faster rate but the phase current rises at a slower rate. As a result, the
phase current may rise at a rate that is similar to or slower than that of the back EMF
during excitation. A secondary freewheel period would then serve only to increase the
harmonic content of the phase current relative to the back EMF. Accordingly, by
employing dual-switch mode at lower speeds and single-switch mode at higher speeds,
the efficiency of the motor may be improved over both speed ranges.
Each mode may comprise exciting the winding with a supply voltage, and varying the
length of the conduction period in response to changes in the supply voltage or the
speed of the motor. As a result, better control may be achieved over the power of the
motor. For reasons noted above, the method may comprise increasing the conduction
period in response to a decrease in the supply voltage and/or an increase in the speed of
the motor.
The present invention further provides a method of controlling a brushless permanentmagnet
motor, the method comprising operating in multi-switch mode over a first speed
range and operating in dual-switch mode over a second speed range, wherein the second
speed range is higher than the first speed range, multi-switch mode comprises
sequentially exciting and freewheeling a winding of the motor multiple times during
each half of an electrical cycle, the winding being freewheeled when current in the
winding exceeds a predefined limit, and dual-switch mode comprises: dividing each
half of an electrical cycle into a conduction period followed by a primary freewheel
period; dividing the conduction period into a first excitation period, a secondary
freewheel period and a second excitation period; exciting the winding during each
excitation period; and freewheeling the winding during each freewheel period.
When operating over the first speed range, the length of each electrical half-cycle is
relatively long and thus the rate at which the back EMF rises is relatively slow.
Additionally, the magnitude of the back EMF is relatively low and thus the rate at
which the phase current rises is relatively fast. Consequently, when operating over the
first speed range, the phase current rises at a much faster rate than that of the back EMF.
The phase winding is therefore freewheeled whenever the phase current exceeds a
predefined limit. This then protects the hardware used to implement the method from
excessive phase currents. As the speed of the motor increases, the length of each
electrical half-cycle decreases and thus the back EMF rises at faster rate. Additionally,
the magnitude of the back EMF increases and thus the phase current rises at a slower
rate. When operating over the second speed range, the phase current does not exceed
the predefined limit. The phase current nevertheless rises at a faster rate than that of the
back EMF during excitation. By introducing the secondary freewheel period, the rise in
the phase current is checked momentarily such that the rise in the phase current more
closely follows the rise in the back EMF. As a result, the efficiency of the motor is
improved. Accordingly, by employing multi-switch mode at lower speeds and dualswitch
mode at higher speeds, the hardware may be protected from excessive phase
currents at lower speeds whilst the efficiency of the motor may be improved at higher
speeds.
The present invention further provides a control circuit configured to perform a method
as described in any one of the preceding paragraphs, as well as a motor assembly
comprising a brushless permanent-magnet motor and the control circuit.
The control circuit may comprise an inverter for coupling to a winding of the motor, a
gate driver module and a controller. The gate driver module then controls switches of
the inverter in response to control signals received from the controller. The controller is
responsible for dividing each half of an electrical cycle into the conduction period and
the primary freewheel period, and for dividing the conduction period into the first
excitation period, the secondary freewheel period, and the second excitation period.
The controller then generates control signals to excite the winding during each
excitation period and to freewheel the winding during each freewheel period.
In order that the present invention may be more readily understood, an embodiment of
the invention will now be described, by way of example, with reference to the
accompanying drawings, in which:
Figure 1 is a block diagram of a motor assembly in accordance with the present
invention;
Figure 2 is a schematic diagram of the motor assembly;
Figure 3 details the allowed states of the inverter in response to control signals issued by
the controller of the motor assembly;
Figure 4 illustrates various waveforms of the motor assembly when operating in multiswitch
mode;
Figure 5 illustrates various waveforms of the motor assembly when operating in singleswitch
mode; and
Figure 6 illustrates various waveforms of the motor assembly when operating in dualswitch
mode.
The motor assembly 1 of Figures 1 and 2 is powered by a DC power supply 2 and
comprises a brushless motor 3 and a control circuit 4 .
The motor 3 comprises a four-pole permanent-magnet rotor 5 that rotates relative to a
four-pole stator 6 . Conductive wires wound about the stator 6 are coupled together to
form a single phase winding 7 .
The control circuit 4 comprises a filter 8, an inverter 9, a gate driver module 10, a
current sensor 11, a voltage sensor 12, a position sensor 13, and a controller 14.
The filter 8 comprises a link capacitor CI that smoothes the relatively high-frequency
ripple that arises from switching of the inverter 9 .
The inverter 9 comprises a full bridge of four power switches Q1-Q4 that couple the
phase winding 7 to the voltage rails. Each of the switches Q1-Q4 includes a freewheel
diode.
The gate driver module 10 drives the opening and closing of the switches Q1-Q4 in
response to control signals received from the controller 14.
The current sensor 11 comprises a shunt resistor Rl located between the inverter and the
zero-volt rail. The voltage across the current sensor 11 provides a measure of the
current in the phase winding 7 when connected to the power supply 2 . The voltage
across the current sensor 11 is output to the controller 14 as signal, I PHASE.
The voltage sensor 12 comprises a potential divider R2,R3 located between the DC
voltage rail and the zero volt rail. The voltage sensor outputs a signal, V DC, to the
controller 14 that represents a scaled-down measure of the supply voltage provided by
the power supply 2 .
The position sensor 13 comprises a Hall-effect sensor located in a slot opening of the
stator 6 . The sensor 13 outputs a digital signal, HALL, that is logically high or low
depending on the direction of magnetic flux through the sensor 13. The HALL signal
therefore provides a measure of the angular position of the rotor 5 .
The controller 14 comprises a microcontroller having a processor, a memory device,
and a plurality of peripherals (e.g. ADC, comparators, timers etc.). The memory device
stores instructions for execution by the processor, as well as control parameters and
lookup tables that are employed by the processor during operation. The controller 14 is
responsible for controlling the operation of the motor 3 and generates four control
signals S1-S4 for controlling each of the four power switches Q1-Q4. The control
signals are output to the gate driver module 10, which in response drives the opening
and closing of the switches Q1-Q4.
Figure 3 summarises the allowed states of the switches Q1-Q4 in response to the control
signals S1-S4 output by the controller 14. Hereafter, the terms 'set' and 'clear' will be
used to indicate that a signal has been pulled logically high and low respectively. As
can be seen from Figure 3, the controller 14 sets SI and S4, and clears S2 and S3 in
order to excite the phase winding 7 from left to right. Conversely, the controller 14 sets
S2 and S3, and clears SI and S4 in order to excite the phase winding 7 from right to left.
The controller 14 clears SI and S3, and sets S2 and S4 in order to freewheel the phase
winding 7 . Freewheeling enables current in phase the winding 7 to re-circulate around
the low-side loop of the inverter 9 . In the present embodiment, the power switches QlQ4
are capable of conducting in both directions. Accordingly, the controller 14 closes
both low-side switches Q2,Q4 during freewheeling such that current flows through the
switches Q2,Q4 rather than the less efficient diodes. Conceivably, the inverter 9 may
comprise power switches that conduct in a single direction only. In this instance, the
controller 14 would clear SI, S2 and S3, and set S4 so as to freewheel the phase
winding 7 from left to right. The controller 14 would then clear SI, S3 and S4, and set
S2 in order to freewheel the phase winding 7 from right to left. Current in the low-side
loop of the inverter 9 then flows down through the closed low-side switch (e.g. Q4) and
up through the diode of the open low-side switch (e.g. Q2).
The controller 14 operates in one of three modes depending on the speed of the rotor 5 .
At speeds below a first threshold, the controller 14 operates in multi-switch mode. At
speeds above the first threshold but below a second threshold, the controller 14 operates
in dual-switch mode. And at speeds above the third speed threshold, the controller 14
operates in single-switch mode. The speed of the rotor 5 is determined from the interval
between successive edges of the HALL signal, which will hereafter be referred to as the
HALL period.
Multi- switch mode is employed during acceleration of the motor 3 whilst dual-switch
mode and single-switch mode are employed during steady state. A description of each
mode is provided below. Dual-switch mode involves a small but significant change to
single-switch mode. Accordingly, in order that the nature and the significance of the
change may be better appreciated, a description of single-switch mode will be provided
before that of dual-switch mode.
In all three modes the controller 14 commutates the phase winding 7 in response to
edges of the HALL signal. Each HALL edge corresponds to a change in the polarity of
the rotor 5, and thus a change in the polarity of the back EMF induced in the phase
winding 7 . More particularly, each HALL edge corresponds to a zero-crossing in the
back EMF. Commutation involves reversing the direction of current through the phase
winding 7 . Consequently, if current is flowing through the phase winding 7 in a
direction from left to right, commutation involves exiting the winding from right to left.
In order to simplify the present discussion, it will be assumed that the controller 14
commutates the phase winding 7 in synchrony with the HALL edges, i.e. in synchrony
with the zero-crossings in the back EMF. In reality, however, the controller 14 may
advance, synchronise or retard commutation relative to the HALL edges.
Multi- Switch Mode
When operating in multi-switch mode, the controller 14 sequentially excites and
freewheels the phase winding 7 over each half of an electrical cycle. More particularly,
the controller 14 excites the phase winding 7, monitors the current signal, I PHASE,
and freewheels the phase winding 7 when the current in the phase winding 7 exceeds a
predefined limit. Freewheeling then continues for a predefined freewheel period during
which time current in the phase winding 7 falls to a level below the current limit. At the
end of the freewheel period the controller 14 again excites the phase winding 7 . This
process of exciting and freewheeling the phase winding 7 continues over the full length
of the electrical half-cycle. The controller 14 therefore switches from excitation to
freewheeling multiple times during each electrical half-cycle.
Figure 4 illustrates the waveforms of the HALL signal, the back EMF, the phase
current, the phase voltage, and the control signals over a couple of HALL periods when
operating in multi-switch mode.
At relatively low speeds, the magnitude of the back EMF induced in the phase
winding 7 is relatively small. Current in the phase winding 7 therefore rises relatively
quickly during excitation, and falls relatively slowly during freewheeling. Additionally,
the length of each HALL period and thus the length of each electrical half-cycle is
relatively long. Consequently, the frequency at which the controller 14 switches from
excitation to freewheeling is relatively high. However, as the rotor speed increases, the
magnitude of the back EMF increases and thus current rises at a slower rate during
excitation and falls at a quicker rate during freewheeling. Additionally, the length of
each electrical half-cycle decreases. As a result, the frequency of switching decreases.
Single-Switch Mode
When operating in single-switch mode, the controller 14 divides each half of an
electrical cycle into a conduction period followed by a freewheel period. The
controller 14 then excites the phase winding 7 during the conduction period and
freewheels the phase winding 7 during the freewheel period. When operating within
single-switch mode, the phase current does not exceed the current limit during
excitation. Consequently, the controller 14 switches from excitation to freewheeling
only once during each electrical half-cycle.
Figure 5 illustrates the waveforms of the HALL signal, the back EMF, the phase
current, the phase voltage, and the control signals over a couple of HALL periods when
operating in single-switch mode.
The magnitude of the supply voltage used to excite the phase winding 7 may vary. For
example, the power supply 2 may comprise a battery that discharges with use.
Alternatively, the power supply 2 may comprise an AC source, rectifier and smoothing
capacitor that provide a relatively smooth voltage, but the RMS voltage of the AC
source may vary. Changes in the magnitude of the supply voltage will influence the
amount of current that is driven into the phase winding 7 during the conduction period.
As a result, the power of the motor 3 is sensitive to changes in the supply voltage. In
addition to the supply voltage, the power of the motor 3 is sensitive to changes in the
speed of the rotor 5 . As the speed of the rotor 5 varies (e.g. in response to changes in
load), so too does the magnitude of the back EMF. Consequently, the amount of current
driven into the phase winding 7 during the conduction period may vary. The
controller 14 therefore varies the length of the conduction period in response to changes
in the speed of the rotor 5 and/or the magnitude of the supply voltage. As a result, the
controller 14 is better able to control the power of the motor 3 in response to changes in
the rotor speed and/or the supply voltage.
In order to vary the length of the conduction period, the controller 14 stores a lookup
table of different conduction periods for different voltages and/or speeds. The
controller 14 then indexes the lookup table (e.g. in response to each or every nth HALL
edge) using the supply voltage and/or the rotor speed to select a conduction period. The
speed of the rotor 5 is obtained from the length of the HALL period, whilst the supply
voltage is obtained from the V DC signal.
The lookup table stores conduction periods that achieve a particular output power at
each voltage and/or speed point. As the supply voltage decreases, less current and thus
less power are driven into the motor 3 over the same conduction period. Similarly, as
the rotor speed increases, the magnitude of the back EMF increases. Accordingly, less
current and thus less power is driven into the motor 3 over the same conduction period.
Accordingly, in order to compensate for this behaviour, the controller 14 may employ a
conduction period that increases in response to a decrease in the supply voltage or an
increase in the rotor speed.
Dual-Switch Mode
During excitation of the phase winding, the torque-to-current ratio is at a maximum
when the waveform of the phase current matches that of the back EMF. Improvements
in the efficiency of the motor 3 are therefore achieved by shaping the waveform of the
phase current such that it better matches the waveform of the back EMF, i.e. by
reducing the harmonic content of the phase current waveform relative to the back EMF
waveform. The applicant has found that, when operating at lower speeds within singleswitch
mode, an improvement in the efficiency of the motor 3 is achieved by inserting a
relatively small secondary freewheel period into the conduction period.
At lower rotor speeds, the length of the HALL period is longer and thus the back EMF
rises at a slower rate. Additionally, the magnitude of the back EMF is lower and thus,
assuming the supply voltage is unchanged, current in the phase winding rises at a faster
rate. Consequently, at lower speeds, the back EMF rises at a slower rate but the phase
current rises at a faster rate. As a result, the phase current rises at a faster rate than that
of the back EMF during the early part of the conduction period. The applicant has
found that, by introducing a relatively small secondary freewheel period during the
conduction period, the rise in the phase current is checked momentarily such that the
rise in the phase current more closely follows the rise in the back EMF. As a result, the
harmonic content of the phase current waveform relative to the back EMF waveform is
reduced and thus the efficiency of the motor 3 is increased.
When operating in dual-switch mode, the controller 14 divides each half of an electrical
cycle into a conduction period followed by a primary freewheel period. The
controller 14 then divides the conduction period into a first excitation period, followed
by a secondary freewheel period, followed by a second excitation period. The
controller 14 then excites the phase winding 7 during each of the two excitation periods
and freewheels the phase winding 7 during each of the two freewheel periods. As in
single-switch mode, the phase current does not exceed the current limit during
excitation. Accordingly, the controller 14 switches from excitation to freewheeling
twice during each electrical half-cycle.
Figure 6 illustrates the waveforms of the HALL signal, the back EMF, the phase
current, the phase voltage, and the control signals over a couple of HALL periods when
operating in dual-switch mode.
The controller 14 operates in dual-switch mode at lower speeds where conventionally
the controller 14 would operate in single-switch mode. As in single-switch mode, the
controller 14 varies the length of the conduction period in response to changes in the
magnitude of the supply voltage and/or the speed of the rotor 5 . To this end, the
controller 14 stores a lookup table of different excitation periods for different voltages
and/or speeds. The controller 14 then indexes the lookup table using the supply voltage
and/or the rotor speed to select an excitation period. The selected excitation period is
then used to define both the first excitation period and the second excitation period, i.e.
the controller 14 excites the phase winding 7 for the selected excitation period,
freewheels the phase winding 7 for the secondary freewheel period, and excites the
phase winding 7 again for the selected excitation period.
Since the first excitation period and the second excitation period are of the same length,
the secondary freewheel period occurs at the centre of the conduction period. This has
at least two advantages. First, the harmonic content of the phase current is better
balanced over the two excitation periods. As a result, the total harmonic content of the
phase current over the conduction period is likely to be lower than if the two excitation
periods were of different lengths. Second, the lookup table need only store one
excitation period for each voltage and/or speed point. As a result, less memory is
required for the lookup table and thus a cheaper controller may be used. In spite of the
aforementioned advantages, it may be desirable to alter the position of the secondary
freewheel period in response to changes in the supply voltage and/or the rotor speed.
This may be achieved by employing a lookup table that stores a first excitation period
and a second excitation period for each voltage and/or speed point.
The controller 14 employs a secondary freewheel period that is fixed in length. This
then has the advantage of reducing the memory requirements of the controller 14, i.e.
the controller 14 need only store a single secondary freewheel period. Alternatively,
however, the controller 14 might employ a secondary freewheel period that varies in
response to changes in the supply voltage and/or the rotor speed. In particular, the
controller 14 may employ a secondary freewheel period that increases in response to an
increase in the supply voltage or a decrease in the rotor speed. As the supply voltage
increases, current in the phase winding 7 rises at a faster rate during excitation,
assuming that the rotor speed and thus the magnitude of the back EMF are unchanged.
As a result, the harmonic content of the phase current waveform relative to the back
EMF waveform is likely to increase. By increasing the length of the secondary
freewheel period in response to an increase in the supply voltage, the rise in the phase
current is checked for a longer period and thus the harmonic content of the phase
current waveform may be reduced. As the rotor speed decreases, the length of the
HALL period increases and thus the back EMF rises at a slower rate. Additionally, the
magnitude of the back EMF decreases and thus current in the phase winding 7 rises at a
faster rate, assuming that the supply voltage is unchanged. Consequently, as the rotor
speed decreases, the back EMF rises at a slower rate but the phase current rises at a
faster rate. The harmonic content of the phase current waveform relative to the back
EMF waveform is therefore likely to increase. By increasing the secondary freewheel
period in response to a decrease in the rotor speed, the rise in the phase current is
checked for a longer period and thus the harmonic content of the phase current
waveform may be reduced. Accordingly, increasing the secondary freewheel period in
response to an increase in the supply voltage and/or a decrease in the rotor speed may
result in further improvements in efficiency.
The length of the secondary freewheel period is relatively short and is intended only to
check momentarily the rise in the phase current. Accordingly, the secondary freewheel
period is shorter than both the primary freewheel period and each of the excitation
periods. The actual length of the secondary freewheel period will depend upon the
particular characteristics of the motor assembly 1, e.g. the inductance of the phase
winding 7, the magnitude of the supply voltage, the magnitude of the back EMF etc.
Irrespective of the length, the secondary freewheel period occurs during a period of
rising back EMF in the phase winding 7 . This is contrast to the primary freewheel
period, which occurs principally if not wholly during a period of falling back EMF. The
primary freewheel period makes use of the inductance of the phase winding 7 such that
torque continues to be generated by the phase current without any additional power
being drawn from the power supply 2 . As the back EMF falls, less torque is generated
for a given phase current. Accordingly, by freewheeling the phase winding 7 during the
period of falling back EMF, the efficiency of the motor 3 may be improved without
adversely affecting the torque.
In the embodiment described above, the controller 14 operates in two different modes
during steady state. Dual-switch mode is employed when operating over a first speed
range, and single-switch mode is employed when operating over a second, higher speed
range. This then improves the efficiency of the motor 3 over the full range of speeds
within steady state. If the secondary freewheel period were employed when operating at
the higher speed range then the efficiency of the motor 3 would worsen, at least in the
present embodiment. This is because, when operating at the higher speed range, the
phase current generally rises at a rate that is similar to or slower than that of the back
EMF. Accordingly, the introduction of a secondary freewheel period would only serve
to increase the harmonic content of the phase current relative to the back EMF.
In spite of the comments made in the previous paragraph, dual-switch mode could
conceivably be used at higher speeds as well as lower speeds. For example, the
inductance of the phase winding 7 may be relatively low such that, even when operating
at relatively high speeds, the time constant (L/R) associated with the phase inductance is
particularly short in comparison to the length of the HALL period. As a result, the
phase current rises relatively quickly in comparison to the back EMF. Alternatively,
perhaps the magnitude of the supply voltage relative to the back EMF at higher speeds
is relatively high such that the phase current rises relatively quickly in comparison to
the back EMF. In both these instances, improvements in efficiency may be achieved by
employing dual-switch mode at higher speeds as well as at lower speeds.
In the embodiment described above, the controller 14 stores a lookup table of
conduction periods for use in single-switch mode and excitation periods for use in dualswitch
mode. The primary freewheel period can then be calculated by subtracting the
conduction period from the HALL period. Alternatively, if the phase winding 7 is
commutated in synchrony with each HALL edge, primary freewheeling may continue
indefinitely until the next HALL edge is sensed by the controller 14. Although the
controller 14 stores a lookup table of conduction periods and excitation periods, it will
be appreciated that the same level of control may be achieved by different means. For
example, rather than storing a lookup table of conduction periods and excitation
periods, the controller 14 could store a lookup table of primary freewheel periods,
which is likewise indexed using the magnitude of the supply voltage and/or the speed of
the rotor 5 . The conduction period would then be obtained by subtracting the primary
freewheel period from the HALL period, and each excitation period would be obtained
by subtracting the primary and the secondary freewheel periods from the HALL period
and dividing the result by two:
T CD = T HALL - T FW l
T EXC = (T HALL - T FW l - T_FW_2)/2
where T CD is the conduction period, T EXC is each of the first and second excitation
periods, T HALL is the HALL period, T FW l is the primary freewheel period, and
T FW 2 is the secondary freewheel period.

CLAIMS
1. A method of controlling a brushless permanent-magnet motor, the method
comprising: dividing each half of an electrical cycle into a conduction period followed
by a primary freewheel period; dividing the conduction period into a first excitation
period, a secondary freewheel period, and a second excitation period; exciting a winding
of the motor during each excitation period; and freewheeling the winding during each
freewheel period, wherein the secondary freewheel period has a position and length
within the conduction period that reduces the harmonic content of current in the
winding relative to back EMF in the winding.
2 . A method as claimed in claim 1, wherein the secondary freewheel period occurs
at a time when back EMF in the winding is rising, and the primary freewheel period
occurs at a time when back EMF is principally falling.
3 . A method as claimed in claim 1 or 2, wherein the length of the secondary
freewheel period is less than each of the primary freewheel period, the first excitation
period and the second excitation period.
4 . A method as claimed in any one of the preceding claims, wherein the method
comprises exciting the winding with a supply voltage, and varying the length of the
conduction period in response to changes in the supply voltage or the speed of the
motor.
5 . A method as claimed in claim 4, wherein the method comprises increasing the
length of the conduction period in response to a decrease in the supply voltage or an
increase in the speed of the motor.
6 . A method as claimed in any one of the preceding claims, wherein the first
excitation period and the second excitation period have the same length.
7 . A method as claimed in any one of the preceding claims, wherein the length of
the secondary freewheel period is fixed.
8 . A method as claimed in any one of the preceding claims, wherein the method
comprises exciting the winding with a supply voltage, and varying the secondary
freewheel period in response to changes in the supply voltage or the speed of the motor.
9 . A method as claimed in claim 8, wherein the method comprises increasing the
length of the secondary freewheel period in response to an increase in the supply
voltage or a decrease in the speed of the motor.
10. A method of controlling a brushless permanent-magnet motor, the method
comprising operating in dual-switch mode over a first speed range and operating in
single-switch mode over a second speed range, wherein the second speed range is
higher than the first speed range, each mode comprises dividing each half of an
electrical cycle into a conduction period followed by a primary freewheel period, the
single-switch mode comprises exciting a winding of the motor during the conduction
period and freewheeling the winding during the freewheel period, and dual-switch mode
comprises: dividing the conduction period into a first excitation period, a secondary
freewheel period and a second excitation period; exciting the winding during each
excitation period; and freewheeling the winding during each freewheel period.
11. A method as claimed in claim 10, wherein each mode comprises exciting the
winding with a supply voltage, and varying the length of the conduction period in
response to changes in the supply voltage or the speed of the motor.
12. A method of controlling a brushless permanent-magnet motor, the method
comprising operating in multi-switch mode over a first speed range and operating in
dual-switch mode over a second speed range, wherein the second speed range is higher
than the first speed range, multi-switch mode comprises sequentially exciting and
freewheeling a winding of the motor multiple times during each half of an electrical
cycle, the winding being freewheeled when current in the winding exceeds a predefined
limit, and dual-switch mode comprises: dividing each half of an electrical cycle into a
conduction period followed by a primary freewheel period; dividing the conduction
period into a first excitation period, a secondary freewheel period and a second
excitation period; exciting the winding during each excitation period; and freewheeling
the winding during each freewheel period.
13. A control circuit for a brushless permanent-magnet motor, the control circuit
being configured to perform a method as claimed in any one of the preceding claims.
14. A control circuit as claimed in claim 13, wherein the control circuit comprises
an inverter for coupling to a winding of the motor, a gate driver module and a controller,
the gate driver module controls switches of the inverter in response to control signals
received from the controller, and the controller generates control signals to excite the
winding during each excitation period and to freewheel the winding during each
freewheel period.
15. A motor assembly comprising a brushless permanent-magnet motor and a
control circuit as claimed in claim 13 or 14.