Method of Controlling of a Brushless Permanent-Magnet Motor
The present invention relates to a method of controlling a brushless permanent-magnet
motor.
There is a growing need to improve the efficiency of brushless permanent-magnet
motors.
The present invention provides a method of controlling a brushless permanent-magnet
motor, the method comprising commutating a winding of the motor at times that are
retarded relative to zero-crossings of back EMF in the winding when operating at
speeds greater than 50 krpm.
At speeds greater than 50 krpm, the length of each electrical half-cycle is relatively
short and the magnitude of the back EMF is relatively large. Both of these factors
would suggest that advanced commutation is necessary in order to drive sufficient
current and thus power into the phase winding in order to maintain such speeds.
However, the applicant has identified that, once the motor is at this speed,
improvements in the efficiency of the motor may be achieved by retarding
commutation. For a permanent-magnet motor, the torque-to-current ratio is at a
maximum when the waveform of the phase current matches that of the back EMF.
Improvements in the efficiency of the motor are therefore achieved by shaping the
waveform of the phase current such that it better matches the waveform of the back
EMF. In instances for which the phase current rises faster than the back EMF at around
zero-crossings in the back EMF, advanced commutation would cause the phase current
to quickly lead the back EMF. By retarding commutation until after each zero-crossing
in the back EMF, the rise in the phase current may be made to more closely follow that
of the back EMF. As a result, the efficiency of the motor may be improved.
The method may comprise commutating the winding at times that are retarded relative
to zero-crossings of back EMF when operating over a speed range spanning at least
5 krpm, and more preferably at least 10 krpm. As a result, improvements in the
efficiency of the motor may be achieved over a relatively large range of speeds.
Changes in the magnitude of the supply voltage used to excite the winding will
influence the rate at which the phase current rises. Changes in the speed of the motor
will influence the length of each electrical half-cycle and thus the rate at which the back
EMF rises. Additionally, changes in the speed of the motor will influence the
magnitude of the back EMF and thus the rate at which the phase current rises.
Accordingly, the method may comprise retarding commutation by a retard period and
varying the retard period in response to changes in the supply voltage and/or the speed
of the motor. This then has the advantage that the efficiency of the motor may be
improved as the motor operates over a range of supply voltages and/or motor speeds.
Additionally, the amount of current and thus power that is driven into the winding is
sensitive to changes in the supply voltage and/or the motor speed. By varying the retard
period in response to changes in the supply voltage and/or motor speed, better control
may be achieved over the input or output power of the motor.
The method may comprise increasing the retard period in response to an increase in the
supply voltage and/or a decrease in the motor speed. As the magnitude of the supply
voltage increases, the phase current rises at a faster rate. As the speed of the motor
decreases, the back EMF rises at a slower rate. Additionally, the magnitude of the back
EMF decreases and thus the phase current rises at a faster rate. By increasing the retard
period in response to an increase in the supply voltage and/or a decrease in the motor
speed, the waveform of the phase current may be made to better match that of the back
EMF in response to changes in the supply voltage and/or the motor speed. As a result,
the efficiency of the motor may be improved when operating over a range of supply
voltages and/or speeds.
The method may comprise dividing each electrical half-cycle into a conduction period
followed by a freewheel period. The winding is then excited during the conduction
period and freewheeled during the freewheel period. Moreover, the method may
comprise dividing the conduction period into a first excitation period, a further
freewheel period and a second excitation period, and the winding may be excited during
each excitation period and the winding may be freewheeled during the further freewheel
period. Although commutation is retarded, the phase current may nevertheless rise at a
faster rate than that of the back EMF. As a result, the phase current may eventually lead
the back EMF. The secondary freewheel period serves to check momentarily the rise in
the phase current. Consequently, the phase current may be made to more closely follow
the rise of the back EMF during the conduction period, thereby improving the
efficiency.
The method may comprise commutating the winding at times that are retarded relative
to zero-crossings of back EMF when operating over a first speed range, commutating
the winding at times that are advanced relative to zero-crossings of back EMF in the
winding when operating over a second speed range, and the second speed range is
higher than the first speed range. When operating over the first speed range, the phase
current rises at a faster rate than that of the back EMF at around zero-crossings.
Accordingly, by retarding commutation, improvements in the efficiency of the motor
may be achieved. When operating over the second speed range, the length of each
electrical half-cycle is shorter and thus the back EMF rises at a faster rate. Additionally,
the magnitude of the back EMF is higher and thus the phase current rises at a slower
rate. The back EMF therefore rises at a faster rate but the phase current rises at a slower
rate. As a result, the phase current rises at a slower rate than that of the back EMF.
Retarding commutation would then only serve to worsen the efficiency of the motor.
Moreover, it may not be possible to drive sufficient current and power into the winding
when operating over the second speed range if commutation is retarded. Accordingly,
by employing retarded commutation over the first speed range and advanced
commutation over the second speed range, the efficiency of the motor may be improved
over both speed ranges.
The method may comprise dividing each electrical half-cycle of the motor into a
conduction period followed by a freewheel period when operating over the first speed
range and the second speed range, the winding being excited during the conduction
period and freewheeled during the freewheel period.
The method may comprise driving the motor at constant power (be it input power or
output power) over the first speed range and the second speed range. The power of the
motor over the first speed range is then lower than that over the second speed range.
The method may comprise retarding commutation by a retard period when the motor
operates over the first speed range, and advancing commutation by an advance period
when the motor operates over the second speed range. Constant power may then be
achieved by varying the retard period and the advance period in response to changes in
the speed of the motor. Additionally or alternatively, constant power may be achieved
by varying the length of the conduction period in response to changes in the speed of
the motor.
The first speed range and the second speed range may each span at least 5 krpm, and
more preferably at least 10 krpm. As a result, improvements in the efficiency of the
motor may be achieved over relatively large speed ranges.
The present invention also provides a control circuit configured to perform a method
described in any one of the preceding paragraphs, as well as a motor assembly that
comprises the control circuit and a brushless permanent-magnet motor.
The control circuit may comprise an inverter for coupling to a winding of the motor, a
gate driver module and a controller. The gate driver module then controls switches of
the inverter in response to control signals received from the controller, and the
controller generates control signals to commutate the winding. More particularly, the
controller generates control signals to commutate the winding at times that are retarded
relative to zero-crossings of back EMF in the winding when the speed of the motor is
greater than 50 krpm.
In order that the present invention may be more readily understood, an embodiment of
the invention will now be described, by way of example, with reference to the
accompanying drawings, in which:
Figure 1 is a block diagram of a motor assembly in accordance with the present
invention;
Figure 2 is a schematic diagram of the motor assembly;
Figure 3 details the allowed states of the inverter in response to control signals issued by
the controller of the motor assembly;
Figure 4 illustrates various waveforms of the motor assembly when operating in
acceleration mode;
Figure 5 illustrates various waveforms of the motor assembly when operating in highpower
mode; and
Figure 6 illustrates various waveforms of the motor assembly when operating in lowpower
mode.
The motor assembly 1 of Figures 1 and 2 is powered by a DC power supply 2 and
comprises a brushless motor 3 and a control circuit 4 .
The motor 3 comprises a four-pole permanent-magnet rotor 5 that rotates relative to a
four-pole stator 6 . Conductive wires wound about the stator 6 are coupled together to
form a single phase winding 7 .
The control circuit 4 comprises a filter 8, an inverter 9, a gate driver module 10, a
current sensor 11, a voltage sensor 12, a position sensor 13, and a controller 14.
The filter 8 comprises a link capacitor CI that smoothes the relatively high-frequency
ripple that arises from switching of the inverter 9 .
The inverter 9 comprises a full bridge of four power switches Q1-Q4 that couple the
phase winding 7 to the voltage rails. Each of the switches Q1-Q4 includes a freewheel
diode.
The gate driver module 10 drives the opening and closing of the switches Q1-Q4 in
response to control signals received from the controller 14.
The current sensor 11 comprises a shunt resistor Rl located between the inverter and the
zero-volt rail. The voltage across the current sensor 11 provides a measure of the
current in the phase winding 7 when connected to the power supply 2 . The voltage
across the current sensor 11 is output to the controller 14 as signal, I PHASE.
The voltage sensor 12 comprises a potential divider R2,R3 located between the DC
voltage rail and the zero volt rail. The voltage sensor outputs a signal, V DC, to the
controller 14 that represents a scaled-down measure of the supply voltage provided by
the power supply 2 .
The position sensor 13 comprises a Hall-effect sensor located in a slot opening of the
stator 6 . The sensor 13 outputs a digital signal, HALL, that is logically high or low
depending on the direction of magnetic flux through the sensor 13. The HALL signal
therefore provides a measure of the angular position of the rotor 5 .
The controller 14 comprises a microcontroller having a processor, a memory device,
and a plurality of peripherals (e.g. ADC, comparators, timers etc.). The memory device
stores instructions for execution by the processor, as well as control parameters and
lookup tables that are employed by the processor during operation of the motor
assembly 1. The controller 14 is responsible for controlling the operation of the motor 3
and generates four control signals S1-S4 for controlling each of the four power switches
Q1-Q4. The control signals are output to the gate driver module 10, which in response
drives the opening and closing of the switches Q1-Q4.
Figure 3 summarises the allowed states of the switches Q1-Q4 in response to the control
signals S1-S4 output by the controller 14. Hereafter, the terms 'set' and 'clear' will be
used to indicate that a signal has been pulled logically high and low respectively. As
can be seen from Figure 3, the controller 14 sets SI and S4, and clears S2 and S3 in
order to excite the phase winding 7 from left to right. Conversely, the controller 14 sets
S2 and S3, and clears SI and S4 in order to excite the phase winding 7 from right to left.
The controller 14 clears SI and S3, and sets S2 and S4 in order to freewheel the phase
winding 7 . Freewheeling enables current in phase the winding 7 to re-circulate around
the low-side loop of the inverter 9 . In the present embodiment, the power switches Ql-
Q4 are capable of conducting in both directions. Accordingly, the controller 14 closes
both low-side switches Q2,Q4 during freewheeling such that current flows through the
switches Q2,Q4 rather than the less efficient diodes. Conceivably, the inverter 9 may
comprise power switches that conduct in a single direction only. In this instance, the
controller 14 would clear SI, S2 and S3, and set S4 so as to freewheel the phase
winding 7 from left to right. The controller 14 would then clear SI, S3 and S4, and set
S2 in order to freewheel the phase winding 7 from right to left. Current in the low-side
loop of the inverter 9 then flows down through the closed low-side switch (e.g. Q4) and
up through the diode of the open low-side switch (e.g. Q2).
The controller 14 operates in one of three modes: acceleration mode, low-power mode
and high-power mode. Low-power mode and high-power mode are both steady-state
modes. The controller 14 receives and periodically monitors a power- mode signal,
POWER MODE, in order to determine which steady-state mode should be employed.
If the power-mode signal is logically low, the controller 14 selects low-power mode,
and if the power-mode signal is logically high, the controller 14 selects high-power
mode. When operating in low-power mode, the controller 14 drives the motor 3 over an
operating speed range of 60-70 krpm. When operating in high-power mode, the
controller 14 drives the motor 3 over an operating speed range of 90-100 krpm.
Acceleration mode is then used to accelerate the motor 3 from stationary to the lower
limit of each operating speed range.
In all three modes the controller 14 commutates the phase winding 7 in response to
edges of the HALL signal. Each HALL edge corresponds to a change in the polarity of
the rotor 5, and thus a change in the polarity of the back EMF induced in the phase
winding 7 . More particularly, each HALL edge corresponds to a zero-crossing in the
back EMF. Commutation involves reversing the direction of current through the phase
winding 7 . Consequently, if current is flowing through the phase winding 7 in a
direction from left to right, commutation involves exiting the winding from right to left.
In the discussion below, reference is made frequently to the speed of the motor 3 . The
speed of the motor 3 is determined from the interval between successive edges of the
HALL signal, which will hereafter be referred to as the HALL period.
Acceleration Mode
At speeds below 20 krpm, the controller 14 commutates the phase winding 7 in
synchrony with each HALL edge. At speeds at or above 20 krpm, the controller 14
commutates the phase winding 7 in advance of each HALL edge. In order to
commutate the phase winding 7 in advance of a particular HALL edge, the controller 14
acts in response to the preceding HALL edge. In response to the preceding HALL edge,
the controller 14 subtracts an advance period, T ADV, from the HALL period,
T HALL, in order to obtain a commutation period, T COM:
T COM = T HALL - T ADV
The controller 14 then commutates the phase winding 7 at a time, T COM, after the
preceding HALL edge. As a result, the controller 14 commutates the phase winding 7
in advance of the subsequent HALL edge by the advance period, T ADV.
Irrespective of whether commutation in synchronous or advanced, the controller 14
sequentially excites and freewheels the phase winding 7 over each half of an electrical
cycle when operating in acceleration mode. More particularly, the controller 14 excites
the phase winding 7, monitors the current signal, I PHASE, and freewheels the phase
winding 7 when the current in the phase winding 7 exceeds a predefined limit.
Freewheeling then continues for a predefined freewheel period during which time
current in the phase winding 7 falls to a level below the current limit. At the end of the
freewheel period the controller 14 again excites the phase winding 7 . This process of
exciting and freewheeling the phase winding 7 continues over the full length of the
electrical half-cycle. The controller 14 therefore switches from excitation to
freewheeling multiple times during each electrical half-cycle.
Figure 4 illustrates the waveforms of the HALL signal, the back EMF, the phase
current, the phase voltage, and the control signals over a couple of HALL periods when
operating in acceleration mode. In Figure 4 the phase winding 7 is commutated in
synchrony with the HALL edges.
At relatively low speeds, the magnitude of the back EMF induced in the phase
winding 7 is relatively small. Current in the phase winding 7 therefore rises relatively
quickly during excitation, and falls relatively slowly during freewheeling. Additionally,
the length of each HALL period and thus the length of each electrical half-cycle is
relatively long. Consequently, the frequency at which the controller 14 switches from
excitation to freewheeling is relatively high. However, as the rotor speed increases, the
magnitude of the back EMF increases and thus current rises at a slower rate during
excitation and falls at a quicker rate during freewheeling. Additionally, the length of
each electrical half-cycle decreases. As a result, the frequency of switching decreases.
The controller 14 continues to operate in acceleration mode until the speed of the rotor 5
reaches the lower limit of the operating speed range of the selected power mode. So, for
example, if high-power mode is selected, the controller 14 continues to operate in
acceleration mode until the speed of the rotor 5 reaches 90 krpm.
High-Power Mode
The controller 14 commutates the phase winding in advance of each HALL edge.
Advanced commutation is achieved in the same manner as that described above for
acceleration mode.
When operating in high-power mode, the controller 14 divides each half of an electrical
cycle into a conduction period followed by a freewheel period. The controller 14 then
excites the phase winding 7 during the conduction period and freewheels the phase
winding 7 during the freewheel period. The phase current is not expected to exceed the
current limit during excitation. Consequently, the controller 14 switches from
excitation to freewheeling only once during each electrical half-cycle.
The controller 14 excites the phase winding 7 for a conduction period, T CD. At the
end of the conduction period, the controller 14 freewheels the phase winding 7 .
Freewheeling then continues indefinitely until such time as the controller 14
commutates the phase winding 7 . The controller 14 therefore controls operation of the
motor 3 using two parameters: the advance period, T ADV, and the conduction period,
T CD.
Figure 5 illustrates the waveforms of the HALL signal, the back EMF, the phase
current, the phase voltage, and the control signals over a couple of HALL periods when
operating in high-power mode.
The magnitude of the supply voltage used to excite the phase winding 7 may vary. For
example, the power supply 2 may comprise a battery that discharges with use.
Alternatively, the power supply 2 may comprise an AC source, rectifier and smoothing
capacitor that provide a relatively smooth voltage, but the RMS voltage of the AC
source may vary. Changes in the magnitude of the supply voltage will influence the
amount of current that is driven into the phase winding 7 during the conduction period.
As a result, the power of the motor 3 is sensitive to changes in the supply voltage. In
addition to the supply voltage, the power of the motor 3 is sensitive to changes in the
speed of the rotor 5 . As the speed of the rotor 5 varies (e.g. in response to changes in
load), so too does the magnitude of the back EMF. Consequently, the amount of current
driven into the phase winding 7 during the conduction period may vary. The
controller 14 therefore varies the advance period and the conduction period in response
to changes in the magnitude of the supply voltage. The controller 14 also varies the
advance period in response to changes in the speed of the rotor 5 .
The controller 14 stores a voltage lookup table that comprises an advance period,
T ADV, and a conduction period, T CD, for each of a plurality of different supply
voltages. The controller 14 also stores a speed lookup table that comprises a speedcompensation
value for each of a plurality of different rotor speeds and different supply
voltages. The lookup tables store values that achieve a particular input or output power
at each voltage and speed point. In the present embodiment, the lookup tables store
values that achieve constant output power for the motor 3 over a range of supply
voltages as well as over the operating speed range for high-power mode.
The V DC signal output by the voltage sensor 12 provides a measure of the supply
voltage, whilst the length of the HALL period provides a measure of the rotor speed.
The controller 14 indexes the voltage lookup table using the supply voltage to select a
phase period and a conduction period. The controller 14 then indexes the speed lookup
table using the rotor speed and the supply voltage to select a speed-compensation value.
The controller 14 then adds the selected speed-compensation value to the selected phase
period so as to obtain a speed-compensated phase period. The commutation period,
T COM, is then obtained by subtracting the speed-compensated phase period from the
HALL period, T HALL.
The speed lookup table stores speed-compensation values that depend not only on the
speed of the rotor 5 but also on the magnitude of the supply voltage. The reason for this
is that, as the supply voltage decreases, a particular speed-compensation value has a
smaller net effect on the output power of the motor 3 . By storing speed-compensation
values that depend on both the rotor speed and the supply voltage, better control over
the output power of the motor 3 may be achieved in response to changes in the rotor
speed.
It will be noted that two lookup tables are used to determine the advance period. The
first lookup table (i.e. the voltage lookup table) is indexed using the supply voltage.
The second lookup table (i.e. the speed lookup table) is indexed using both the rotor
speed and the supply voltage. Since the second lookup table is indexed using both rotor
speed and supply voltage, one might question the need for two lookup tables. However,
the advantage of using two lookup tables is that different voltage resolutions may be
used. The output power of the motor 3 is relatively sensitive to the magnitude of the
supply voltage. In contrast, the effect that the speed-compensation value has on the
output power is less sensitive to the supply voltage. Accordingly, by employing two
lookup tables, a finer voltage resolution may be used for the voltage lookup table, and
coarser voltage resolution may be used for the speed lookup table. As a result,
relatively good control over the output power of the motor 3 may be achieved through
the use of smaller lookup tables, which then reduces the memory requirements of the
controller 14.
Low-Power Mode
The controller 14 commutates the phase winding 7 at times that are retarded relative to
the HALL edges. Retarded commutation is achieved in a manner similar to that for
advanced commutation. In response to a HALL edge, the controller 14 adds a retard
period, T RET, to the HALL period, T HALL, in order to obtain a commutation
period, T COM:
T COM = T HALL + T RET
The controller 14 then commutates the phase winding 7 at a time, T COM, after the
HALL edge. As a result, the controller 14 commutates the phase winding 7 at a time
T RET after the subsequent HALL edge.
When operating in low-power mode, the controller 14 divides each half of an electrical
cycle into a conduction period followed by a primary freewheel period. The
controller 14 then divides the conduction period into a first excitation period, followed
by a secondary freewheel period, followed by a second excitation period. The
controller 14 then excites the phase winding 7 during each of the two excitation periods
and freewheels the phase winding 7 during each of the two freewheel periods. As in
high-power mode, the phase current is not expected to exceed the current limit during
excitation. Accordingly, the controller 14 switches from excitation to freewheeling
twice during each electrical half-cycle.
Figure 6 illustrates the waveforms of the HALL signal, the back EMF, the phase
current, the phase voltage, and the control signals over a couple of HALL periods when
operating in low-power mode.
As in high-power mode, the controller 14 varies the retard period and the conduction
period in response to changes in the magnitude of the supply voltage, and the
controller 14 varies the retard period in response to changes in the speed of the rotor 5 .
The controller 14 therefore stores a further voltage lookup table that comprises different
retard periods, T RET, and different excitation periods, T EXC, for different supply
voltages. The controller 14 also stores a further speed lookup table that comprises
speed-compensation values for different rotor speeds and different supply voltages. The
lookup tables employed in low-power mode therefore differ from those employed in
high-power mode only in that tables store retard periods rather than advance periods,
and excitation periods rather than conduction periods. As in high-power mode, the
lookup tables employed in low-power mode store values that achieve constant output
power for the motor 3 over the same range of supply voltages and over the operating
speed range for low-power mode.
During operation, the controller 14 indexes the voltage lookup table using the supply
voltage to select a retard period and an excitation period. The selected excitation period
is then used to define both the first excitation period and the second excitation period,
i.e. during the conduction period the controller 14 excites the phase winding 7 for the
selected excitation period, freewheels the phase winding 7 for the secondary freewheel
period, and excites the phase winding 7 again for the selected excitation period. As a
result, the secondary freewheel period occurs at the centre of the conduction period.
In comparison to high-power mode, the excitation of the phase winding 7 differs in two
important ways. First, the controller 14 retards commutation. Second, the controller 14
introduces a secondary freewheel period into the conduction period. The reasons for
and benefits of these two differences will now be explained.
When operating in high-power mode, advanced commutation is necessary in order to
achieve the necessary output power. As the speed of the rotor 5 increases, the HALL
period decreases and thus the time constant (L/R) associated with the phase inductance
becomes increasingly important. Additionally, the back EMF induced in the phase
winding 7 increases, which in turn influences the rate at which phase current rises. It
therefore becomes increasingly difficult to drive current and thus power into the phase
winding 7 . By commutating the phase winding 7 in advance of each HALL edge, and
thus in advance of the zero-crossings in the back EMF, the supply voltage is boosted
momentarily by the back EMF. As a result, the direction of current through the phase
winding 7 is more quickly reversed. Additionally, the phase current is caused to lead
the back EMF, which helps to compensate for the slower rate of current rise. Although
this then generates a short period of negative torque, this is normally more than
compensated by the subsequent gain in positive torque.
When operating in low-power mode, the length of the HALL period is longer and thus
the back EMF rises at a slower rate. Additionally, the magnitude of the back EMF is
lower and thus current in the phase winding 7 rises at a faster rate for a given supply
voltage. The back EMF therefore rises at a slower rate but the phase current rises at a
faster rate. It is not therefore necessary to commutate the phase winding 7 in advance of
the HALL edges in order to achieve the desired output power. Moreover, for reasons
that will now be explained, the efficiency of the motor assembly 3 is improved by
retarding commutation.
During excitation, the torque-to-current ratio is at a maximum when the waveform of
the phase current matches that of the back EMF. Improvements in the efficiency of the
motor 3 are therefore achieved by shaping the waveform of the phase current such that
it better matches the waveform of the back EMF, i.e. by reducing the harmonic content
of the phase current waveform relative to the back EMF waveform. As noted in the
preceding paragraph, when operating in low-power mode, the back EMF rises at a
slower rate but the phase current rises at a faster rate. Indeed, when operating in lowpower
mode, the phase current rises faster than the back EMF when the magnitude of
the back EMF is relatively low, i.e. at around zero-crossings in the back EMF.
Consequently, if the phase winding 7 were commutated in advance of or in synchrony
with the HALL edges, the phase current would quickly lead the back EMF. In highpower
mode, it was necessary for the phase current to initially lead the back EMF in
order to compensate for the shorter HALL period and the slower rise in the phase
current. In low-power mode, however, it is not necessary for the phase current to lead
the back EMF in order to achieve the necessary output power. By retarding
commutation until after each HALL edge, the phase current more closely follows the
rise in the back EMF. As a result, the efficiency of the motor assembly 1 is improved.
The secondary freewheel period acts to further improve the efficiency of the motor 3 .
As a result of retarding commutation, the phase current more closely matches that of the
back EMF. Nevertheless, the phase current continues to rise at a faster rate than that of
the back EMF. Consequently, the phase current eventually overtakes the back EMF.
By introducing a relatively small secondary freewheel period into the conduction
period, the rise in the phase current is checked momentarily such that the rise in the
phase current more closely follows the rise in the back EMF. As a result, the harmonic
content of the phase current waveform relative to the back EMF waveform is further
reduced and thus the efficiency of the motor 3 is further increased.
Acceleration mode is used to accelerate the motor 3 from stationary to the lower limit of
each operating speed range. Consequently, the controller 14 operates in acceleration
mode between 0 and 90 krpm when high-power mode is selected, and between 0 and
60 krpm when low-power mode is selected. Irrespective of which power mode has been
selected, the controller 14 commutates the phase winding 7 in synchrony with the
HALL edges between 0 and 20 krpm. The controller 14 then commutates the phase
winding 7 in advance of the HALL edges as the motor 3 accelerates from 20 to 90 krpm
(high-power mode) or from 20 to 60 krpm (low-power mode). When operating in
acceleration mode, the controller 14 employs an advance period that remains fixed as
the rotor 5 accelerates. Irrespective of whether high-power mode or low-power mode is
selected, the controller 14 indexes the voltage lookup table employed in high-power
mode using the supply voltage in order to select an advance period. The selected
advance period is then used by the controller 14 during acceleration mode.
Conceivably, the efficiency of the motor 3 may be improved by employing an advance
period that varies with rotor speed. However, this would then require an additional
lookup table. Moreover, acceleration mode is typically short-lived, with the
controller 14 operating predominantly in low-power mode or high-power mode.
Consequently, any efficiency improvements that may be made by varying the advance
period within acceleration mode are unlikely to contribute significantly to the overall
efficiency of the motor 3 .
On switching from acceleration mode to low-power mode, the controller 14 switches
from advanced commutation to retarded commutation. When operating within the lowpower
mode, the controller 14 is able to drive sufficient current and thus power into the
phase winding 7 during each electrical half-cycle whilst simultaneously employing
retarded commutation in order to improve the efficiency of the motor assembly 1. In
contrast, during acceleration, advanced commutation is necessary in order to ensure that
sufficient current and thus power is driven into the phase winding 7 during each
electrical half-cycle. If the controller 14 were to retard or synchronise commutation
during acceleration, the rotor 5 would fail to accelerate to the required speed.
Accordingly, whilst retarded commutation may be employed in order to maintain the
rotor speed over a speed range of 60-70 krpm, advanced commutation is necessary in
order to ensure that the rotor accelerates to 60 krpm.
Whilst it is known to retard commutation when operating at relatively low speeds, it is
completely unknown to retard commutation when operating at relatively high speeds,
i.e. at speeds in excess of 50 krpm. At these relatively high speeds, the relatively short
length of each HALL period and the magnitude of the back EMF would suggest that
advanced commutation is necessary. Indeed, advanced commutation is necessary in
order to accelerate the motor to such speeds. The applicant has, however, identified that
once at these speeds, commutation may be retarded so as to improve the efficiency of
the motor 3.
In the embodiment described above, the controller 14 employs two steady-state modes:
high-power mode and low-power mode. In high-power mode, the controller 14
commutates the phase winding 7 in advance of zero-crossings in the back EMF. In lowpower
mode, the controller 14 commutates the phase winding 7 in retard of zerocrossings
in the back EMF. The advance period and the retard period may each be
regarded as a phase period, T PHASE, and the commutation period, T COM may be
defined as:
T COM = T HALL - T PHASE
If the phase period is positive, commutation occurs before the HALL edge (i.e.
advanced commutation), and if the phase period is negative, commutation occurs after
the HALL edge (i.e. retarded commutation). Employing the same scheme to commutate
the phase winding 7 in both high-power mode and low-power mode simplifies the
control. However, conceivably different methods may be used to commutate the phase
winding 7 . For example, when operating in low power mode, the controller 14 may
simply commutate the phase winding 7 at a time, T RET, after each HALL edge.
In the embodiment described above, the controller 14 varies only the phase period (i.e.
the advance period in high-power mode and the retard period in low-power mode) in
response to changes in the rotor speed. In comparison to the conduction period, the
input power of the motor 3 is typically more sensitive to changes in the phase period.
Accordingly, better control over the output power of the motor 3 may be achieved by
varying the phase period. Nevertheless, in spite of these advantages, the controller 14
may instead vary only the conduction period in response to changes in the rotor speed.
Alternatively, the controller 14 may vary both the phase period and the conduction
period in response to changes in the rotor speed. This may be necessary if, for example,
the output power of the motor 3 cannot be controlled adequately by varying just the
phase period. Alternatively, improvements in the efficiency of the motor 3 may be
achieved by varying both the phase period and the conduction period in response to
changes in the rotor speed. However, a disadvantage of varying both the phase period
and the conduction period is that additional lookup tables are required, thus placing
additional demands on the memory of the controller 14.
In the embodiment described above, the controller 14 varies the phase period and the
conduction period in response to changes in the supply voltage. This then has the
advantage that the efficiency of the motor 3 may be better optimised at each voltage
point. Nevertheless, it may be possible to achieve the desired control over the output
power of the motor 3 by varying just one of the phase period and the conduction period.
Since the output power of the motor 3 is more sensitive to changes in the phase period,
better control over the output power of the motor 3 may be achieved by varying the
phase period.
The controller 14 may therefore be said to vary the phase period and/or the conduction
period in response to changes in the supply voltage and the rotor speed. Whilst the two
periods may be varied in response to changes in the supply voltage and the rotor speed,
the controller 14 could conceivably vary the periods in response to only one of the
supply voltage and the rotor speed. For example, the voltage provided by the power
supply 2 may be relatively stable. In the instance, the controller 14 might vary the
phase period and/or the conduction period in response to changes in the rotor speed
only. Alternatively the motor 3 may be required to operate at constant speed or over a
relatively small range of speeds within low-power mode and high-power mode. In this
instance, the controller 14 might vary the phase period and/or the conduction period in
response to changes in the supply voltage only. Accordingly, in a more general sense,
the controller 14 may be said to vary the phase period and/or the conduction period in
response to changes in the supply voltage and/or the rotor speed. Moreover, rather than
storing a voltage lookup table or a speed lookup table, the controller 14 may be said to
store a power lookup table that comprises different control values for different supply
voltages and/or rotor speeds. Each control value then achieves a particular output
power at each voltage and/or speed point. The controller 14 then indexes the power
lookup table using the supply voltage and/or the rotor speed to select a control value
from the power lookup table. The control value is then used to define the phase period
or the conduction period.
In the embodiment described above, the controller 14 stores lookup tables that
comprises conduction periods for use in high-power mode and excitation periods for use
in low-power mode. However, the same level of control may be achieved by different
means. For example, rather than storing a lookup table of conduction periods and
excitation periods, the controller 14 could store a lookup table of primary freewheel
periods, which is likewise indexed using the magnitude of the supply voltage and/or the
speed of the rotor 5 . The conduction period would then be obtained by subtracting the
primary freewheel period from the HALL period, and each excitation period would be
obtained by subtracting the primary and the secondary freewheel periods from the
HALL period and dividing the result by two:
T CD = T HALL - T FW l
T EXC = (T HALL - T FW l - T_FW_2)/2
where T CD is the conduction period, T EXC is each of the first and second excitation
periods, T HALL is the HALL period, T FW l is the primary freewheel period, and
T FW 2 is the secondary freewheel period.
In the embodiment described above, the secondary freewheel period occurs at the very
centre of the conduction period. This is achieved by ensuring that the same excitation
period is used to define the lengths of the first excitation period and the second
excitation period. There are at least two advantages in ensuring that the secondary
freewheel period occurs at the centre of the conduction period. First, the harmonic
content of the phase current is better balanced over the two excitation periods. As a
result, the total harmonic content of the phase current over the conduction period is
likely to be lower than if the two excitation periods were of different lengths. Second,
the lookup table need only store one excitation period for each voltage point. As a
result, less memory is required for the lookup table. In spite of the aforementioned
advantages, it may be desirable to alter the position of the secondary freewheel period in
response to changes in the supply voltage and/or the rotor speed. This may be achieved
by employing a lookup table that stores a first excitation period and a second excitation
period for different voltages and/or speeds.
The controller 14 employs a secondary freewheel period that is fixed in length. This
then has the advantage of reducing the memory requirements of the controller 14.
Alternatively, however, the controller 14 might employ a secondary freewheel period
that varies in response to changes in the supply voltage and/or the rotor speed. In
particular, the controller 14 may employ a secondary freewheel period that increases in
response to an increase in the supply voltage or a decrease in the rotor speed. As the
supply voltage increases, current in the phase winding 7 rises at a faster rate during
excitation, assuming that the rotor speed and thus the magnitude of the back EMF is
unchanged. As a result, the harmonic content of the phase current waveform relative to
the back EMF waveform is likely to increase. By increasing the length of the secondary
freewheel period in response to an increase in the supply voltage, the rise in the phase
current is checked for a longer period and thus the harmonic content of the phase
current waveform may be reduced. As the rotor speed decreases, the length of the
HALL period increases and thus the back EMF rises at a slower rate. Additionally, the
magnitude of the back EMF decreases and thus current in the phase winding 7 rises at a
faster rate, assuming that the supply voltage is unchanged. Consequently, as the rotor
speed decreases, the back EMF rises at a slower rate but the phase current rises at a
faster rate. The harmonic content of the phase current waveform relative to the back
EMF waveform is therefore likely to increase. By increasing the secondary freewheel
period in response to a decrease in the rotor speed, the rise in the phase current is
checked for a longer period and thus the harmonic content of the phase current
waveform may be reduced. Accordingly, increasing the secondary freewheel period in
response to an increase in the supply voltage and/or a decrease in the rotor speed may
result in further improvements in efficiency.
The length of the secondary freewheel period is relatively short and is intended only to
check momentarily the rise in the phase current. Accordingly, the secondary freewheel
period is shorter than both the primary freewheel period and each of the excitation
periods. The actual length of the secondary freewheel period will depend upon the
particular characteristics of the motor assembly 1, e.g. the inductance of the phase
winding 7, the magnitude of the supply voltage, the magnitude of the back EMF etc.
Irrespective of the length, the secondary freewheel period occurs during a period of
rising back EMF in the phase winding 7 . This is contrast to the primary freewheel
period, which occurs principally if not wholly during a period of falling back EMF. The
primary freewheel period makes use of the inductance of the phase winding 7 such that
torque continues to be generated by the phase current without any additional power
being drawn from the power supply 2 . As the back EMF falls, less torque is generated
for a given phase current. Accordingly, by freewheeling the phase winding 7 during the
period of falling back EMF, the efficiency of the motor assembly 1 may be improved
without adversely affecting the torque.

CLAIMS
1. A method of controlling a brushless permanent-magnet motor, the method
comprising commutating a winding of the motor at times that are retarded relative to
zero-crossings of back EMF in the winding when operating at speeds greater than
50 krpm.
2 . A method as claimed in claim 1, wherein the method comprises commutating
the winding at times that are retarded relative to zero-crossings of back EMF when
operating over a speed range spanning at least 5 krpm.
3 . A method as claimed in claim 1 or 2, wherein the method comprising
commutating the winding at times that are retarded relative to zero-crossings of back
EMF in the winding by a retard period, and varying the retard period in response to
changes in a supply voltage used to excite the winding or the speed of the motor.
4 . A method as claimed in claim 3, wherein the method comprises increasing the
retard period in response to increase in the supply voltage or a decrease in the speed of
the motor.
5 . A method as claimed in any one of the preceding claims, wherein the method
comprises dividing each electrical half-cycle into a conduction period followed by a
freewheel period, the winding being excited during the conduction period and
freewheeled during the freewheel period.
6 . A method as claimed in claim 5, wherein the method comprises dividing the
conduction period into a first excitation period, a further freewheel period and a second
excitation period, the winding is excited during each excitation period and the winding
is freewheeled during the further freewheel period.
7 . A method as claimed in any one of the preceding claims, wherein the method
comprises commutating the winding at times that are retarded relative to zero-crossings
of back EMF when operating over a first speed range, commutating the winding at
times that are advanced relative to zero-crossings of back EMF in the winding when
operating over a second speed range, and the second speed range is higher than the first
speed range.
8 . A method as claimed in claim 7, wherein the method comprises dividing each
electrical half-cycle of the motor into a conduction period followed by a freewheel
period when operating over the first speed range and the second speed range, the
winding being excited during the conduction period and freewheeled during the
freewheel period.
9 . A method as claimed in claim 7 or 8, wherein the method comprises driving the
motor at constant power over the first speed range and driving the motor at a higher
constant power over the second speed range.
10. A method as claimed in any one of claims 7 to 9, wherein the first speed range
and the second speed range each span at least 5 krpm.
11. A control circuit for a brushless permanent-magnet motor, the control circuit
being configured to perform a method as claimed in any one of the preceding claims.
12. A control circuit as claimed in claim 11, wherein the control circuit comprises
an inverter for coupling to a winding of the motor, a gate driver module and a controller,
the gate driver module controls switches of the inverter in response to control signals
received from the controller, and the controller generates control signals to commutate
the winding.
13. A motor assembly comprising a brushless permanent-magnet motor and a
control circuit as claimed in claim 11 or 12.