DESCRIPTION [TITLE OF THE INVENTION] METHOD OF SIGNAL GENERATION AND SIGNAL GENERATING DEVICE 5 [Technical Field] [OOO 11 (CROSS-REFERENCE TO RELATED APPLICATIONS) This application is based on applications No. 201 1-033771 filed February 1 8,20 1 1,20 1 1-05 1 842 filed March 9,20 1 1,20 1 1-093544 filed April 19,20 1 1, and 10 201 1-1 021 01 filed April 28, 201 1 in Japan, the contents of which are hereby incorporated by reference. The present invention relates to a transmission device and a reception device for communication using multiple antennas. 1 5 [Background Art] [0002] A MIMO (Multiple-Input, Multiple-Output) system is an example of a conventional communication system using multiple antennas. In multi-antenna communication, of which the MIMO system is typical, multiple transmission signals 20 are each modulated, and each modulated signal is simultaneously transmitted fiom a different antenna in order to increase the transmission speed of the data. [0003] Fig. 23 illustrates a sample configuration of a transmission and reception device having two transmit antennas and two receive antennas, and using two 25 transmit modulated signals (transmit streams). In the transmission device, encoded data are interleaved, the interleaved data are modulated, and fiequency conversion and the like are performed to generate transmission signals, which are then transmitted fiom antennas. In this case, the scheme for simultaneously transmitting 1 different modulated signals fiom different transmit antennas at the same time and on a common fiequency is a spatial multiplexing MIMO system. [0004] 5 In this context, Patent Literature 1 suggests using a transmission device provided with a different interleaving pattern for each transmit antenna. That is, the transmission device fiom Fig. 23 should use two distinct interleaving patterns performed by two interleavers (IT, and IT,,). As for the reception device, Non-Patent Literature 1 and Non-Patent Literature 2 describe improving reception quality by 10 iteratively using soft values for the detection scheme (by the MIMO detector of Fig. 23). [0005] As it happens, models of actual propagation environments in wireless communications include NLOS (Non Line-Of-Sight), typified by a Rayleigh fading 15 environment is representative, and LOS (Line-Of-Sight), typified by a Rician fading environment. When the transmission device transmits a single modulated signal, and the reception device performs maximal ratio combination on the signals received by a plurality of antennas and then demodulates and decodes the resulting signals, excellent reception quality can be achieved in a LOS environment, in 20 particular in an environment where the Rician factor is large. The Rician factor represents the received power of direct waves relative to the received power of scattered waves. However, depending on the transmission system (e.g., a spatial multiplexing MIMO system), a problem occurs in that the reception quality deteriorates as the Rician factor increases (see Non-Patent Literature 3). 25 Figs. 24A and 24B illustrate an example of simulation results of the BER (Bit Error Rate) characteristics (vertical axis: BER, horizontal axis: SNR (signal-to-noise ratio) for data encoded with LDPC (low-density parity-check) codes and transmitted over a 2 x 2 (two transmit antennas, two receive antennas) spatial 2 multiplexing MIMO system in a Rayleigh fading environment and in a Rician fading environment with Rician factors of K = 3, 10, and 16 dB. Fig. 24A gives the Max-Log approximation-based log-likelihood ratio (Max-log APP) BER characteristics without iterative detection (see Non-Patent Literature I and 5 Non-Patent Literature 2), while Fig. 24B gives the Max-log APP BER characteristic with iterative detection (see Non-Patent Literature 1 and Non-Patent Literature 2) (number of iterations: five). Figs. 24A and 24B clearly indicate that, regardless of whether or not iterative detection is performed, reception quality degrades in the spatial multiplexing MIMO system as the Rician factor increases. Thus, the 10 problem of reception quality degradation upon stabilization of the propagation environment in the spatial multiplexing MIMO system, which does not occur in a conventional single-modulation signal system, is unique to the spatial multiplexing MIMO system. [0006] 15 Broadcast or multicast communication is a service applied to various propagation environments. The radio wave propagation environment between the broadcaster and the receivers belonging to the users is often a LOS environment. When using a spatial multiplexing MIMO system having the above problem for broadcast or multicast communication, a situation may occur in which the received 20 electric field strength is high at the reception device, but in which degradation in reception quality makes service reception difficult. In other words, in order to use a spatial multiplexing MIMO system in broadcast or multicast communication in both the NLOS environment and the LOS environment, a MIMO system that offers a certain degree of reception quality is desirable. 25 [0007] Non-Patent Literature 8 describes a scheme for selecting a codebook used in precoding (i.e. a precoding matrix, also referred to as a precoding weight matrix) based on feedback information from a communication party. However, Non-Patent 3 Literature 8 does not at all disclose a scheme for precoding in an environment in which feedback information cannot be acquired fiom the other party, such as in the above broadcast or multicast communication. [OOOS] 5 On the other hand, Non-Patent Literature 4 discloses a scheme for switching the precoding matrix over time. This scheme is applicable when no feedback information is available. Non-Patent Literature 4 discloses using a unitary matrix as the precoding matrix, and switching the unitary matrix at random, but does not at all disclose a scheme applicable to degradation of reception quality in the 10 above-described LOS environment. Non-Patent Literature 4 simply recites hopping between precoding matrices at random. Obviously, Non-Patent Literature 4 makes no mention whatsoever of a precoding method, or a structure of a precoding matrix, for remedying degradation of reception quality in a LOS environment. [Citation List] 15 [Patent Literature] [0009] [Patent Literature 11 International Patent Application Publication No. W020051050885 won-Patent Literature] 20 [OOlO] won-Patent Literature 11 "Achieving near-capacity on a multiple-antenna channel" IEEE Transaction on communications, vo1.5 1, no.3, pp.389-399, March 2003 won-Patent Literature 21 25 "Performance analysis and design optimization of LDPC-coded MIMO OFDM systems" IEEE Trans. Signal Processing, vo1.52, no.2, pp.348-361, Feb. 2004 won-Patent Literature 31 "BER performance evaluation in 2x2 MIMO spatial multiplexing systems under Rician fading channels" IEICE Trans. Fundamentals, vol.E91 -A, no. 10, pp.2798-2807, Oct. 2008 won-Patent Literature 41 "Turbo space-time codes with time varying linear transformations" IEEE Trans. Wireless communications, vo1.6, no.2, pp.486-493, Feb. 2007 won-Patent Literature 51 "Likelihood function for QR-MLD suitable for soft-decision turbo decoding and its performance" IEICE Trans. Commun., vol.E88-B, no.1, pp.47-57, Jan. 2004 won-Patent Literature 61 "A tutorial on 'Parallel concatenated (Turbo) coding', 'Turbo (iterative) decoding' and related topics" IEICE, Technical Report IT98-5 1 [Non-Patent Literature 71 "Advanced signal processing for PLCs: Wavelet-OFDM Proc. of IEEE International symposium on ISPLC 2008, pp. 187- 192,2008 won-Patent Literature 81 D. J. Love and R. W. Heath Jr., "Limited feedback unitary preceding for spatial multiplexing systems" IEEE Trans. Inf. Theory, vo1.51, no.8, pp.2967-1976, Aug. 2005 won-Patent Literature 91 DVB Document A122, Framing structure, channel coding and modulation for a second generation digital terrestrial television broadcasting system (DVB-T2), June 2008 won-Patent Literature 101 L. Vangelista, N. Benvenuto, and S. Tomasin "Key technologies for next-generation terrestrial digital television standard DVB-T2," IEEE Commun. Magazine, vo.47, no.10, pp.146-153, Oct. 2009 won-Patent Literature 111 5 T. Ohgane, T. Nishimura, and Y. Ogawa, "Application of space division multiplexing and those performance in a MIMO channel" IEICE Trans. Cornmun., vo.88-B, no.5, pp.1843-1851, May 2005 Won-Patent Literature 121 5 R. G. Gallager "Low-density parity-check codes," IRE Trans. Inform. Theory, IT-8, pp.21-28, 1962 won-Patent Literature 131 D. J. C. Mackay, "Good error-correcting codes based on very sparse matrices," IEEE Trans. Inform. Theory, vo1.45, no.2, pp.399-431, March 1999. 10 won-Patent Literature 141 ETSI EN 302 307, "Second generation framing structure, channel coding and modulation systems for broadcasting, interactive services, news gathering and other broadband satellite applications" v.1.1.2, June 2006 won-Patent Literature 151 15 Y.-L. Ueng, and C.-C. Cheng "A fast-convergence decoding method and memory-efficient VLSI decoder architecture for irregular LDPC codes in the IEEE 802.16e standards" IEEE VTC-2007 Fall, pp. 1255-1259 won-Patent Literature 161 S. M. Alamouti "A simple transmit diversity technique for wireless 20 communications" IEEE J. Select. Areas Commun., vol. 16, no.8, pp. 145 1-1458, Oct 1998 won-Patent Literature 171 V. Tarokh, H. Jafrkhani, and A. R. Calderbank "Space-time block coding for wireless communications: Performance results" IEEE J. Select. Areas Commun., 25 vo1.17,no.3,no.3,pp.451-460,March1999 [Summary of Invention] [Technical Problem] [OOl 11 An object of the present invention is to provide a MIMO system that improves reception quality in a LOS environment. [Solution to Problem] [OO 121 5 The present invention provides a signal generation method for generating, from a plurality of baseband signals, a plurality of signals for transmission on a common frequency band and at a common time, comprising the steps of: multiplying a fust baseband signal sl generated from a fust set of bits by u, and multiplying a second baseband signal s2 generated from a second set of bits by v, 10 where u and v denote real numbers different from each other; performing a change of phase on each of the first baseband signal sl multiplied by u and the second baseband signal s2 multiplied by v, thus generating a first post-phase-change baseband signal u x sl' and a second post-phase-change baseband signal v x s2'; and applying weighting according to a predetermined matrix F to the first 15 post-phase-change baseband signal u x sl' and to the second post-phase-change baseband signal v x s2', thus generating the plurality of signals for transmission on the common frequency band and at the common time as a fust weighted signal zl and a second weighted signal 22, wherein the first weighted signal zl and the second weighted signal 22 satis@ the relation: (zl, = F(u x sl', v x ~ 2 ' a)n~d the 20 change of phase is performed on the fust baseband signal sl multiplied by u and the second baseband signal s2 multiplied by v by using a phase modification value sequentially selected from among N phase modification value candidates, each of the N phase modification value candidates being selected at least once within a predetermined period. 25 [0013] The present invention also provides a signal generation apparatus for generating, from a plurality of baseband signals, a plurality of signals for transmission on a common frequency band and at a common time, comprising :a 7 power changer multiplying a first baseband signal sl generated fiom a fust set of bits by u, and multiplying a second baseband signal s2 generated fiom a second set of bits by v, where u and v denote real numbers different fiom each other; a phase changer performing a change of phase on each of the first baseband signal sl 5 multiplied by u and the second baseband signal s2 multiplied by v, thus generating a first post-phase-change baseband signal u x sl' and a second post-phase-change baseband signal v x s2'; and a weighting unit applying weighting according to a predetermined matrix F to the first post-phase-change baseband signal u x sl' and to the second post-phase-change baseband signal v x s2', thus generating the plurality 10 of signals for transmission on the common frequency band and at the common time as a first weighted signal zl and a second weighted signal 22, wherein the fvst weighted signal zl and the second weighted signal 22 satis@ the relation: (zl, = F(u x sl', v x ~ 2 ' a)n~d the change of phase is performed on the first baseband signal sl multiplied by u and the second baseband signal s2 multiplied by v by using 15 a phase modification value sequentially selected from among N phase modification value candidates, each of the N phase modification value candidates being selected at least once within a predetermined period. [Advantageous Effects of Invention] [00 141 20 According to the above structure, the present invention provides a signal generation method and a signal generation apparatus that remedy degradation of reception quality in a LOS environment, thereby providing high-quality service to LOS users during broadcast or multicast communication. [Brief Description of Drawings] 25 [0015] Fig. 1 illustrates an example of a transmission and reception device in a spatial multiplexing MIMO system. Fig. 2 illustrates a sample frame configuration. 8 Fig. 3 illustrates an example of a transmission device applying a phase changing scheme. Fig. 4 illustrates another example of a transmission device applying a phase changing scheme. 5 Fig. 5 illustrates another sample fiame configuration. Fig. 6 illustrates a sample phase changing scheme. Fig. 7 illustrates a sample configuration of a reception device. Fig. 8 illustrates a sample configuration of a signal processor in the reception device. 10 Fig. 9 illustrates another sample configuration of a signal processor in the reception device. Fig. 10 illustrates an iterative decoding scheme. Fig. 1 1 illustrates sample reception conditions. Fig. 12 illustrates a further example of a transmission device applying a 15 phase changing scheme. Fig. 13 illustrates yet a further example of a transmission device applying a phase changing scheme. Fig. 14 illustrates a further sample frame configuration. Fig. 15 illustrates yet another sample fiame configuration. Fig. 16 illustrates still another sample frame configuration. Fig. 17 illustrates still yet another sample frame configuration. Fig. 18 illustrates yet a firher sample fiame configuration. Figs. 19A and 19B illustrate examples of a mapping scheme. Figs. 20A and 20B illustrate further examples of a mapping scheme. Fig. 21 illustrates a sample configuration of a weighting unit. Fig. 22 illustrates a sample symbol rearrangement scheme. Fig. 23 illustrates another example of a transmission and reception device in a spatial multiplexing MIMO system. 9 Figs. 24A and 24B illustrate sample BER characteristics. Fig. 25 illustrates another sample phase changing scheme. Fig. 26 illustrates yet another sample phase changing scheme. Fig. 27 illustrates a further sample phase changing scheme. Fig. 28 illustrates still a further sample phase changing scheme. Fig. 29 illustrates still yet a further sample phase changing scheme. Fig. 30 illustrates a sample symbol arrangement for a modulated signal providing high received signal quality. Fig. 31 illustrates a sample frame configuration for a modulated signal 10 providing high received signal quality. Fig. 32 illustrates another sample symbol arrangement for a modulated signal providing high received signal quality. Fig. 33 illustrates yet another sample symbol arrangement for a modulated signal providing high received signal quality. 15 Fig. 34 illustrates variation in numbers of symbols and slots needed per coded block when block codes are used. Fig. 35 illustrates variation in numbers of symbols and slots needed per pair of coded blocks when block codes are used. Fig. 36 illustrates an overall configuration of a digital broadcasting system. Fig. 37 is a block diagram illustrating a sample receiver. Fig. 38 illustrates multiplexed data configuration. Fig. 39 is a schematic diagram illustrating multiplexing of encoded data into streams. Fig. 40 is a detailed diagram illustrating a video stream as contained in a 25 PES packet sequence. Fig. 41 is a structural diagram of TS packets and source packets in the multiplexed data. Fig. 42 illustrates PMT data configuration. 10 Fig. 43 illustrates information as configured in the multiplexed data. Fig. 44 illustrates the configuration of stream attribute information. Fig. 45 illustrates the configuration of a video display and audio output device. 5 Fig. 46 illustrates a sample configuration of a communications system. Figs. 47A and 47B illustrate a variant sample symbol arrangement for a modulated signal providing high received signal quality. Figs. 48A and 48B illustrate another variant sample symbol arrangement for a modulated signal providing high received signal quality. 10 Figs. 49A and 49B illustrate yet another variant sample symbol arrangement for a modulated signal providing high received signal quality. Figs. 50A and 50B illustrate a further variant sample symbol arrangement for a modulated signal providing high received signal quality. Fig. 5 1 illustrates a sample configuration of a transmission device. 15 Fig. 52 illustrates another sample configuration of a transmission device. Fig. 53 illustrates a further sample configuration of a transmission device. Fig. 54 illustrates yet a further sample configuration of a transmission device. Fig. 55 illustrates a baseband signal switcher. 20 Fig. 56 illustrates yet still a further sample configuration of a transmission device. Fig. 57 illustrates sample operations of a distributor. Fig. 58 illustrates further sample operations of a distributor. Fig. 59 illustrates a sample communications system indicating the 25 relationship between base stations and terminals. Fig. 60 illustrates an example of transmit signal fiequency allocation. Fig. 61 illustrates another example of transmit signal frequency allocation. Fig. 62 illustrates a sample communications system indicating the relationship between a base station, repeaters, and terminals. Fig. 63 illustrates an example of transmit signal frequency allocation with respect to the base station. Fig. 64 illustrates an example of transmit signal frequency allocation with respect to the repeaters. Fig. 65 illustrates a sample configuration of a receiver and transmitter in the repeater. Fig. 66 illustrates a signal data format used for transmission by the base station. Fig. 67 illustrates yet still another sample configuration of a transmission device. Fig. 68 illustrates another baseband signal switcher. Fig. 69 illustrates a weighting, baseband signal switching, and phase changing scheme. Fig. 70 illustrates a sample configuration of a transmission device using an OFDM scheme. Figs. 71A and 71B illustrate further sample frame configurations. Fig. 72 illustrates the numbers of slots and phase changing values corresponding to a modulation scheme. Fig. 73 further illustrates the numbers of slots and phase changing values corresponding to a modulation scheme. Fig. 74 illustrates the overall frame configuration of a signal transmitted by a broadcaster using DVB-T2. Fig. 75 illustrates two or more types of signals at the same time. Fig. 76 illustrates still a fbrther sample configuration of a transmission device. Fig. 77 illustrates an alternate sample frame configuration. 12 Fig. 78 illustrates another alternate sample frame configuration. Fig. 79 illustrates a hrther alternate sample frame configuration. Fig. 80 illustrates an example of a signal point layout for 16-QAM in the IQ plane. 5 Fig. 81 illustrates an example of a signal point layout for QPSK in the IQ plane. Fig. 82 schematically shows absolute values of a log-likelihood ratio obtained by the reception device. Fig. 83 schematically shows absolute values of a log-likelihood ratio 10 obtained by the reception device. Fig. 84 is an example of a structure of a signal processor pertaining to a weighting unit. Fig. 85 is an example of a structure of the signal processor pertaining to the weighting unit. 15 Fig. 86 illustrates an example of a signal point layout for 64-QAM in the IQ plane. Fig. 87 shows the modulation scheme, the power changing value and the phase changing value to be set at each time. Fig. 88 shows the modulation scheme, the power changing value and the 20 phase changing value to be set at each time. Fig. 89 is an example of a structure of the signal processor pertaining to the weighting unit. Fig. 90 is an example of a structure of the signal processor pertaining to the weighting unit. 25 Fig. 91 shows the modulation scheme, the power changing value and the phase changing value to be set at each time. Fig. 92 shows the modulation scheme, the power changing value and the phase changing value to be set at each time. 13 Fig. 93 is an example of a structure of the signal processor pertaining to the weighting unit. Fig. 94 illustrates an example of a signal point layout for 16QAM and QPSK in the IQ plane. 5 Fig. 95 illustrates an example of a signal point layout for 16QAM and QPSK in the IQ plane. [Description of Embodiments] [00 1 61 Embodiments of the present invention are described below with reference to 10 the accompanying drawings. pmbodirnent 11 The following describes, in detail, a transmission scheme, a transmission device, a reception scheme, and a reception device pertaining to the present embodiment. 15 [0017] Before beginning the description proper, an outline of transmission schemes and decoding schemes in a conventional spatial multiplexing MIMO system is provided. [OO 1 81 20 Fig. 1 illustrates the structure of an NtxNr spatial multiplexing MIMO system. An information vector z is encoded and interleaved. The encoded bit vector u = (u,, ... uNt) is obtained as the interleave output. Here, ui = (uil, ... U& (where M is the number of transmitted bits per symbol). For a transmit vector s = (sl, ... SNt), a received signal si = map(ui) is found for transmit antenna #i. 25 Normalizing the transmit energy, this is expressible as ~ ( 1 ~=~ E1s/~N)t ( where E, is the total energy per channel). The receive vector y = (yl, ... yNr)T is expressed in formula 1, below. [OO 1 91 Math. 11 (formula 1) [0020] 5 Here, HNtNirs the channel matrix, n = (nl, ... nNr)i s the noise vector, and the average value of ni is zero for independent and identically distributed (i.i.d) complex Gaussian noise of variance 02. Based on the relationship between transmitted symbols introduced into a receiver and the received symbols, the probability distribution of the received vectors can be expressed as formula 2, below, for a 10 multi-dimensional Gaussian distribution. [002 11 [Math. 21 (formula 2) Here, a receiver performing iterative decoding is considered. Such a receiver is illustrated in Fig. 1 as being made up of an outer soft-inlsoft-out decoder and a MIMO detector. The log-likelihood ratio vector (L-value) for Fig. 1 is given by formula 3 through formula 5, as follows. 20 [0023] Math. 31 (formula 3) [0024] Math. 41 (formula 4) [0025] [Math. 51 (formula 5) [0026] (Iterative Detection Scheme) The following describes the MIMO signal iterative detection performed by the NpN, spatial multiplexing MIMO system. The log-likelihood ratio of u, is defined by formula 6. [0027] [Math. 61 (formula 6) [0028] Through application of Bayes' theorem, formula 6 can be expressed as formula 7. [0029] 16 [Math. 71 (formula 7) [0030] 5 Note that U,, = {ulu, = +I}. Through the approximation lnCaj - max In a,, formula 7 can be approximated as formula 8. The symbol - is herein used to signi@ approximation. [003 11 wath. 81 10 (formula 8) [0032] In formula 8, P(ulu,,) and In P(ulu,,) can be expressed as follows. [0033] 15 Wath.91 (formula 9) [0034] [Math. lo] (formula 10) [0035] [Math. 111 (formula 1 1) 10 [0036] Note that the log-probability of the formula given in formula 2 can be expressed as formula 12. [0037] [Math. 121 15 (formula 12) [003 81 Accordingly, given formula 7 and formula 13, the posterior L-value for the MAP or APP (a posteriori probability) can be can be expressed as follows. . 5 [0039] [Math. 131 (formula 13) [0040] 10 This is hereinafter termed iterative APP decoding. Also, given formula 8 and formula 12, the posterior L-value for the Max-log APP can be can be expressed as follows. . [004 11 [Math. 141 1 5 (formula 14) [0042] [Math. 151 (formula 15) This is hereinafter referred to as iterative Max-log APP decoding. As such, the external information required by the iterative decoding system is obtainable by subtracting prior input fiom formula 13 or fiom formula 14. (System Model) 5 Fig. 23 illustrates the basic configuration of a system related to the following explanations. The illustrated system is a 2x2 spatial multiplexing MIMO system having an outer decoder for each of two streams A and B. The two outer decoders perform identical LDPC encoding (Although the present example considers a configuration in which the outer encoders use LDPC codes, the outer 10 encoders are not restricted to the use of LDPC as the error-correcting codes. The example may also be realized using other error-correcting codes, such as turbo codes, convolutional codes, or LDPC convolutional codes. Further, while the outer encoders are presently described as individually configured for each transmit antenna, no limitation is intended in this regard. A single outer encoder may be 15 used for a plurality of transmit antennas, or the number of outer encoders may be greater than the number of transmit antennas. The system also has interleavers (z, zb) for each of the streams A and B. Here, the modulation scheme is 2 h -(i.e~., ~ ~ h bits transmitted per symbol). [0044] 20 The receiver performs iterative detection (iterative APP (or Max-log APP) decoding) of MIMO signals, as described above. The LDPC codes are decoded using, for example, sum-product decoding. Fig. 2 illustrates the fiame configuration and describes the symbol order after interleaving. Here, (i,j,) and (ibjb)c an be expressed as follows. 25 [0045] Math. 161 (formula 16) [0046] math. 171 (formula 17) [0047] Here, i, and ib represent the symbol order after interleaving, j, and jb represent the bit position in the modulation scheme (where j,jb = I, ... h), ITa,n d IT^ represent the interleaven of streams A and B, and fiajwa and obibrjebpr esent the data 10 order of streams A and B before interleaving. Note that Fig. 2 illustrates a situation where i, = ib. (Iterative Decoding) The following describes, in detail, the sum-product decoding used in decoding the LDPC codes and the MIMO signal iterative detection algorithm, both 15 used by the receiver. [0048] Sum-Product Decoding A two-dimensional MxN matrix H = {Hm) is used as the check matrix for LDPC codes subject to decoding. For the set[l,N] = (1, 2 ... N), the partial sets 20 A(m) and B(n) are defined as follows. [0049] wath. 181 (formula 18) wath. 191 (formula 19) [005 11 5 Here, A(m) signifies the set of column indices equal to 1 for row rn of check matrix H, while B(n) signifies the set of row indices equal to 1 for row n of check matrix H. The sum-product decoding algorithm is as follows. Step A-1 (Initialization): For all pairs (m,n) satisfling H, = 1, set the prior log ratio P, = 1. Set the loop variable (number of iterations) l,, = 1, and set the 10 maximum number of loops Isurn,. Step A-2 (Processing): For all pairs (m,n) satisfling H, = 1 in the order m = 1, 2, ... M , update the extrinsic value log ratio a, using the following update formula. [0052] wath. 201 15 (formula 20) [0053] [Math. 211 (formula 2 1) [0054] [Math. 221 (formula 22) exp(x) + 1 f (x) = In exp(x) - 1 [0055] where f is the Gallager function. h, can then be computed as follows. Step A-3 (Column Operations): For all pairs (m,n) satisfling H,, = 1 in the order n 5 = 1, 2, ... N , update the extrinsic value log ratio using the following update formula. [0056] Math. 231 (formula 23) [0057] Step A-4 (Log-likelihood Ratio Calculation): For nE[l,N], the log-likelihood ratio Ln is computed as follows. [0058] 15 Math. 241 (formula 24) [0059] Step A-5 (Iteration Count): If I, ,,,I ,,, < 1 then l,, is incremented and the 20 process returns to step A-2. Sum-product decoding ends when I, = l,,,. The above describes one iteration of sum-product decoding operations. Afterward, MIMO signal iterative detection is performed. The variables m, n, a,, p,, L,,, and L, used in the above explanation of sum-product decoding operations are expressed as m, n,, aa-, pammmL , and L, for stream A and as mb, nb, ab m bnb, 25 pbmbnhb,b ,a nd Lnbf or stream B. 23 (MIMO Signal Iterative Detection) The following describes the calculation of I,, for MIMO signal iterative detection. 5 The following formula is derivable fiom formula 1. [006 11 Math. 251 (formula 25) Given the frame configuration illustrated in Fig. 2, the following functions are derivable from formula 16 and formula 17. COO631 wath. 261 15 (formula 26) [0064] Math. 271 (formula 27) [0065] where n,nb E[l,N]. For iteration k of MIMO signal iterative detection, the variables L, L, Lb, and Lnb are expressed as hk,, Lk,nruL ,nb,a nd Lk,nb. Step B-1 (Initial Detection; k = 0) 25 For initial wave detection, h,, and &,,nb are calculated as follows. 24 For iterative APP decoding: [0066] [Math. 281 (formula 28) [0067] For iterative Max-log APP decoding: [0068] [Math. 291 10 (formula 29) = max {y(u(ix), y(ix))]- max {y(u(i,), Y (i,))} /20.nx Uo,nx,+, uo.nx,-l [0069] [Math. 301 (formula 30) [0070] where X = a,b. Next, the iteration count for the MIMO signal iterative detection is set to I,,, = 0, with the maximum iteration count being Imimo,-. Step B-2 (Iterative Detection; Iteration k): When the iteration count is k, 20 formula 1 1, formula 13) through formula 15), formula 16), and formula 17) can be expressed as formula 3 1) through formula 34), below. Note that @,Y) = (a,b)(b,a). For iterative APP decoding: [007 11 25 [Math. 311 (formula 3 1) [0072] 5 wath. 321 (formula 32) [0073] For iterative Max-log APP decoding: 10 [0074] [Math. 331 (formula 33) [0075] 15 [Math. 341 (formula 34) Step B-3 (Iteration Count and Codeword Estimation) If lmim n. This is because the phase of direct waves fluctuates slowly in the time domain relative to the frequency domain. Accordingly, the present Embodiment 25 performs a regular change of phase that reduces the influence of steady direct waves. Thus, the phase changing period (cycle) should preferably reduce direct wave fluctuations. Accordingly, m should be greater than n. Taking the above into consideration, using the time and frequency domains together for reordering, as 53 shown in Figs. 17A and 17B, is preferable to using either of the frequency domain or the time domain alone due to the strong probability of the direct waves becoming regular. As a result, the effects of the present invention are more easily obtained. However, reordering in the frequency domain may lead to diversity gain due the fact 5 that frequency-domain fluctuations are abrupt. As such, using the frequency and time domains together for reordering is not always ideal. [0 1601 Figs. 18A and 18B indicate frequency on the horizontal axes and time on the vertical axes thereof, and illustrate an example of a symbol reordering scheme 10 used by the reorderers 1301A and 1301B from Fig. 13 that differs from that of Figs. 17A and 14B. Fig. 18A illustrates a reordering scheme for the symbols of modulated signal zl, while Fig. 18B illustrates a reordering scheme for the symbols of modulated signal 22. Much like Figs. 17A and 17B, Figs. 18A and 18B illustrate the use of the time and frequency domains, together. However, in 15 contrast to Figs. 17A and 17B, where the frequency domain is prioritized and the time domain is used for secondary symbol arrangement, Figs. 18A and 18B prioritize the time domain and use the frequency domain for secondary symbol arrangement. In Fig. 1 8B, symbol group 1802 corresponds to one period (cycle) of symbols when the phase changing scheme is used. 20 [0161] In Figs. 17A, 17B, 18A, and 18B, the reordering scheme applied to the symbols of modulated signal zl and the symbols of modulated signal 22 may be identical or may differ as in Figs. 15A and 15B. Both approaches allow good reception quality to be obtained. Also, in Figs. 17A, 17B, 18A, and 18B, the 25 symbols may be arranged non-sequentially as in Figs. 16A and 16B. Both approaches allow good reception quality to be obtained. [0 1 621 Fig. 22 indicates frequency on the horizontal axis and time on the vertical axis thereof, and illustrates an example of a symbol reordering scheme used by the reorderers 1301A and 1301B from Fig. 13 that differs from the above. Fig. 22 illustrates a regular phase changing scheme using four slots, similar to time u 5 through u+3 from Fig. 6. The characteristic feature of Fig. 22 is that, although the symbols are reordered with respect the frequency domain, when read along the time axis, a periodic shift of n (n = 1 in the example of Fig. 22) symbols is apparent. The frequency-domain symbol group 2210 in Fig. 22 indicates four symbols to which the change of phase is applied at time u through u+3 from Fig. 6. 10 [0163] Here, symbol #O is obtained through a change of phase at time u, symbol #1 is obtained through a change of phase at time u+l, symbol #2 is obtained through a change of phase at time u+2, and symbol #3 is obtained through a change of phase at time u+3. 15 Similarly, for frequency-domain symbol group 2220, symbol #4 is obtained through a change of phase at time u, symbol #5 is obtained through a change of phase at time u+l, symbol #6 is obtained through a change of phase at time u+2, and symbol #7 is obtained through a change of phase at time u+3. [0 1 641 20 The above-described change of phase is applied to the symbol at time $1. However, in order to apply periodic shifting in the time domain, the following phase changes are applied to symbol groups 2201,2202,2203, and 2204. For time-domain symbol group 2201, symbol #O is obtained through a change of phase at time u, symbol #9 is obtained through a change of phase at time 25 u+l, symbol #18 is obtained through a change of phase at time u+2, and symbol #27 is obtained through a change of phase at time u+3. [0165] For time-domain symbol group 2202, symbol #28 is obtained through a change of phase at time u, symbol #1 is obtained through a change of phase at time u+l, symbol #10 is obtained through a change of phase at time u+2, and symbol #19 is obtained through a change of phase at time u+3. 5 For time-domain symbol group 2203, symbol #20 is obtained through a change of phase at time u, symbol #29 is obtained through a change of phase at time u+l, symbol #2 is obtained through a change of phase at time u+2, and symbol #11 is obtained through a change of phase at time u+3. [0 1661 10 For time-domain symbol group 2204, symbol #12 is obtained through a change of phase at time u, symbol #21 is obtained through a change of phase at time u+l, symbol #30 is obtained through a change of phase at time u+2, and symbol #3 is obtained through a change of phase at time u+3. The characteristic feature of Fig. 22 is seen in that, taking symbol #11 as an 15 example, the two neighbouring symbols thereof having the same time in the frequency domain (#lo and #12) are both symbols changed using a different phase than symbol #11, and the two neighbouring symbols thereof having the same carrier in the time domain (#2 and #20) are both symbols changed using a different phase than symbol #I 1. This holds not only for symbol #I 1, but also for any symbol 20 having two neighboring symbols in the frequency domain and the time domain. Accordingly, phase changing is effectively carried out. This is highly likely to improve date reception quality as influence from regularizing direct waves is less prone to reception. [0 1671 25 Although Fig. 22 illustrates an example in which n = 1, the invention is not limited in this manner. The same may be applied to a case in which n = 3. Furthermore, although Fig. 22 illustrates the realization of the above-described effects by arranging the symbols in the frequency domain and advancing in the time 56 domain so as to achieve the characteristic effect of imparting a periodic shift to the symbol arrangement order, the symbols may also be randomly (or regularly) arranged to the same effect. [0168] 5 [Embodiment 21 In Embodiment 1, described above, phase changing is applied to a weighted (precoded with a fixed precoding matrix) signal z(t). The following Embodiments describe various phase changing schemes by which the effects of Embodiment 1 may be obtained. 10 [0169] In the above-described Embodiment, as shown in Figs. 3 and 6, phase changer 3 17B is configured to perform a change of phase on only one of the signals output by the weighting unit 600. However, phase changing may also be applied before precoding is 15 performed by the weighting unit 600. In addition to the components illustrated in Fig. 6, the transmission device may also feature the weighting unit 600 before the phase changer 3 17B, as shown in Fig. 25. [0 1701 In such circumstances, the following configuration is possible. The phase 20 changer 3 17B performs a regular change of phase with respect to baseband signal s2(t), on which mapping has been performed according to a selected modulation scheme, and outputs s2'(t) = s2(t)y(t) (where y(t) varies over time t). The weighting unit 600 executes precoding on s2'4 outputs z2(t) = W2s2'(t) (see formula 42) and the result is then transmitted. 25 [0171] Alternatively, phase changing may be performed on both modulated signals sl(t) and s2(t). As such, the transmission device is configured so as to include a phase changer taking both signals output by the weighting unit 600, as shown in Fig. 26. Like phase changer 3 17B, phase changer 3 17A performs regular a regular change of phase on the signal input thereto, and as such changes the phase of signal 5 zl'(t) precoded by the weighting unit. Post-phase-change signal zl(t) is then output to a transmitter. [0 1 721 However, the phase changing rate applied by the phase changers 3 17A and 3 17B varies simultaneously in order to perform the phase changing shown in Fig. 26. 10 (The following describes a non-limiting example of the phase changing scheme.) For time u, phase changer 3 17A fiom Fig. 26 performs the change of phase such that zl (t) = yl(t)zl'(t), while phase changer 3 17B performs the change of phase such that d(t) = y2(t)z2'(t). For example, as shown in Fig. 26, for time u, yl(u) = and y2(u) = e-jd2, for time u+l, yl(u+l) = e'"I4 and y2(u+l) = e- j3d4 , and for time u+k, 15 yl(u+k) = e'kd4 and y2(u+k) = e'(k3d4-xn). Here, the regular phase changing period (cycle) may be the same for both phase changers 3 17A and 3 17B, or may vary for each. [0 1731 Also, as described above, a change of phase may be performed before 20 precoding is performed by the weighting unit. In such a case, the transmission device should be configured as illustrated in Fig. 27. When a change of phase is carried out on both modulated signals, each of the transmit signals is, for example, control information that includes information about the phase changing pattern. By obtaining the control information, the 25 reception device knows the phase changing scheme by which the transmission device regularly varies the change, i.e., the phase changing pattern, and is thus able to demodulate (decode) the signals correctly. [0 1 741 5 8 Next, variants of the sample configurations shown in Figs. 6 and 25 are described with reference to Figs. 28 and 29. Fig. 28 differs fiom Fig. 6 in the inclusion of phase change ONIOFF information 2800 and in that the change of phase is performed on only one of zl'(t) and z2'(t) (i.e., performed on one of zl'(t) 5 and z2'(t), which have identical time or a common fiequency). Accordingly, in order to perform the change of phase on one of zl'(t) and z2'(t), the phase changers 317A and 317B shown in Fig. 28 may each be ON, and performing the change of phase, or OFF, and not performing the change of phase. The phase change ONIOFF information 2800 is control information therefor. The phase change 10 ONIOFF information 2800 is output by the signal processing scheme information generator 3 14 shown in Fig. 3. [0 1751 Phase changer 317A of Fig. 28 changes the phase to produce zl(t) = yl(t)zl'(t), while phase changer 317B changes the phase to produce z2(t) = 15 y2(t)z2'(t). Here, a change of phase having a period (cycle) of four is, for example, applied to zl'(t). (Meanwhile, the phase of z2'(t) is not changed.) Accordingly, for time u, yl(u) = do and y2(u) = 1, for time u+l, yl(u+l) = dd2 and y2(u+l) = 1, for time u+2, yl(u+2) = e'" and y2(u+2) = 1, and for time u+3, yl(u+3) = e'3"n and 20 y2(u+3) = 1. [0 1761 Next, a change of phase having a period (cycle) of four is, for example, applied to z2'(t). (Meanwhile, the phase of zl'(t) is not changed.) Accordingly, for time u+4, yl(u+4) = 1 and y2(u+4) = do, for time u+5, yl(u+5) = 1 and y2(u+5) = dd2, 25 for time u+6, yl(u+6) = 1 and y2(u+6) = e'", and for time u+7, yl(u+7) = 1 and y2(u+7) = d3"n. [0 1771 Accordingly, given the above examples. 5 9 for any time 8k, y1(8k) = do and y2(8k) = 1, for iny time 8k+l, y1(8k+l) = Pn and y2(8k+l) = 1, for any time 8k+2, y1(8k+2) = e'" and y2(8k+2) = 1, for any time 8k+3, y1(8k+3) = d3"" and y2(8k+3) = 1, 5 for any time 8k+4, y1(8k+4) = 1 and y2(8k+4) = do, for any time 8k+5, y1(8k+3) = 1 and y2(8k+5) = p", for any time 8k+6, y1(8k+6) = 1 and y2(8k+6) = e'", and for any time 8k+7, y1(8k+7) = 1 and y2(8k+7) = e'3d2. [0178] 10 As described above, there are two intervals, one where the change of phase is performed on zl'(t) only, and one where the change of phase is performed on z2'(t) only. Furthermore, the two intervals form a phase changing period (cycle). While the above explanation describes the interval where the change of phase is performed on zl'(t) only and the interval where the change of phase is performed on 15 z2'(t) only as being equal, no limitation is intended in this manner. The two intervals may also differ. In addition, while the above explanation describes performing a change of phase having a period (cycle) of four on zl'(t) only and then performing a change of phase having a period (cycle) of four on z2'(t) only, no limitation is intended in this manner. The changes of phase may be performed on 20 zl'(t) and on z2'(t) in any order (e.g., the change of phase may alternate between being performed on zl'(t) and on z2'(t), or may be performed in random order). Phase changer 317A of Fig. 29 changes the phase to produce sl'(t) = yl(t)sl(t), while phase changer 3 17B changes the phase to produce s2'(t) = y2(t)s2(t). [0 1791 25 Here, a change of phase having a period (cycle) of four is, for example, applied to sl(t). (Meanwhile, s2(t) remains unchanged). Accordingly, for time u, yl(u) = do and y2(u) = 1, for time u+l, yl(u+l) = e'"" and y2(u+l) = 1, for time u+2, yl(u+2) = d" and y2(u+2) = 1, and for time u+3, yl(u+3) = e'3"/2 and y2(u+3) = 1. 60 [O 1 801 Next, a change of phase having a period (cycle) of four is, for example, applied to s2(t). (Meanwhile, sl(t) remains unchanged). Accordingly, for time u+4, yl(u+4) = 1 and y2(u+4) = d', for time u+5, yl(u+5) = 1 and y2(u+5) = Pn, for time 5 u+6, yl(u+6) = 1 and y2(u+6) = P, and for time u+7, yl(u+7) = 1 and y2(u+7) = e'3" [0181] Accordingly, given the above examples, for any time 8k, y1(8k) = e' and y2(8k) = 1, 10 for any time 8k+l, y1(8k+l) = d"" and y2(8k+l) = 1, for any time 8k+2, y1(8k+2) = P and y2(8k+2) = 1, for any time 8k+3, y1(8k+3) = d3"" and y2(8k+3) = 1, for any time 8k+4, y1(8k+4) = 1 and y2(8k+4) = do, for any time 8k+5, y1(8k+5) = 1 and y2(8k+5) = Pn, 15 for any time 8k+6, y1(8k+6) = 1 and y2(8k+6) = e'", and for any time 8k+7, y1(8k+7) = 1 and y2(8k+7) = e'3d2. [0 1 821 As described above, there are two intervals, one where the change of phase is performed on sl(t) only, and one where the change of phase is performed on s2(t) 20 only. Furthermore, the two intervals form a phase changing period (cycle). Although the above explanation describes the interval where the change of phase is performed on sl(t) only and the interval where the change of phase is performed on s2(t) only as being equal, no limitation is intended in this manner. The two intervals may also differ. In addition, while the above explanation describes 25 performing the change of phase having a period (cycle) of four on sl(t) only and then performing the change of phase having a period (cycle) of four on s2(t) only, no limitation is intended in this manner. The changes of phase may be performed on sl(t) and on s2(t) in any order (e.g., may alternate between being performed on sl(t) and on s2(t), or may be performed in random order). Accordingly, the reception conditions under which the reception device receives each transmit signal zl(t) and z2(t) are equalized. By periodically 5 switching the phase of the symbols in the received signals zl(t) and z2(t), the ability of the error corrected codes to correct errors may be improved, thus ameliorating received signal quality in the LOS environment. [0 1 831 Accordingly, Embodiment 2 as described above is able to produce the same 10 results as the previously described Embodiment 1. Although the present Embodiment used a single-carrier scheme, i.e., time domain phase changing, as an example, no limitation is intended in this regard. The same effects are also achievable using multi-carrier transmission. Accordingly, the present Embodiment may also be realized using, for example, spread-spectrum 15 communications, OFDM, SC-FDMA (Single Carrier Frequency-Division Multiple Access), SC-OFDM, wavelet OFDM as described in Non-Patent Literature 7, and so on. As previously described, while the present Embodiment explains the change of phase as changing the phase with respect to the time domain t, the phase may alternatively be changed with respect to the fkequency domain as described in 20 Embodiment 1. That is, considering the phase changing scheme in the time domain t described in the present Embodiment and replacing t with f (f being the ((sub-) carrier) fkequency) leads to a change of phase applicable to the frequency domain. Also, as explained above for Embodiment 1, the phase changing scheme of the present Embodiment is also applicable to changing the phase with respect 25 both the time domain and the frequency domain. [0 1 841 Accordingly, although Figs. 6, 25, 26, and 27 illustrate changes of phase in the time domain, replacing time t with carrier f in each of Figs. 6, 25, 26, and 27 62 corresponds to a change of phase in the frequency domain. In other words, replacing (t) with (t, f ) where t is time and f is frequency corresponds to performing the change of phase on time-frequency blocks. [0185] 5 Furthermore, in the present Embodiment, symbols other than data symbols, such as pilot symbols (preamble, unique word, etc) or symbols transmitting control information, may be arranged within the frame in any manner. [Embodiment 31 Embodiments 1 and 2, described above, discuss regular changes of phase. 10 Embodiment 3 describes a scheme of allowing the reception device to obtain good received signal quality for data, regardless of the reception device arrangement, by considering the location of the reception device with respect to the transmission device. [0186] 15 Embodiment 3 concerns the symbol arrangement within signals obtained through a change of phase. Fig. 31 illustrates an example of frame configuration for a portion of the symbols within a signal in the time-frequency domain, given a transmission scheme where a regular change of phase is performed for a multi-carrier scheme such as 20 OFDM. First, an example is explained in which the change of phase is performed one of two baseband signals, precoded as explained in Embodiment 1 (see Fig. 6). [0 1 871 (Although Fig. 6 illustrates a change of phase in the time domain, switching 25 time t with carrier f in Fig. 6 corresponds to a change of phase in the fiequency domain. In other words, replacing (t) with (t, f) where t is time and f is fiequency corresponds to performing phase changes on time-frequency blocks.) Fig. 31 illustrates the frame configuration of modulated signal z2', which is input to phase changer 317B from Fig. 12. Each square represents one symbol (although both signals sl and s2 are included for precoding purposes, depending on the precoding matrix, only one of signals sl and s2 may be used). 5 [0188] Consider symbol 3 100 at carrier 2 and time $2 of Fig. 3 I. The carrier here described may alternatively be termed a sub-carrier. Within camer 2, there is a very strong correlation between the channel conditions for symbol 3100 at carrier 2, time $2 and the channel conditions for the 10 time domain nearest-neighbour symbols to time $2, i.e., symbol 3013 at time $1 and symbol 3 101 at time $3 within carrier 2. [0189] Similarly, for time $2, there is a very strong correlation between the channel conditions for symbol 3100 at carrier 2, time $2 and the channel conditions for the 15 frequency-domain nearest-neighbour symbols to carrier 2, i.e., symbol 3 104 at carrier 1, time $2 and symbol 3 104 at time $2, carrier 3. As described above, there is a very strong correlation between the channel conditions for symbol 3 100 and the channel conditions for symbols 3 101, 3 102, 3103, and 3104. 20 [0190] The present description considers N different phases (N being an integer, N 3 2) for multiplication in a transmission scheme where the phase is regularly changed. The symbols illustrated in Fig. 31 are indicated as @, for example. This signifies that this symbol is signal 22' from Fig. 6 phase-changed through 25 multiplication by go. That is, the values indicated in Fig. 31 for each of the symbols are the values of y(t) from formula 42, which are also the values of z2(t) = y2(t)z2'(t) described in Embodiment 2. [0191] 64 The present Embodiment takes advantage of the high correlation in channel conditions existing between neigbouring symbols in the frequency domain andlor neighbouring symbols in the time domain in a symbol arrangement enabling high data reception quality to be obtained by the reception device receiving the 5 phase-changed symbols. In order to achieve this high data reception quality, conditions #1 and #2 are necessary. [0 1 921 (Condition # 1) 10 As shown in Fig. 6, for a transmission scheme involving a regular change of phase performed on precoded baseband signal 22' using multi-carrier transmission such as OFDM, time X, carrier Y is a symbol for transmitting data (hereinafter, data symbol), neighbouring symbols in the time domain, i.e., at time X-1, carrier Y and at time X+1, carrier Y are also data symbols, and a different change of phase should 15 be performed on precoded baseband signal 22' corresponding to each of these three data symbols, i-e., on precoded baseband signal 22' at time X, carrier Y, at time X-1, carrier Y and at time X+l, carrier Y. [0 1 931 (Condition #2) 20 As shown in Fig. 6, for a transmission scheme involving a regular change of phase performed on precoded baseband signal 22' using multi-carrier transmission such as OFDM, time X, carrier Y is a data symbol, neighbouring symbols in the fi-equency domain, i.e., at time X, carrier Y-1 and at time X, carrier Y+l are also data symbols, and a different change of phase should be performed on precoded 25 baseband signal 22' corresponding to each of these three data symbols, i.e., on precoded baseband signal 22' at time X, carrier Y, at time X, carrier Y-1 and at time X, carrier Y+l. [0 1 941 Ideally, data symbols satisfling Condition #1 should be present. Similarly, data symbols satisfling Condition #2 should be present. The reasons supporting Conditions #1 and #2 are as follows. A very strong correlation exists between the channel conditions of given 5 symbol of a transmit signal (hereinafter, symbol A) and the channel conditions of the symbols neighbouring symbol A in the time domain, as described above. [0 1951 Accordingly, when three neighbouring symbols in the time domain each have different phases, then despite reception quality degradation in the LOS 10 environment (poor signal quality caused by degradation in conditions due to direct wave phase relationships despite high signal quality in terms of SNR) for symbol A, the two remaining symbols neighbouring symbol A are highly likely to provide good reception quality. As a result, good received signal quality is achievable after error correction and decoding. 15 [0196] Similarly, a very strong correlation exists between the channel conditions of given symbol of a transmit signal (hereinafter, symbol A) and the channel conditions of the symbols neighbouring symbol A in the fiequency domain, as described above. Accordingly, when three neighbouring symbols in the fiequency domain 20 each have different phases, then despite reception quality degradation in the LOS environment (poor signal quality caused by degradation in conditions due to direct wave phase relationships despite high signal quality in terms of SNR) for symbol A, the two remaining symbols neighbouring symbol A are highly likely to provide good reception quality. As a result, good received signal quality is achievable after error 25 correction and decoding. [0197] Combining Conditions #1 and #2, ever greater data reception quality is likely achievable for the reception device. Accordingly, the following Condition #3 can be derived. (Condition #3) 5 As shown in Fig. 6, for a transmission scheme involving a regular change of phase performed on precoded baseband signal 22' using multi-carrier transmission such as OFDM, time X, carrier Y is a data symbol, neighbouring symbols in the time domain, i.e., at time X-I, carrier Y and at time X+1, carrier Y are also data symbols, and neighbouring symbols in the frequency domain, i.e., at time X, carrier 10 Y-1 and at time X, carrier Y+1 are also data symbols, and a different change in phase should be performed on precoded baseband signal 22' corresponding to each of these five data symbols, i.e., on precoded baseband signal 22' at time X, carrier Y, at time X, carrier Y-I, at time X, carrier Y+1, at a time X-I, carrier Y, and at time X+1, carrier Y. 15 [0198] Here, the different changes in phase are as follows. Changes in phase are defined from 0 radians to 2n radians. For example, for time X, carrier Y, a phase change of e'OxY is applied to precoded baseband signal 22' from Fig. 6, for time X-I, carrier Y, a phase change of e' 'OX-1,Y is applied to precoded baseband signal 22' from 20 Fig. 6, for time X+1, carrier Y, a phase change of e'ex+'.Yis applied to precoded baseband signal 22' from Fig. 6, such that 0 5 €IxY < 271, 0 5 OX-I,Y < 2n, and 0 5 < 2n, all units being in radians. Accordingly, for Condition #I, it follows that OYy # OX-^,^, €IYY #OX+I,Y, and that OX-l,Y # OX+l,Y. Similarly, for Condition #2, it follows that OYY # OYY-I, OYY # OYY+I, and that OxY-I # OYY+l. And, for 25 Condition #3, it follows that # OX-l,Y, OxY# OX+I,Y, €IxY# OxY-I, €IxY # OxY-I, OX-I,Y # ~ x + I , Y , ~ x - I , Y # ~ Y Y - I , OX-I,Y # ~x+I,Y~ X, + I , Y# OX-I,Y~, X + I , Y# OYy+1, and that ~YY-#l exy+1- [0 1991 67 Ideally, a data symbol should satisfj Condition #3. Fig. 31 illustrates an example of Condition #3 where symbol A corresponds to symbol 3100. The symbols are arranged such that the phase by which precoded baseband signal 22' fiom Fig. 6 is multiplied differs for symbol 3100, for both 5 neighbouring symbols thereof in the time domain 3101 and 3102, and for both neighbouring symbols thereof in the fiequency domain 3102 and 3104. Accordingly, despite received signal quality degradation of symbol 3100 for the receiver, good signal quality is highly likely for the neighbouring signals, thus guaranteeing good signal quality after error correction. 10 [0200] Fig. 32 illustrates a symbol arrangement obtained through phase changes under these conditions. As evident from Fig. 32, with respect to any data symbol, a different change in phase is applied to each neighbouring symbol in the time domain and in the 15 fiequency domain. As such, the ability of the reception device to correct errors may be improved. [020 11 In other words, in Fig. 32, when all neighbouring symbols in the time domain are data symbols, Condition #1 is satisfied for all Xs and all Ys. 20 Similarly, in Fig. 32, when all neighbouring symbols in the fiequency domain are data symbols, Condition #2 is satisfied for all Xs and all Ys. Similarly, in Fig. 32, when all neighbouring symbols in the fiequency domain are data symbols and all neighbouring symbols in the time domain are data symbols, Condition #3 is satisfied for all Xs and all Ys. 25 [0202] The following describes an example in which a change of phase is performed on two precoded baseband signals, as explained in Embodiment 2 (see Fig. 26). 68 When a change of phase is performed on precoded baseband signal zl' and precoded baseband signal 22' as shown in Fig. 26, several phase changing schemes are possible. The details thereof are explained below. [0203] 5 Scheme 1 involves a change in phase performed on precoded baseband signal 22' as described above, to achieve the change in phase illustrated by Fig. 32. In Fig. 32, a change of phase having a period (cycle) of 10 is applied to precoded baseband signal 22'. However, as described above, in order to satisfl Conditions #1, #2, and #3, the change in phase applied to precoded baseband signal 22' at each 10 (sub-)carrier varies over time. (Although such changes are applied in Fig. 32 with a period (cycle) of ten, other phase changing schemes are also possible.) Then, as shown in Fig. 33, the change in phase performed on precoded baseband signal zl' produces a constant value that is one-tenth of that of the change in phase performed on precoded baseband signal 22'. In Fig. 33, for a period (cycle) (of change in 15 phase performed on precoded baseband signal 22') including time $1, the value of the change in phase performed on precoded baseband signal zl' is do. Then, for the next period (cycle) (of change in phase performed on precoded baseband signal 22') including time $2, the value of the change in phase performed on precoded baseband signal zl' is dd9, and so on. 20 [0204] The symbols illustrated in Fig. 33 are indicated as do, for example. This signifies that this symbol is signal zl' from Fig. 26 on which a change in phase as been applied through multiplication by do. That is, the values indicated in Fig. 33 for each of the symbols are the values of zl'(t) = y2(t)zl'(t) described in 25 Embodiment 2 for yl(t). [0205] As shown in Fig. 33, the change in phase performed on precoded baseband signal zl' produces a constant value that is one-tenth that of the change in phase 69 performed on precoded baseband signal 22' such that the phase changing value varies with the number of each period (cycle). (As described above, in Fig. 33, the value is do for the first period (cycle), dnD for the second period (cycle), and so on.) As described above, the change in phase performed on precoded baseband 5 signal 22' has a period (cycle) of ten, but the period (cycle) can be effectively made greater than ten by taking the change in phase applied to precoded baseband signal zl' and to precoded baseband signal 22' into consideration. Accordingly, data reception quality may be improved for the reception device. [0206] 10 Scheme 2 involves a change in phase of precoded baseband signal 22' as described above, to achieve the change in phase illustrated by Fig. 32. In Fig. 32, a change of phase having a period (cycle) of ten is applied to precoded baseband signal 22'. However, as described above, in order to satisfy Conditions #1, #2, and #3, the change in phase applied to precoded baseband signal 22' at each (sub-)carrier 15 varies over time. (Although such changes are applied in Fig. 32 with a period (cycle) of ten, other phase changing schemes are also possible.) Then, as shown in Fig. 30, the change in phase performed on precoded baseband signal zl' differs from that performed on precoded baseband signal 22' in having a period (cycle) of three rather than ten. 20 [0207] The symbols illustrated in Fig. 30 are indicated as e", for example. This signifies that this symbol is signal zl' from Fig. 26 to which a change in phase has been applied through multiplication by do. That is, the values indicated in Fig. 30 for each of the symbols are the values of zl(t) = yl(t)zl'(t) described in Embodiment 25 2foryl(t). [020 81 As described above, the change in phase performed on precoded baseband signal 22' has a period (cycle) of ten, but by taking the changes in phase applied to 70 precoded baseband signal zl' and precoded baseband signal 22' into consideration, the period (cycle) can be effectively made equivalent to 30 for both precoded baseband signals zl' and 22'. Accordingly, data reception quality may be improved for the reception device. An effective way of applying scheme 2 is to 5 perform a change in phase on precoded baseband signal zl' with a period (cycle) of N and perform a change in phase on precoded baseband signal 22' with a period (cycle) of M such that N and M are coprime. As such, by taking both precoded baseband signals zl' and 22' into consideration, a period (cycle) of NxM is easily achievable, effectively making the period (cycle) greater when N and M are 10 coprime. [0209] The above describes an example of the phase changing scheme pertaining to Embodiment 3. The present invention is not limited in this manner. As explained for Embodiments 1 and 2, a change in phase may be performed with respect the 15 frequency domain or the time domain, or on time-frequency blocks. Similar improvement to the data reception quality can be obtained for the reception device in all cases. The same also applies to h e s having a configuration other than that described above, where pilot symbols (SP (Scattered Pilot) and symbols transmitting 20 control information are inserted among the data symbols. The details of change in phase in such circumstances are as follows. [02 lo] Figs. 47A and 47B illustrate the frame configuration of modulated signals (precoded baseband signals) zl or zl' and 22' in the time-frequency domain. Fig. 25 47A illustrates the frame configuration of modulated signal (precoded baseband signals) zl or zl' while Fig. 47B illustrates the fiarne configuration of modulated signal (precoded baseband signals) 22'. In Figs. 47A and 47B, 4701 marks pilot symbols while 4702 marks data symbols. The data symbols 4702 are symbols on which precoding or precoding and a change in phase have been performed. [02 1 11 Figs. 47A and 47B, like Fig. 6, indicate the arrangement of symbols when a 5 change in phase is applied to precoded baseband signal 22' (while no change of phase is performed on precoded baseband signal zl). (Although Fig. 6 illustrates a change in phase with respect to the time domain, switching time t with carrier f in Fig. 6 corresponds to a change in phase with respect to the frequency domain. In other words, replacing (t) with (t, f) where t is time and f is frequency corresponds to 10 performing a change of phase on time-frequency blocks.) Accordingly, the numerical values indicated in Figs. 47A and 47B for each of the symbols are the values of precoded baseband signal 22' after the change in phase. No values are given for the symbols of precoded baseband signal zl' (21) as no change in phase is performed thereon. 15 [0212] The key point of Figs. 47A and 47B is that the change in phase is performed on the data symbols of precoded baseband signal z2', i.e., on precoded symbols. (The symbols under discussion, being precoded, actually include both symbols sl and s2.) Accordingly, no change of phase is performed on the pilot symbols inserted 20 into 22'. [02 1 31 Figs. 48A and 48B illustrate the fiame configuration of modulated signals (precoded baseband signals) zl or zl' and 22' in the time-frequency domain. Fig. 48A illustrates the m e configuration of modulated signal (precoded baseband 25 signals) zl or zl' while Fig. 47B illustrates the fiame configuration of modulated signal (precoded baseband signals) 22'. In Figs. 48A and 48B, 4701 marks pilot symbols while 4702 marks data symbols. The data symbols 4702 are symbols on which precoding, or precoding and a change in phase, have been performed. 72 [02 141 Figs. 48A and 48B, like Fig. 26, indicate the arrangement of symbols when a change in phase is applied to precoded baseband signal zl' and to precoded baseband signal 22'. (Although Fig. 26 illustrates a change in phase with respect to 5 the time domain, switching time t with carrier f in Fig. 26 corresponds to a change in phase with respect to the frequency domain. In other words, replacing (t) with (t, ij where t is time and f is frequency corresponds to performing a change of phase on time-frequency blocks.) Accordingly, the numerical values indicated in Figs. 48A and 48B for each of the symbols are the values of precoded baseband signal zl' and 10 22' after the change in phase. [02 1 51 The key point of Fig. 47 is that a change of phase is performed on the data symbols of precoded baseband signal zl', that is, on the precoded symbols thereof, and on the data symbols of precoded baseband signal 22', that is, on the precoded 15 symbols thereof. (The symbols under discussion, being precoded, actually include both symbols sl and s2.) Accordingly, no change of phase is performed on the pilot symbols inserted in zl', nor on the pilot symbols inserted in 22'. [02 1 61 Figs. 49A and 49B illustrate the frame configuration of modulated signals 20 (precoded baseband signals) zl or zl' and 22' in the time-frequency domain. Fig. 49A illustrates the frame configuration of modulated signal (precoded baseband signals) zl or zl' while Fig. 49B illustrates the fiarne configuration of modulated signal (precoded baseband signal) 22'. In Figs. 49A and 49B, 4701 marks pilot symbols, 4702 marks data symbols, and 4901 marks null symbols for which the 25 in-phase component of the baseband signal I = 0 and the quadrature component Q = 0. As such, data symbols 4702 are symbols on which precoding or precoding and the change in phase have been performed. Figs. 49A and 49B differ from Figs. 47A and 47B in the configuration scheme for symbols other than data symbols. 73 The times and carriers at which pilot symbols are inserted into modulated signal zl' are null symbols in modulated signal 22'. Conversely, the times and carriers at which pilot symbols are inserted into modulated signal 22' are null symbols in modulated signal zl '. 5 [0217] Figs. 49A and 49B, like Fig. 6, indicate the arrangement of symbols when a change in phase is applied to precoded baseband signal 22' (while no change of phase is performed on precoded baseband signal zl). (Although Fig. 6 illustrates a change of phase with respect to the time domain, switching time t with carrier f in 10 Fig. 6 corresponds to a change of phase with respect to the frequency domain. In other words, replacing (t) with (t, f ) where t is time and f is frequency corresponds to performing a change of phase on time-frequency blocks.) Accordingly, the numerical values indicated in Figs. 49A and 49B for each of the symbols are the values of precoded baseband signal 22' after a change of phase is performed. No 15 values are given for the symbols of precoded baseband signal zl' (zl) as no change of phase is performed thereon. [02 1 81 The key point of Figs. 49A and 49B is that a change of phase is performed on the data symbols of precoded baseband signal 22'' i.e., on precoded symbols. 20 (The symbols under discussion, being precoded, actually include both symbols sl and s2.) Accordingly, no change of phase is performed on the pilot symbols inserted into 22'. [02 1 91 Figs. 50A and 50B illustrate the frame configuration of modulated signals 25 (precoded baseband signals) zl or zl' and 22' in the time-frequency domain. Fig. 50A illustrates the frame configuration of modulated signal (precoded baseband signal) zl or zl' while Fig. 50B illustrates the frame configuration of modulated signal (precoded baseband signal) 22'. In Figs. 50A and 50B, 4701 marks pilot 74 symbols, 4702 marks data symbols, and 4901 marks null symbols for which the in-phase component of the baseband signal I = 0 and the quadrature component Q = 0. As such, data symbols 4702 are symbols on which precoding, or precoding and a change of phase, have been performed. Figs. 50A and 50B differ from Figs. 48A 5 and 48B in the configuration scheme for symbols other than data symbols. The times and carriers at which pilot symbols are inserted into modulated signal zl' are null symbols in modulated signal 22'. Conversely, the times and carriers at which pilot symbols are inserted into modulated signal 22' are null symbols in modulated signal zl ' . 10 [0220] Figs. 50A and 50B, like Fig. 26, indicate the arrangement of symbols when a change of phase is applied to precoded baseband signal zl' and to precoded baseband signal 22'. (Although Fig. 26 illustrates a change of phase with respect to the time domain, switching time t with carrier f in Fig. 26 corresponds to a change of 15 phase with respect to the frequency domain. In other words, replacing (t) with (t, f) where t is time and f is frequency corresponds to performing a change of phase on time-frequency blocks.) Accordingly, the numerical values indicated in Figs. 50A and 50B for each of the symbols are the values of precoded baseband signal zl' and 22' after a change of phase. 20 [0221] The key point of Figs. 50A and 50B is that a change of phase is performed on the data symbols of precoded baseband signal zl', that is, on the precoded symbols thereof, and on the data symbols of precoded baseband signal 22', that is, on the precoded symbols thereof. (The symbols under discussion, being precoded, 25 actually include both symbols sl and s2.) Accordingly, no change of phase is performed on the pilot symbols inserted in zl', nor on the pilot symbols inserted in 22'. [0222] Fig. 51 illustrates a sample configuration of a transmission device generating and transmitting modulated signal having the frame configuration of Figs. 47A, 47B, 49A, and 49B. Components thereof performing the same operations as those of Fig. 4 use the same reference symbols thereas. 5 In Fig. 51, the weighting units 308A and 308B and phase changer 317B only operate at times indicated by the frame configuration signal 313 as corresponding to data symbols. [0223] In Fig. 5 1, a pilot symbol generator 5 101 (that also generates null symbols) 10 outputs baseband signals 5 102A and 5 102B for a pilot symbol whenever the frame configuration signal 3 13 indicates a pilot symbol (or a null symbol). Although not indicated in the frame configurations from Figs. 47A through 50B, when precoding (or phase rotation) is not performed, such as when transmitting a modulated signal using only one antenna (such that the other antenna transmits no 15 signal) or when using a space-time coding transmission scheme (particularly, space-time block coding) to transmit control information symbols, then the frame configuration signal 313 takes control information symbols 5104 and control information 5103 as input. When the frame configuration signal 3 13 indicates a control information symbol, baseband signals 5 102A and 5 102B thereof are output. 20 [0224] Wireless units 310A and 310B of Fig. 51 take a plurality of baseband signals as input and select a desired baseband signal according to the frame configuration signal 3 13. Wireless units 310A and 310B then apply OFDM signal processing and output modulated signals 31 1A and 31 1B conforming to the fiame 25 configuration. Fig. 52 illustrates a sample configuration of a transmission device generating and transmitting modulated signal having the fi-ame configuration of Figs. 48A, 48B, 50A, and 50B. Components thereof performing the same operations as 76 those of Figs. 4 and 51 use the same reference symbols thereas. Fig. 51 features an additional phase changer 317A that only operates when the h e configuration signal 3 13 indicates a data symbol. At all other times, the operations are identical to those explained for Fig. 5 1. 5 [0225] Fig. 53 illustrates a sample configuration of a transmission device that differs fiom that of Fig. 51. The following describes the points of difference. As shown in Fig. 53, phase changer 317B takes a plurality of baseband signals as input. Then, when the frame configuration signal 313 indicates a data symbol, phase 10 changer 3 17B performs a change of phase on precoded baseband signal 3 16B. When kame configuration signal 3 13 indicates a pilot symbol (or null symbol) or a control information symbol, phase changer 3 17B pauses phase changing operations, such that the symbols of the baseband signal are output as-is. (This may be interpreted as performing forced rotation corresponding to go.) 15 A selector 5301 takes the plurality of baseband signals as input and selects a baseband signal having a symbol indicated by the fiame configuration signal 3 13 for output. [0226] Fig. 54 illustrates a sample configuration of a transmission device that 20 differs from that of Fig. 52. The following describes the points of difference. As shown in Fig. 54, phase changer 3 17B takes a plurality of baseband signals as input. Then, when the frame configuration signal 313 indicates a data symbol, phase changer 317B performs a change of phase on precoded baseband signal 316B. When frame configuration signal 3 13 indicates a pilot symbol (or null symbol) or a 25 control information symbol, phase changer 3 17B pauses phase changing operations such that the symbols of the baseband signal are output as-is. (This may be interpreted as performing forced rotation corresponding to do.) Similarly, as shown in Fig. 54, phase changer 5201 takes a plurality of baseband signals as input. Then, when the fiame configuration signal 3 13 indicates a data symbol, phase changer 5201 performs a change of phase on precoded baseband signal 309A. When fiame configuration signal 313 indicates a pilot 5 symbol (or null symbol) or a control information symbol, phase changer 5201 pauses phase changing operations such that the symbols of the baseband signal are output as-is. (This may be interpreted as performing forced rotation corresponding to do.) The above explanations are given using pilot symbols, control symbols, and 10 data symbols as examples. However, the present invention is not limited in this manner. When symbols are transmitted using schemes other than precoding, such as single-antenna transmission or transmission using space-time block coding, not performing a change of phase is important. Conversely, performing a change of phase on symbols that have been precoded is the key point of the present invention. 15 [0227] Accordingly, a characteristic feature of the present invention is that the change of phase is not performed on all symbols within the frame configuration in the time-frequency domain, but only performed on signals that have been precoded. [Embodiment 41 20 Embodiments 1 and 2, described above, discuss a regular change of phase. Embodiment 3, however, discloses performing a different change of phase on neighbouring symbols. [0228] The present Embodiment describes a phase changing scheme that varies 25 according to the modulation scheme and the coding rate of the error-correcting codes used by the transmission device. Table 1, below, is a list of phase changing scheme settings corresponding to the settings and parameters of the transmission device. 78 [0229] [Table 11 No. of Modulated Transmission Signals 2 2 2 2 2 2 2 2 2 2 2 . . Modulation Scheme #1 :QPSK, #2: QPSK #1 :QPSK, #2: QPSK #I :QPSK, #2: QPSK # 1 :QPSK, #2: QPSK # 1 :QPSK, #2: QPSK #l:QPSK,#2:16-QAM # 1 : QPSK, #2: 16-QAM #1: QPSK, #2: 16-QAM # 1 : QPSK, #2: 16-QAM # 1 : QPSK, #2: 16-QAM # 1 : 16-QAM, #2: 16-QAM Coding Rate #1: 112, #2 213 #1: 112, #2: 314 #1: 213, #2: 315 # 1 : 213, #2: 213 # 1 : 313, #2: 213 #1:112,#2: 213 # 1 : 112, #2: 314 #1: 112, #2: 315 # 1 : 213, #2: 314 # 1 : 213, #2: 516 #1: 112, #2: 213 Phase Changing Pattern # 1 : -, #2:A #1: A, #2: B #I: A, #2: C #1: C,#2: - #1: D, #2: E #1: B, #2: A #1: A, #2: C #1: -, #2:E #I: D, #2: - #I : D, #2: B #1: -, #2:E [023 01 In Table 1, #1 denotes modulated signal sl fiom Embodiment 1 described above (baseband signal sl modulated with the modulation scheme set by the 5 transmission device) and #2 denotes modulated signal s2 (baseband signal s2 modulated with the modulation scheme set by the transmission device). The coding rate column of Table 1 indicates the coding rate of the error-correcting codes for modulation schemes #1 and #2. The phase changing pattern column of Table 1 indicates the phase changing scheme applied to precoded baseband signals zl (zl') 10 and 22 (22'), as explained in Embodiments 1 through 3. Although the phase changing patterns are labeled A, B, C, D, E, and so on, this refers to the phase change degree applied, for example, in a phase changing pattern given by formula 46 and formula 47, above. In the phase changing pattern column of Table 1, the dash signifies that no change of phase is applied. 15 [0231] The combinations of modulation scheme and coding rate listed in Table 1 are examples. Other modulation schemes (such as 128-QAM and 256-QAM) and coding rates (such as 718) not listed in Table 1 may also be included. Also, as described in Embodiment 1, the error-correcting codes used for sl and s2 may differ 20 (Table 1 is given for cases where a single type of error-correcting codes is used, as in Fig. 4). Furthermore, the same modulation scheme and coding rate may be used with different phase changing patterns. The transmission device transmits information indicating the phase changing patterns to the reception device. The reception device specifies the phase changing pattern by cross-referencing the 25 information and Table 1, then performs demodulation and decoding. When the modulation scheme and error-correction scheme determine a unique phase changing pattern, then as long as the transmission device transmits the modulation scheme and information regarding the error-correction scheme, the reception device knows the 80 phase changing pattern by obtaining that information. As such, information pertaining to the phase changing pattern is not strictly necessary. [0232] In Embodiments 1 through 3, the change of phase is applied to precoded 5 baseband signals. However, the amplitude may also be modified along with the phase in order to apply periodical, regular changes. Accordingly, an amplification modification pattern regularly modifling the amplitude of the modulated signals may also be made to conform to Table 1. In such circumstances, the transmission device should include an amplification modifier that modifies the amplification after 10 weighting unit 308A or weighting unit 308B from Fig. 3 or 4. In addition, amplification modification may be performed on only one of or on both of the precoded baseband signals zl(t) and z2(t) (in the former case, the amplification modifier is only needed after one of weighting unit 308A and 308B). [023 31 Furthermore, although not indicated in Table 1 above, the mapping scheme may also be regularly modified by the mapper, without a regular change of phase. That is, when the mapping scheme for modulated signal sl(t) is 16-QAM and the mapping scheme for modulated signal s2(t) is also 16-QAM, the mapping scheme applied to modulated signal s2(t) may be regularly changed as follows: from 20 16-QAM to 16-APSK, to 16-QAM in the IQ plane, to a first mapping scheme producing a signal point layout unlike 16-APSK, to 16-QAM in the IQ plane, to a second mapping scheme producing a signal point layout unlike 16-APSK, and so on. As such, the data reception quality can be improved for the reception device, much like the results obtained by a regular change of phase described above. 25 [0234] In addition, the present invention may use any combination of schemes for a regular change of phase, mapping scheme, and amplitude, and the transmit signal may transmit with all of these taken into consideration. 8 1 The present Embodiment may be realized using single-carrier schemes as well as multi-carrier schemes. Accordingly, the present Embodiment may also be realized using, for example, spread-spectrum communications, OFDM, SC-FDMA, SC-OFDM, wavelet OFDM as described in Non-Patent Literature 7, and so on. As 5 described above, the present Embodiment describes changing the phase, amplitude, and mapping schemes by performing phase, amplitude, and mapping scheme modifications with respect to the time domain t. However, much like Embodiment 1, the same changes may be carried out with respect to the frequency domain. That is, considering the phase, amplitude, and mapping scheme modification in the time 10 domain t described in the present Embodiment and replacing t with f (f being the ((sub-) carrier) frequency) leads to phase, amplitude, and mapping scheme modification applicable to the frequency domain. Also, the phase, amplitude, and mapping scheme modification of the present Embodiment is also applicable to phase, amplitude, and mapping scheme modification in both the time domain and the 1 5 frequency domain. [0235] Furthermore, in the present Embodiment, symbols other than data symbols, such as pilot symbols (preamble, unique word, etc) or symbols transmitting control information, may be arranged within the frame in any manner. 20 [Embodiment A 1 ] The present Embodiment describes a scheme for regularly changing the phase when encoding is performed using block codes as described in Non-Patent Literature 12 through 15, such as QC (Quasi-Cyclic) LDPC Codes (not only 25 QC-LDPC but also LDPC codes may be used), concatenated LDPC and BCH (Bose-Chaudhuri-Hocquenghem) codes, Turbo codes or Duo-Binary Turbo Codes using tail-biting, and so on. The following example considers a case where two streams sl and s2 are transmitted. However, when encoding has been performed 82 using block codes and control information and the like is not required, the number of bits making up each coded block matches the number of bits making up each block code (control information and so on described below may yet be included). When encoding has been performed using block codes or the like and control information 5 or the like (e.g., CRC (cyclic redundancy check) transmission parameters) is required, then the number of bits making up each coded block is the sum of the number of bits making up the block codes and the number of bits making up the information. [023 61 10 Fig. 34 illustrates the varying numbers of symbols and slots needed in each coded block when block codes are used. Fig. 34 illustrates the varying numbers of symbols and slots needed in each coded block when block codes are used when, for example, two streams sl and s2 are transmitted as indicated by the transmission device fiom Fig. 4, and the transmission device has only one encoder. (Here, the 15 transmission scheme may be any single-carrier scheme or multi-carrier scheme such as OFDM.) As shown in Fig. 34, when block codes are used, there are 6000 bits making up a single coded block. In order to transmit these 6000 bits, the number of required symbols depends on the modulation scheme, being 3000 symbols for QPSK, 20 1500 symbols for 16-QAM, and 1000 symbols for 64-QAM. [023 71 Then, given that the transmission device fiom Fig. 4 transmits two streams simultaneously, 1500 of the aforementioned 3000 symbols needed when the modulation scheme is QPSK are assigned to sl and the other 1500 symbols are 25 assigned to s2. As such, 1500 slots for transmitting the 1500 symbols (hereinafter, slots) are required for each of sl and s2. [023 81 By the same reasoning, when the modulation scheme is 16-QAM, 750 slots are needed to transmit all of the bits making up a single coded block, and when the modulation scheme is 64-QAM, 500 slots are needed to transmit all of the bits making up a single coded block. 5 The following describes the relationship between the above-defined slots and the phase of multiplication, as pertains to schemes for a regular change of phase. [023 91 Here, five different phase changing values (or phase changing sets) are assumed as having been prepared for use in the scheme for a regular change of phase. 10 That is, five different phase changing values (or phase changing sets) have been prepared for the phase changer of the transmission device from Fig. 4 (equivalent to the period (cycle) from Embodiments 1 through 4) (As in Fig. 6, five phase changing values are needed in order to perform a change of phase with a period (cycle) of five on precoded baseband signal 22' only. Also, as in Fig. 26, two 15 phase changing values are needed for each slot in order to perform the change of phase on both precoded baseband signals zl' and 22'. These two phase changing values are termed a phase changing set. Accordingly, five phase changing sets should ideally be prepared in order to perform the change of phase with a period (cycle) of five in such circumstances). These five phase changing values (or phase 20 changing sets) are expressed as PHASE[O], PHASE[l], PHASE[2], PHASE[3], and PHASE[4]. [0240] For the above-described 1500 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is QPSK, PHASE[O] is used 25 on 300 slots, PHASE[l] is used on 300 slots, PHASE[2] is used on 300 slots, PHASE[3] is used on 300 slots, and PHASE[4] is used on 300 slots. This is due to the fact that any bias in phase usage causes great influence to be exerted by the more frequently used phase, and that the reception device is dependent on such influence for data reception quality. [024 11 Similarly, for the above-described 700 slots needed to transmit the 6000 bits 5 making up a single coded block when the modulation scheme is 16-QAM, PHASE[O] is used on 150 slots, PHASE[l] is used on 150 slots, PHASE[2] is used on 150 slots, PHASE[3] is used on 150 slots, and PHASE[4] is used on 150 slots. [0242] Furthermore, for the above-described 500 slots needed to transmit the 6000 10 bits making up a single coded block when the modulation scheme is 64-QAM, PHASE[O] is used on 100 slots, PHASE[l] is used on 100 slots, PHASE[2] is used on 100 slots, PHASE[3] is used on 100 slots, and PHASE[4] is used on 100 slots. [0243] As described above, a scheme for a regular change of phase requires the 15 preparation of N phase changing values (or phase changing sets ) (where the N different phases are expressed as PHASE[O], PHASE[l], PHASE[2] ... PHASE[N-21, PHASE[N-11). As such, in order to transmit all of the bits making up a single coded block, PHASE[O] is used on & slots, PHASE[l] is used on Kt slots, PHASE[i] is used on Ki slots (where i = 0, 1, 2...N-1 (i denotes an integer that 20 satisfies OFilN-I)), and PHASE[N-I] is used on KN-I slots, such that Condition #A01 is met. (Condition #A01) &=K1...=Ki=...KN-l. Thatis,Ka=Kb(VaandVbwherea,b,=O, 1,2 ... N-1 (a 25 denotes an integer that satisfies 05am-1, b denotes an integer that satisfies OSb9-1), a # b). [0244] Then, when a communication system that supports multiple modulation schemes selects one such supported modulation scheme for use, Condition #A01 is preferably satisfied for the supported modulation scheme. However, when multiple modulation schemes are supported, each such 5 modulation scheme typically uses symbols transmitting a different number of bits per symbols (though some may happen to use the same number), Condition #A01 may not be satisfied for some modulation schemes. In such a case, the following condition applies instead of Condition #A01. [0245] 10 (Condition #A02) The difference between K, and Kb satisfies 0 or 1. That is, IK, - Kbl satisfies 0 or 1 (Va, Vb, where a, b = 0, 1, 2 ... N-1 (a denotes an integer that satisfies WIN-1, b denotes an integer that satisfies Olbg-1), a # b) Fig. 35 illustrates the varying numbers of symbols and slots needed in two 15 coded blocks when block codes are used. Fig. 35 illustrates the varying numbers of symbols and slots needed in each coded block when block codes are used when, for example, two streams sl and s2 are transmitted as indicated by the transmission device fiom Fig. 3 and Fig. 12, and the transmission device has two encoders. (Here, the transmission scheme may be any single-carrier scheme or multi-carrier scheme 20 such as OFDM.) As shown in Fig. 35, when block codes are used, there are 6000 bits making up a single coded block. In order to transmit these 6000 bits, the number of required symbols depends on the modulation scheme, being 3000 symbols for QPSK, 1500 symbols for 16-QAM, and 1000 symbols for 64-QAM. 25 [0246] The transmission device fiom Fig. 3 and the transmission device fiom Fig. 12 each transmit two streams at once, and have two encoders. As such, the two streams each transmit different code blocks. CLAIMS 1. A signal generation method for generating, from a plurality of baseband signals, a plurality of signals for transmission on a common frequency band and at a common time, comprising the steps of: 5 multiplying a first baseband signal si generated from a first set of bits by u, and multiplying a second baseband signal s2 generated from a second set of bits by V, where u and v denote real numbers different fix)m each other; performing a change of phase on each of the first baseband signal si multiplied by u and the second baseband signal s2 multiplied by v, thus generating a 10 first post-phase-change baseband signal u x si' and a second post-phzise-change baseband signal v x s2'; and applying weighting according to a predetermined matrix F to the first post-phase-cheinge baseband signal u x si' and to the second post-phase-change baseband signal v x s2', thus generating the plurality of signals for transmission on 15 the common frequency band and at the common time as a first weighted signal zl and a second weighted signal z2, wherein the first weighted signal zl and the second weighted signal z2 satisfy the relation: (zl,z2/ = F(uxsl',vxs2')'^ 20 and the change of phase is performed on the first baseband signal si multiplied by u and the second baseband signal s2 multiplied by v by using a phase modification value sequentially selected from among N phase modification value candidates, each of the N phase modification value candidates being selected at least once within a predetermined period. 25 2. A signal generation apparatus for generating, from a plurality of baseband signals, a plurality of signals for fransmission on a common frequency band and at a common time, comprising: 370 a power changer multiplying a first baseband signal si generated from a first set of bits by u, and multiplying a second baseband signal s2 generated from a second set of bits by v, where u and v denote real numbers different from each other; a phase changer performing a change of phase on each of the first baseband 5 signal si multiplied by u and the second baseband signal s2 multiplied by v, thus generating a first post-phase-change baseband signal u x si' and a second post-phase-change baseband signal v x s2'; and a weighting unit applying weighting according to a predetermined matrix F to the first post-phase-change baseband signal u x si' and to the second 10 post-phase-change baseband signal v x s2', thus generating the plurality of signals for transmission on the common frequency band and at the common time as a fu-st weighted signal zl and a second weighted signal z2, wherein the first weighted signal zl and the second weighted signal z2 satisfy the relation: 15 (zl,z2f = F ( u x s l ' , v x s 2 'f and the change of phase is performed on the first baseband signal si multiplied by u and the second baseband signal s2 multiplied by v by using a phase modification value sequentially selected from among N phase modification value candidates, each of the N phase modification value candidates being selected at least 20 once within a predetermmed period.