FORM 2
THE PATENTS ACT, 1970
(39 of 1970)
&
THE PATENTS RULES, 2003
COMPLETE SPECIFICATION
[See section 10, Rule 13]
OVERCURRENT DETECTION CIRCUITRY AND POWER CONVERTER;
MITSUBISHI ELECTRIC CORPORATION, A CORPORATION ORGANISED
AND EXISTING UNDER THE LAWS OF JAPAN, WHOSE ADDRESS IS 7-3,
MARUNOUCHI 2-CHOME, CHIYODA-KU, TOKYO 100-8310, JAPAN
THE FOLLOWING SPECIFICATION PARTICULARLY DESCRIBES THE
INVENTION AND THE MANNER IN WHICH IT IS TO BE PERFORMED.
2
DESCRIPTION
5
Field
[0001] The present disclosure relates to an overcurrent
detection circuitry of a semiconductor switching element
and a power converter including the overcurrent detection
10 circuitry.
Background
[0002] In recent years, semiconductor switching elements
such as an insulated gate bipolar transistor (IGBT) and a
15 metal oxide semiconductor field effect transistor (MOSFET)
are widely used for power converters such as inverters and
converters. Normally, in a semiconductor switching element
of a power converter, a gate voltage rises as the
semiconductor switching element is turned on. Then, when
20 the gate voltage reaches an operation threshold voltage of
the semiconductor switching element, a collector current
starts flowing. Thereafter, a collector voltage starts
decreasing, and decreases to around 0 V at the end.
[0003] However, when an anomaly such as an arm short
25 circuit or a load short circuit occurs in the power
converter, the collector current exceeds a predetermined
threshold value and becomes an overcurrent, and the
collector voltage is maintained at a high voltage without
being lowered, damaging the semiconductor switching
30 element. Therefore, in order to prevent damage to the
semiconductor switching element used in the power
converter, it is important to promptly detect an
overcurrent state of the semiconductor switching element.
3
[0004] In the following Patent Literature 1, by
detecting a collector voltage on the basis of a current
flowing to a voltage dividing circuit connected to a
collector terminal of a semiconductor switching element,
5 and determining whether the collector voltage exceeds a
reference voltage, an overcurrent of a current flowing to
the semiconductor switching element is detected.
Citation List
10 Patent Literature
[0005] Patent Literature 1: Japanese Patent Application
Laid-open No. 2012-195937
Summary
15 Technical Problem
[0006] However, when the collector voltage is reduced by
the voltage dividing circuit as in Patent Literature 1, the
current flowing to the voltage dividing circuit is reduced.
An overcurrent of the semiconductor switching element is
20 detected on the basis of this current, but the reduced
current is used for charging an electrostatic capacitance
being parasitic in a wiring line, and detection of the
overcurrent of the semiconductor switching element is
delayed.
25 [0007] The present disclosure has been made in view of
the above, and an object thereof is to obtain an
overcurrent detection circuitry capable of suppressing a
delay in overcurrent detection of a semiconductor switching
element due to an electrostatic capacitance being parasitic
30 in a wiring line.
Solution to Problem
[0008] To solve the above problems and achieve the
4
object, an overcurrent detection circuitry according to the
present disclosure includes: a voltage dividing circuit
adapted to reduce a voltage applied to a first electrode
and a second electrode of a semiconductor switching element
5 including the first electrode, the second electrode, and a
control electrode; a current amplifier circuit adapted to
amplify and output a current outputted from the voltage
dividing circuit; and an overcurrent determination circuit
adapted to determine whether or not the semiconductor
10 switching element is overcurrent, based on a current
outputted from the current amplifier circuit.
Advantageous Effects of Invention
[0009] According to the present disclosure, the
15 overcurrent detection circuitry exhibits an effect of being
able to suppress a delay in overcurrent detection of the
semiconductor switching element due to an electrostatic
capacitance being parasitic in a wiring line.
20 Brief Description of Drawings
[0010] FIG. 1 is a block diagram illustrating a
configuration of a power converter according to a first
embodiment.
FIG. 2 is a circuit diagram illustrating a
25 configuration of the power converter according to the first
embodiment.
FIG. 3 is a timing chart illustrating an operation of
the power converter according to the first embodiment.
FIG. 4 is a circuit diagram illustrating a
30 configuration of a power converter according to a second
embodiment.
FIG. 5 is a timing chart illustrating an operation of
the power converter according to the second embodiment.
5
FIG. 6 is a circuit diagram illustrating a
configuration of a power converter according to a third
embodiment.
FIG. 7 is a timing chart illustrating an operation of
5 the power converter according to the third embodiment.
FIG. 8 is a block diagram illustrating a configuration
of a power conversion system to which the power converter
according to the first embodiment is applied.
10 Description of Embodiments
[0011] Hereinafter, an overcurrent detection circuitry
and a power converter according to embodiments of the
present disclosure will be described in detail with
reference to the drawings. It should be noted that the
15 present disclosure is not limited by these embodiments.
[0012] First Embodiment.
FIG. 1 is a diagram illustrating a configuration
example of a power converter 500 according to a first
embodiment of the present disclosure. The power converter
20 500 includes a semiconductor module 100 and a drive
controller 300.
[0013] The semiconductor module 100 includes a
semiconductor switching element 101 and a diode 102. In
the present embodiment, an example in which an IGBT is used
25 as the semiconductor switching element 101 has been
described, but the present disclosure is not limited to
this. For example, a MOSFET, a bipolar transistor, or the
like may be used.
[0014] The semiconductor switching element 101 includes:
30 a first electrode 103; a second electrode 104, and a
control electrode 105. The semiconductor switching element
101 switches on/off of a current flowing between the first
electrode 103 and the second electrode 104 in response to a
6
gate signal applied to the control electrode 105, that is,
a gate voltage Vge. Note that the first electrode 103
corresponds to a collector in a case of an IGBT and a
bipolar transistor, and corresponds to a drain in a case of
5 a MOSFET. In addition, the second electrode 104
corresponds to an emitter in a case of an IGBT and a
bipolar transistor, and corresponds to a source in a case
of a MOSFET. Further, the control electrode 105
corresponds to a gate in a case of an IGBT and a MOSFET,
10 and corresponds to a base in a case of a bipolar
transistor.
[0015] The diode 102 is a reflux diode, and is connected
in antiparallel with the semiconductor switching element
101. That is: a cathode of the diode 102 is connected to
15 the first electrode 103 of the semiconductor switching
element 101; and an anode of the diode 102 is connected to
the second electrode 104 of the semiconductor switching
element 101.
[0016] The drive controller 300 includes a drive circuit
20 10 and an overcurrent detection circuitry 200. The drive
circuit 10 and the overcurrent detection circuitry 200 are
connected to the semiconductor module 100. The drive
circuit 10 transmits a gate signal for controlling on/off
of the semiconductor switching element 101 to the control
25 electrode 105 of the semiconductor switching element 101,
by a control circuit such as a microcomputer. The
overcurrent detection circuitry 200 detects an overcurrent
in the semiconductor switching element 101. When the
overcurrent is detected by the overcurrent detection
30 circuitry 200, a gate interruption signal Ssc for turning
off the semiconductor switching element 101 is transmitted
to the drive circuit 10.
[0017] The overcurrent detection circuitry 200 includes:
7
a voltage dividing circuit 20; a current amplifier circuit
30; and an overcurrent determination circuit 40. The
current amplifier circuit 30 is connected between the
voltage dividing circuit 20 and the overcurrent
5 determination circuit 40. The voltage dividing circuit 20
reduces a voltage applied to the first electrode 103 and
the second electrode 104 of the semiconductor switching
element 101. The current amplifier circuit 30 amplifies
and outputs a current outputted from the voltage dividing
10 circuit 20. The overcurrent determination circuit 40
determines whether or not the semiconductor switching
element 101 is overcurrent on the basis of the current
outputted from the current amplifier circuit 30.
[0018] FIG. 2 is a circuit diagram illustrating a
15 configuration example of the power converter 500 according
to the first embodiment of the present disclosure.
[0019] The drive circuit 10 includes: a control circuit
11; a turn-on MOSFET 12; a turn-off MOSFET 13; a turn-on
gate resistor 14; a turn-off gate resistor 15; a first DC
20 power supply 16; and a second DC power supply 17. The
turn-on MOSFET 12 is connected between the turn-on gate
resistor 14 and a positive power supply node 18; and the
turn-off MOSFET 13 is connected between the turn-off gate
resistor 15 and a negative power supply node 19. A
25 connection point between the turn-on gate resistor 14 and
the turn-off gate resistor 15 is connected by a wiring line
G to the control electrode 105 of the semiconductor
switching element 101 incorporated in the semiconductor
module 100. A connection point between the first DC power
30 supply 16 and the second DC power supply 17 is connected by
a wiring line S to the second electrode 104 of the
semiconductor switching element 101 incorporated in the
semiconductor module 100. The control circuit 11 controls
8
on/off of the semiconductor switching element 101 by
controlling the turn-on MOSFET 12 and the turn-off MOSFET
13 in response to a control signal Sg applied from an
external device (not illustrated).
5 [0020] The voltage dividing circuit 20 is configured by
connecting a plurality of resistance elements 21a, 21b,...,
and 21n and a resistance element 22 in series. The
resistance element 21a, which is one end of the plurality
of resistance elements 21a, 21b,..., 21n, is connected to
10 the first electrode 103 side of the semiconductor switching
element 101. In addition, one end of the resistance
element 22 is connected to the resistance element 21n and a
base 31a of an npn transistor 31 to be described later; and
another end is connected to an emitter 31b of the npn
15 transistor 31 to be described later. In the present
embodiment, the voltage dividing circuit 20 includes the
plurality of resistance elements, but may include a
plurality of constant voltage diodes, or may include both a
resistance element and a constant voltage diode. In
20 addition, the present disclosure is not limited to this
configuration as long as a voltage applied to the first
electrode 103 and the second electrode 104 of the
semiconductor switching element 101 is reduced.
[0021] The current amplifier circuit 30 includes the npn
25 transistor 31. A collector 31c of the npn transistor 31 is
directly connected to a positive power supply voltage of
the drive circuit 10 via a wiring line V. The emitter 31b
of the npn transistor 31 is connected to a resistance
element 51 of the overcurrent determination circuit 40 to
30 be described later via a wiring line C. Further, the base
31a and the emitter 31b of the npn transistor 31 are
individually connected to both ends of the resistance
element 22 of the voltage dividing circuit 20, and turn on
9
and off the npn transistor 31 in accordance with an end-toend voltage of the resistance element 22.
[0022] The overcurrent determination circuit 40
includes: an integration circuit 50; and a determination
5 circuit 60. The integration circuit 50: includes the
resistance element 51 and a capacitor 52; and outputs an
output result of the integration circuit 50 on the basis of
a current outputted from the current amplifier circuit 30.
The determination circuit 60 includes: a diode 61; a
10 comparator 62; and a DC power supply 63. On the basis of
the output result of the integration circuit 50, the
determination circuit 60 determines whether or not the
semiconductor switching element 101 is overcurrent by
comparison with a predetermined operation threshold voltage
15 Vref.
[0023] The resistance element 51 of the integration
circuit 50 is connected between the wiring line C and an
input node 64 of the comparator 62. The capacitor 52 is
connected between the input node 64 of the comparator 62
20 and the negative power supply node 19. When a voltage
generated across the capacitor 52 exceeds the operation
threshold voltage Vref of the comparator 62, the gate
interruption signal Ssc for turning off the semiconductor
switching element 101 is outputted from the comparator 62
25 to the control circuit 11.
[0024] An anode of the diode 61 of the determination
circuit 60 is connected to the input node 64 of the
comparator 62, and a cathode of the diode 61 is connected
to a drain of the turn-off MOSFET 13. As a result, when
30 the semiconductor switching element 101 is in an off state,
that is, when the turn-on MOSFET 12 is off and the turn-off
MOSFET 13 is on, electric charges accumulated in the
capacitor 52 are discharged via the diode 61 and the turn-
10
off MOSFET 13.
[0025] FIG. 3 is a timing chart illustrating an
operation example of the power converter 500 according to
the first embodiment of the present disclosure. First, an
5 operation at normal time of the power converter 500 will be
described with reference to FIG. 3. A vertical axis of the
timing chart in FIG. 3 indicates, in order from the top:
the external control signal Sg; the gate voltage Vge of the
semiconductor switching element 101; a collector current Ic
10 flowing in the semiconductor switching element 101; a
collector voltage Vce applied to the semiconductor
switching element 101; a base-emitter voltage Vbe31 of the
npn transistor 31 of the current amplifier circuit 30; and
an end-to-end voltage Vsc of the capacitor 52. A
15 horizontal axis represents a period of time t.
[0026] Before time t0 in FIG. 3, that is, during a turnoff period of the semiconductor module 100, the turn-on
MOSFET 12 is turned off, and the turn-off MOSFET 13 is
turned on. This causes electric charges of the capacitor
20 52 to be discharged via the diode 61 and the turn-off
MOSFET 13, so that a potential of the input node 64 of the
comparator 62 becomes equivalent to that of the negative
power supply node 19.
[0027] At time t0, in response to the external control
25 signal Sg switching from an OFF command to an ON command,
the control circuit 11 turns on the turn-on MOSFET 12 and
turns off the turn-off MOSFET 13. This causes a charge
current to flow from the first DC power supply 16 to an
input capacitance of the semiconductor switching element
30 101 via: the positive power supply node 18; the turn-on
MOSFET 12; the turn-on gate resistor 14; and the wiring
line G. Then the gate voltage Vge of the semiconductor
switching element 101 starts increasing.
11
[0028] As the gate voltage Vge increases, the
semiconductor switching element 101 enters a turn-on
operation. Since a connection point 70 has a higher
potential than the input node 64 of the comparator 62, the
5 discharge of the capacitor 52 is stopped. At this time, a
voltage corresponding to a bus voltage Vdd, which is a main
voltage inputted to the power converter 500, is applied
across the voltage dividing circuit 20. Since the end-toend voltage of the resistance element 22 exceeds an
10 operation threshold voltage of the npn transistor 31, the
npn transistor 31 is turned on. When the npn transistor 31
is turned on, an amplified current Iz flows from the first
DC power supply 16 to a collector of the npn transistor 31
via the wiring line V. This amplified current Iz flows
15 through the wiring line C, and charges the capacitor 52 via
the resistance element 51 of the overcurrent determination
circuit 40.
[0029] In a case where the current amplifier circuit 30
is not provided, a current flowing into the overcurrent
20 determination circuit 40 is small, for example, several
hundreds μA. Therefore, it takes time to charge the
electrostatic capacitance being parasitic in the wiring
line, and as a result, it takes time for the end-to-end
voltage Vsc of the capacitor 52 to rise with respect to the
25 period of time t. However, in the present embodiment due
to the current amplifier circuit 30, the amplified current
Iz flowing to the wiring line C is large enough, for
example several tens of mA, to instantaneously charge the
electrostatic capacitance being parasitic in the wiring
30 line C. Therefore, most of the amplified current Iz flows
into the overcurrent determination circuit 40. Therefore,
the end-to-end voltage Vsc of the capacitor 52 with respect
to the period of time t is determined by: a difference
12
value V+-V- between a positive power supply voltage V+ and
a negative power supply voltage V-; and time constants R51
and C52 determined by a resistance value R51 of the
resistance element 51 and a capacitance value C52 of the
5 capacitor 52. That is, it can be expressed as the
following Formula (1), and the end-to-end voltage Vsc of
the capacitor 52 rises according to Formula (1) as
indicated from time t0 to time t4 in FIG. 3.
[0030] Vsc(t)=(V+-V-)∙(1-exp(-t/(R51∙C52))...(1)
10 [0031] When the gate voltage Vge exceeds a threshold
voltage Vth of the semiconductor switching element 101 at
time t2, the collector current Ic starts flowing from the
first electrode 103 to the second electrode 104. At this
time, an induced electromotive force VL=Ls∙dIc/dt is
15 generated, which is represented by a product of a time
change rate dIc/dt of the collector current Ic and an
inductance Ls being parasitic in the main circuit. Then,
the collector voltage Vce applied to the semiconductor
switching element 101 decreases by an amount of the induced
20 electromotive force VL from the bus voltage Vdd. However,
since the collector voltage Vce still maintains a high
voltage, the on state of the npn transistor 31 is
continued, and the end-to-end voltage Vsc of the capacitor
52 continues to rise according to Formula (1).
25 [0032] A period from time t3 to time t5 is a Miler
period in which the gate voltage Vge becomes constant due
to a Miler effect of the semiconductor switching element
101. During this Miler period, the collector voltage Vce
greatly fluctuates, and the collector voltage Vce decreases
30 to around 0 V at time t5. At time t4, the end-to-end
voltage of the resistance element 22 of the voltage
dividing circuit 20 does not exceed the operation threshold
voltage of the npn transistor 31 of the current amplifier
13
circuit 30, which causes the base-emitter voltage Vbe31 of
the npn transistor 31 to decrease, that is, causes the npn
transistor 31 to be turned off. As a result, the amplified
current Iz for charging the capacitor 52 of the overcurrent
5 determination circuit 40 is interrupted, and the end-to-end
voltage Vsc of the capacitor 52 no longer rises.
[0033] When the Miler period is over at time t5, the
gate voltage Vge increases again, and the turn-on operation
ends when the gate voltage Vge reaches the positive power
10 supply voltage V+ at time t6.
[0034] Next, a turn-off operation from time t7 to time
t13 will be described. At time t7 in FIG. 3, in response
to the external control signal Sg switching from the ON
command to the OFF command, the control circuit 11 turns
15 off the turn-on MOSFET 12 and turns on the turn-off MOSFET
13. This causes: a discharge current to flow from the
input capacitance of the semiconductor switching element
101 to the second DC power supply 17 via the wiring line G,
the turn-off gate resistor 15, the turn-off MOSFET 13, and
20 the negative power supply node 19; and the gate voltage Vge
to start decreasing. As a result, the semiconductor
switching element 101 shifts to the turn-off operation. At
this time, since a potential of the connection point 70
instantaneously becomes V-, residual electric charges of
25 the capacitor 52 are quickly discharged via the diode 61.
[0035] Between time t7 and time t8, the collector
current Ic and the collector voltage Vce do not change.
After time t8, the collector voltage Vce applied to the
semiconductor switching element 101 starts increasing,
30 which provides a Miler period in which the gate voltage Vge
is substantially constant from time t8 to time t10, and the
collector voltage Vce reaches the bus voltage Vdd at time
t10.
14
[0036] When the Miler period is over at time t10, the
gate voltage Vge starts decreasing again. When the gate
voltage Vge falls below the threshold voltage Vth of the
semiconductor switching element 101 at time t11, the
5 collector current Ic no longer flows. Then, when the gate
voltage Vge reaches the negative power supply voltage V- as
at time t13, the turn-off operation ends.
[0037] During the turn-off period, since the collector
voltage Vce of the semiconductor switching element 101
10 becomes equivalent to the bus voltage Vdd, a voltage
corresponding to the bus voltage Vdd is applied across the
voltage dividing circuit 20. Therefore, the end-to-end
voltage of the resistance element 22 of the voltage
dividing circuit 20 exceeds the operation threshold voltage
15 of the npn transistor 31 of the current amplifier circuit
30, which causes the base-emitter voltage Vbe31 of the npn
transistor 31 to rise, that is, causes the npn transistor
31 to be turned on. As a result, the amplified current Iz
flows from the first DC power supply 16 to the collector of
20 the npn transistor 31 via the positive power supply node 18
and the wiring line V, and this amplified current Iz flows
into the overcurrent determination circuit 40 via the
wiring line C. However, since most of the amplified
current Iz is discharged through the resistance element 51
25 and the diode 61, charging of the capacitor 52 is
suppressed. Therefore, after time t7, the end-to-end
voltage Vsc of the capacitor 52 during the turn-off period
is maintained equivalent to 0 V.
[0038] Next, an operation of the power converter 500 at
30 a time of short-circuit anomaly will be described with
reference to FIG. 3. Note that an operation from time t14
to time t16, that is, an operation from when the
semiconductor module 100 shifts from the turn-off state to
15
the turn-on operation until when the end-to-end voltage Vsc
of the capacitor 52 rises according to Formula (1) is
similar to the operation at normal time.
[0039] When the gate voltage Vge exceeds the threshold
5 voltage Vth of the semiconductor switching element 101 at
time t16, the collector current Ic starts flowing from the
first electrode 103 to the second electrode 104. At this
time, in a no-load state due to occurrence of a shortcircuit anomaly, the collector current Ic instantaneously
10 becomes a large current and becomes a larger value than the
collector current Ic during the normal operation. During
the normal operation, since the collector voltage Vce of
the semiconductor switching element 101 has decreased to
around 0 V, a load such as a motor has held the bus voltage
15 Vdd. However, since there is no load such as a motor at
the time of the short-circuit anomaly, the semiconductor
switching element 101 keeps holding most of the bus voltage
Vdd. Therefore, the base-emitter voltage Vbe31 of the npn
transistor 31 of the current amplifier circuit 30 rises,
20 that is, the npn transistor 31 continues to be on, and the
end-to-end voltage Vsc of the capacitor 52 continues to
rise according to Formula (1).
[0040] At time t17, the end-to-end voltage Vsc of the
capacitor 52 reaches the operation threshold voltage Vref
25 of the comparator 62 of the overcurrent determination
circuit 40. Then, the comparator 62 of the overcurrent
determination circuit 40 determines as an overcurrent, and
transmits the gate interruption signal Ssc for turning off
the semiconductor switching element 101 from the comparator
30 62 to the control circuit 11 of the drive circuit 10. Upon
receiving the gate interruption signal Ssc, the control
circuit 11 turns off the turn-on MOSFET 12 and turns on the
turn-off MOSFET 13. This causes: a discharge current to
16
flow from the input capacitance of the semiconductor
switching element 101 to the second DC power supply 17 via
the wiring line G, the turn-off gate resistor 15, the turnoff MOSFET 13, and the negative power supply node 19; and
5 the gate voltage Vge of the semiconductor switching element
101 to decrease. As the gate voltage Vge of the
semiconductor switching element 101 decreases, a shortcircuit current is interrupted at time t17, and the
semiconductor switching element 101 can be protected from
10 the overcurrent.
[0041] As described above, according to the present
embodiment, the overcurrent detection circuitry 200
includes: the voltage dividing circuit 20; the current
amplifier circuit 30; and the overcurrent determination
15 circuit 40. By connecting the current amplifier circuit 30
in between the voltage dividing circuit 20 and the
overcurrent determination circuit 40, a current flowing
through the voltage dividing circuit 20 is amplified. This
makes it possible to instantaneously charge the
20 electrostatic capacitance being parasitic in the wiring
line C, and makes it possible to suppress a delay in
overcurrent detection of the semiconductor switching
element 101 due to the electrostatic capacitance being
parasitic in the wiring line.
25 [0042] Second Embodiment.
FIG. 4 is a circuit diagram illustrating a
configuration example of a power converter 500A according
to a second embodiment of the present disclosure.
Identical parts to those in FIG. 1 are denoted by the same
30 reference numerals, the description thereof is omitted, and
only different parts will be described here. The power
converter 500A according to the present embodiment is
different in including a drive controller 300A as
17
illustrated in FIG. 4. More specifically, the drive
controller 300A has different circuit configurations of a
voltage dividing circuit 20A and a current amplifier
circuit 30A.
5 [0043] The voltage dividing circuit 20A is configured by
connecting a plurality of resistance elements 21a, 21b,...,
and 21n and a resistance element 23 in series. In the
present embodiment, the resistance element 21a, which is
one end of the plurality of resistance elements 21a,
10 21b,..., 21n, is connected to the first electrode 103 side
of the semiconductor switching element 101. In addition,
one end of the resistance element 23 is connected to the
resistance element 21n and a base 32a of an npn transistor
32 to be described later, and another end is connected to
15 the wiring line S which is a source wiring line of the
semiconductor switching element 101. In the present
embodiment, the voltage dividing circuit 20A includes the
plurality of resistance elements, but may include a
plurality of constant voltage diodes, or may include both a
20 resistance element and a constant voltage diode.
[0044] The current amplifier circuit 30A includes the
npn transistor 32, resistance elements 33 and 34, and a pnp
transistor 35. The base 32a of the npn transistor 32 is
connected between the resistance element 21n and the
25 resistance element 23 of the voltage dividing circuit 20A.
A collector 32c of the npn transistor 32 is directly
connected to the control electrode 105 and the drive
circuit 10 of the semiconductor switching element 101 via
the resistance element 33 and the wiring line G. An
30 emitter 32b of the npn transistor 32 is directly connected
to the second electrode 104 of the semiconductor switching
element 101 via the wiring line S. The resistance element
33 functions as a collector-current limiting resistor of
18
the npn transistor 32. The base 32a and the emitter 31b of
the npn transistor 32 are connected to both ends of the
resistance element 23 of the voltage dividing circuit 20A,
so that the npn transistor 32 is turned on and off in
5 accordance with an end-to-end voltage of the resistance
element 23.
[0045] One end of the resistance element 34 is connected
to a base 35a of the pnp transistor 35, and another end of
the resistance element 34 is connected between the
10 collector 32c of the npn transistor 32 and the resistance
element 33. The resistance element 34 functions as a basecurrent limiting resistor of the pnp transistor 35. In the
pnp transistor 35: an emitter 35c is connected to the
wiring line G; the base 35a is connected to the resistance
15 element 34; and a collector 35b is connected to the
resistance element 51 of the overcurrent determination
circuit 40 via the wiring line C.
[0046] FIG. 5 is a timing chart illustrating an
operation example of the power converter 500A according to
20 the second embodiment of the present disclosure. First, an
operation at normal time of the power converter 500A will
be described with reference to FIG. 5. A vertical axis of
the timing chart of FIG. 5 indicates, in order from the
top: the external control signal Sg; the gate voltage Vge
25 of the semiconductor switching element 101; the collector
current Ic flowing in the semiconductor switching element
101; the collector voltage Vce applied to the semiconductor
switching element 101; a base-emitter voltage Vbe32 of the
npn transistor 32 of the current amplifier circuit 30A; a
30 base-emitter voltage Vbe35 of the pnp transistor 35; and
the end-to-end voltage Vsc of the capacitor 52. A
horizontal axis represents a period of time t.
[0047] Before time t0 in FIG. 5, that is, during a turn-
19
off period of the semiconductor module 100, the turn-on
MOSFET 12 is turned off, and the turn-off MOSFET 13 is
turned on. This causes electric charges of the capacitor
52 to be discharged via the diode 61 and the turn-off
5 MOSFET 13, so that a potential of the input node 64 of the
comparator 62 becomes equivalent to that of the negative
power supply node 19.
[0048] At time t0, in response to an external control
signal switching from an OFF command to an ON command, the
10 control circuit 11 turns on the turn-on MOSFET 12 and turns
off the turn-off MOSFET 13. This causes: a charge current
to flow from the first DC power supply 16 to an input
capacitance of the semiconductor switching element 101 via
the positive power supply node 18, the turn-on MOSFET 12,
15 the turn-on gate resistor 14, and the wiring line G; and
the gate voltage Vge of the semiconductor switching element
101 to start increasing.
[0049] As the gate voltage Vge increases, the
semiconductor switching element 101 enters a turn-on
20 operation. Since the connection point 70 has a higher
potential than the input node 64 of the comparator 62, the
discharge of the capacitor 52 is stopped. At this time, a
voltage equivalent to the bus voltage Vdd, which is a main
voltage inputted to the power converter 500A, is applied
25 across the voltage dividing circuit 20A, and reduced by a
total resistance value of the resistance elements 21a,
21b,..., up to 21n and a resistance value of the resistance
element 23. At this time, since the end-to-end voltage of
the resistance element 23 exceeds an operation threshold
30 voltage of the npn transistor 32, the npn transistor 32 is
turned on.
[0050] Further, at time t1, when Vge of the
semiconductor switching element 101 is increased by the
20
turn-on operation, and a potential of the wiring line G
which is a gate wiring line of the semiconductor switching
element 101, becomes higher than a potential of the wiring
line S which is a source a wiring line of the semiconductor
5 switching element 101; a collector current flows from the
first DC power supply 16 to the npn transistor 32 via the
positive power supply node 18, the turn-on MOSFET 12, the
turn-on gate resistor 14, the wiring line G, and the
resistance element 33; and a potential of the collector 32c
10 of the npn transistor 32 becomes equal to a potential of
the wiring line S. Then, a current starts flowing to the
base 35a of the pnp transistor 35, and the pnp transistor
35 is turned on. When the pnp transistor 35 is turned on,
the amplified current Iz flows to the pnp transistor 35 via
15 the positive power supply node 18, the turn-on MOSFET 12,
the turn-on gate resistor 14, and the wiring line G. This
amplified current Iz flows through the wiring line C, and
charges the capacitor 52 via the resistance element 51 of
the overcurrent determination circuit 40.
20 [0051] The amplified current Iz flowing in the wiring
line C is large enough, for example several tens of mA, to
instantaneously charge the electrostatic capacitance being
parasitic in the wiring line C. Therefore, most of the
amplified current Iz flows into the overcurrent
25 determination circuit 40. When the pnp transistor 35 is
turned on at time t1, a potential of the emitter 35b of the
pnp transistor 35 becomes equal to a potential of the
wiring line G. Therefore, the end-to-end voltage Vsc of
the capacitor 52 with respect to the period of time t is
30 determined: by a difference value V+-V- between a positive
power supply voltage V+ and a negative power supply voltage
V-; and the time constants R51 and C52 determined by the
resistance value R51 of the resistance element 51 and the
21
capacitance value C52 of the capacitor 52. That is, it can
be expressed as the following Formula (2), and the end-toend voltage Vsc of the capacitor 52 rises according to
Formula (2).
5 [0052] Vsc(t)=(V+-V-)∙(1-exp(-t/(R51∙C52))...(2)
[0053] When the gate voltage Vge exceeds the threshold
voltage Vth of the semiconductor switching element 101 at
time t2, the collector current Ic starts flowing from the
first electrode 103 to the second electrode 104. At this
10 time, an induced electromotive force VL=Ls∙dIc/dt is
generated, which is represented by a product of the time
change rate dIc/dt of the collector current Ic and the
inductance Ls being parasitic in the main circuit. Then,
the collector voltage Vce applied to the semiconductor
15 switching element 101 decreases by an amount of the induced
electromotive force VL from the bus voltage Vdd. However,
since the collector voltage Vce still maintains a high
voltage, the on state of the npn transistor 32 is
continued, and the end-to-end voltage Vsc of the capacitor
20 52 continues to rise according to Formula (2).
[0054] A period from time t3 to time t5 is a Miler
period in which the gate voltage Vge becomes constant due
to a Miler effect of the semiconductor switching element
101. During the Miler period, the collector voltage Vce
25 greatly fluctuates, and the collector voltage Vce decreases
to around 0 V at time t5. At time t4, the end-to-end
voltage of the resistance element 23 of the voltage
dividing circuit 20A falls below the operation threshold
voltage of the npn transistor 32 of the current amplifier
30 circuit 30A, and the base-emitter voltage Vbe32 of the npn
transistor 32 decreases, that is, the npn transistor 32 is
turned off. When the npn transistor 32 is turned off, a
potential of the collector 32c of the npn transistor 32
22
becomes equal to a potential of the wiring line G, so that
a current no longer flows in the base 35b of the pnp
transistor 35, and the pnp transistor 35 is turned off. As
a result, the amplified current Iz for charging the
5 capacitor 52 of the overcurrent determination circuit 40 is
interrupted, and the end-to-end voltage Vsc of the
capacitor 52 no longer rises, that is, charging of the
capacitor 52 is stopped.
[0055] When the Miler period is over at time t5, the
10 gate voltage Vge increases again, and the turn-on operation
ends when the gate voltage Vge reaches the positive power
supply voltage V+ at time t6.
[0056] Next, a turn-off operation from time t7 to time
t13 will be described. At time t7 in FIG. 5, in response
15 to the external control signal Sg switching from an ON
command to an OFF command, the control circuit 11 turns off
the turn-on MOSFET 12 and turns on the turn-off MOSFET 13.
This causes a discharge current to flow from the input
capacitance of the semiconductor switching element 101 to
20 the second DC power supply 17 via the wiring line G, the
turn-off gate resistor 15, the turn-off MOSFET 13, and the
negative power supply node 19, and the gate voltage Vge
starts decreasing. As a result, the semiconductor
switching element 101 shifts to the turn-off operation. At
25 this time, since a potential of the connection point 70
instantaneously becomes V-, residual electric charges of
the capacitor 52 are quickly discharged via the diode 61.
[0057] Between time t7 and time t8, the collector
current Ic and the collector voltage Vce do not change.
30 After time t8, the collector voltage Vce applied to the
semiconductor switching element 101 starts increasing,
which provides a Miler period in which the gate voltage Vge
is substantially constant from time t8 to time t10, and the
23
collector voltage Vce reaches the bus voltage Vdd at time
t10.
[0058] When the Miler period is over at time t10, the
gate voltage Vge starts decreasing again. When the gate
5 voltage Vge falls below the threshold voltage Vth of the
semiconductor switching element 101 at time t11, the
collector current Ic no longer flows. Then, when the gate
voltage Vge reaches the negative power supply voltage V- as
at time t13, the turn-off operation ends.
10 [0059] During the turn-off period, since the collector
voltage Vce of the semiconductor switching element 101 is
equal to the bus voltage Vdd, a voltage equivalent to the
bus voltage Vdd is applied across the voltage dividing
circuit 20A. Therefore, the end-to-end voltage of the
15 resistance element 23 of the voltage dividing circuit 20A
exceeds the operation threshold voltage of the npn
transistor 32 of the current amplifier circuit 30A, which
causes the base-emitter voltage Vbe32 of the npn transistor
32 to rise, and causes the npn transistor 32 to be turned
20 on.
[0060] When the npn transistor 32 is turned on, the
collector 32c of the npn transistor 32 has a potential
equal to that of the wiring line S. However, since the
potential of the wiring line G is lower than that of the
25 wiring line S at the time of turn-off, a base current of
the pnp transistor 35 no longer flows, the base-emitter
voltage Vbe35 of the pnp transistor 35 decreases, and the
pnp transistor 35 is turned off. As a result, a current
for charging the capacitor 52 of the overcurrent
30 determination circuit 40 is interrupted, and the capacitor
52 is discharged via the diode 61. Therefore, after time
t7, the end-to-end voltage Vsc of the capacitor 52 during
the turn-off period is maintained equivalent to 0 V.
24
[0061] Next, an operation of the power converter 500A at
a time of short-circuit anomaly will be described with
reference to FIG. 5. Note that, an operation from time t14
to time t16, that is, an operation from when the
5 semiconductor module 100 shifts from the turn-off state to
the turn-on operation until when the end-to-end voltage Vsc
of the capacitor 52 rises according to Formula (2) is
similar to the operation at normal time.
[0062] When the gate voltage Vge exceeds the threshold
10 voltage Vth of the semiconductor switching element 101 at
time t16, the collector current Ic starts flowing from the
first electrode 103 to the second electrode 104. At this
time, in a no-load state due to occurrence of a shortcircuit anomaly, the collector current Ic instantaneously
15 becomes a large current and becomes a larger value than the
collector current Ic during the normal operation. During
the normal operation, since the collector voltage Vce of
the semiconductor switching element 101 has decreased to
around 0 V, a load such as a motor to be described later
20 has held the bus voltage Vdd. However, since there is no
load such as a motor at the time of the short-circuit
anomaly, the semiconductor switching element 101 keeps
holding most of the bus voltage Vdd. Therefore, the baseemitter voltage Vbe32 of the npn transistor 32 and the
25 base-emitter voltage Vbe35 of the pnp transistor 35 of the
current amplifier circuit 30A rise, that is, the npn
transistor 32 and the pnp transistor 35 continue to be on,
and the end-to-end voltage Vsc of the capacitor 52
continues to rise according to Formula (2).
30 [0063] At time t17, the end-to-end voltage Vsc of the
capacitor 52 reaches the operation threshold voltage Vref
of the comparator 62 of the overcurrent determination
circuit 40. Then, the comparator 62 of the overcurrent
25
determination circuit 40 determines as an overcurrent, and
transmits the gate interruption signal Ssc for turning off
the semiconductor switching element 101 from the comparator
62 to the control circuit 11 of the drive circuit 10. Upon
5 receiving the gate interruption signal Ssc, the control
circuit 11 turns off the turn-on MOSFET 12 and turns on the
turn-off MOSFET 13. This causes: a discharge current to
flow from the input capacitance of the semiconductor
switching element 101 to the second DC power supply 17 via
10 the wiring line G, the turn-off gate resistor 15, the turnoff MOSFET 13, and the negative power supply node 19; and
the gate voltage Vge of the semiconductor switching element
101 to decrease. As the gate voltage Vge of the
semiconductor switching element 101 decreases, a short15 circuit current is interrupted at time t17, and the
semiconductor switching element 101 can be protected from
the overcurrent.
[0064] As described above, according to the present
embodiment, the voltage dividing circuit 20A and the
20 current amplifier circuit 30A are connected to the drive
circuit 10 and the overcurrent determination circuit 40 by
using the wiring line G, the wiring line C, and the wiring
line S. As a result, the number of wiring lines can be
further reduced in addition to the effects similar to those
25 of the first embodiment.
[0065] Third Embodiment.
FIG. 6 is a circuit diagram illustrating a
configuration example of a power converter 500B according
to a third embodiment of the present disclosure. Identical
30 parts to those in FIGS. 1, 2, and 4 are denoted by the same
reference numerals, the description thereof is omitted, and
only different parts will be described here. The power
converter 500B according to the present embodiment is
26
different in including a drive controller 300B as
illustrated in FIG. 6. More specifically, the drive
controller 300B has a different circuit configuration of a
voltage dividing circuit 20B.
5 [0066] The voltage dividing circuit 20B is configured by
connecting a plurality of resistance elements 21a, 21b,...,
and 21n and the resistance element 23 in series, and
further connecting capacitors 24a, 24b,..., and 24n to both
ends of each of the resistance elements 21a, 21b,..., and
10 21n. That is, the resistance elements 21a, 21b,..., and
21n and the capacitors 24a, 24b,..., and 24n are connected
in parallel. In the present embodiment, the resistance
element 21a, which is one end of the plurality of
resistance elements 21a, 21b,..., 21n, is connected to the
15 first electrode 103 side of the semiconductor switching
element 101. In addition, one end of the resistance
element 23 is connected to the resistance element 21n and
the base 32a of the npn transistor 32, and another end is
connected to the wiring line S which is a source wiring
20 line of the semiconductor switching element 101. In the
present embodiment, the voltage dividing circuit 20B
includes the plurality of resistance elements, but may
include a plurality of constant voltage diodes, or may
include both a resistance element and a constant voltage
25 diode.
[0067] FIG. 7 is a timing chart illustrating an
operation example of the power converter 500B according to
the third embodiment of the present disclosure. First, an
operation at normal time of the power converter 500B will
30 be described with reference to FIG. 7. The timing chart of
FIG. 7 indicates, in order from the top of a vertical axis,
the external control signal Sg, the gate voltage Vge of the
semiconductor switching element 101, the collector current
27
Ic flowing to the semiconductor switching element 101, the
collector voltage Vce applied to the semiconductor
switching element 101, the base-emitter voltage Vbe32 of
the npn transistor 32 of the current amplifier circuit 30A,
5 the base-emitter voltage Vbe35 of the pnp transistor 35,
and the end-to-end voltage Vsc of the capacitor 52. A
horizontal axis represents a period of time t. Note that,
regarding the operation at normal time of the power
converter 500B, an operation until time t3 and an operation
10 from time t5 to time t13 are similar to those in the second
embodiment, and thus only an operation from time t3 to time
t5 will be described.
[0068] A period from time t3 to time t5 is a Miler
period in which the gate voltage Vge becomes constant due
15 to a Miler effect of the semiconductor switching element
101. During this Miler period, the collector voltage Vce
greatly fluctuates, and the collector voltage Vce decreases
to around 0 V at time t5.
[0069] At this time, a voltage change equivalent to the
20 collector voltage Vce also occurs across the resistance
elements 21a, 21b,..., and 21n of the voltage dividing
circuit 20B. Therefore, a displacement current Idp flows
via capacitors 24a, 24b,..., 24n according to the voltage
change. Since the displacement current Idp also flows to
25 the resistance element 23, a potential of the base 32a is
lower than a potential of the emitter 32b of the npn
transistor 32. Therefore, at time t3, the base-emitter
voltage Vbe32 of the npn transistor 32 decreases, and the
npn transistor 32 is turned off.
30 [0070] When the npn transistor 32 is turned off, the
collector 32c of the npn transistor 32 has a potential
equal to that of the wiring line G, so that a current no
longer flows to the base 35a of the pnp transistor 35, and
28
the pnp transistor 35 is also turned off at time t3. As a
result, the amplified current Iz for charging the capacitor
52 of the overcurrent determination circuit 40 is
interrupted, and the end-to-end voltage Vsc of the
5 capacitor 52 no longer rises, that is, charging of the
capacitor 52 is stopped.
[0071] Next, an operation of the power converter 500B at
a time of a short-circuit anomaly will be described with
reference to FIGS. 6 and 7. Note that an operation from
10 time t14 to time t16, that is, an operation from when the
semiconductor module 100 shifts from the turn-off state to
the turn-on operation until when the end-to-end voltage Vsc
of the capacitor 52 rises according to Formula (2) is
similar to the operation at normal time of the second and
15 third embodiments.
[0072] When the gate voltage Vge exceeds the threshold
voltage Vth of the semiconductor switching element 101 at
time t16, the collector current Ic starts flowing from the
first electrode 103 to the second electrode 104. At this
20 time, in a no-load state due to occurrence of a shortcircuit anomaly, the collector current Ic instantaneously
becomes a large current and becomes a larger value than the
collector current Ic during the normal operation. During
the normal operation, since the collector voltage Vce of
25 the semiconductor switching element 101 has decreased to
around 0 V, a load such as a motor has held the bus voltage
Vdd. However, since there is no load such as a motor at
the time of the short-circuit anomaly, the semiconductor
switching element 101 keeps holding most of the bus voltage
30 Vdd. Therefore, the base-emitter voltage Vbe32 of the npn
transistor 32 and the base-emitter voltage Vbe35 of the pnp
transistor 35 of the current amplifier circuit 30A rise,
that is, the npn transistor 32 and the pnp transistor 35
29
continue to be on, and the end-to-end voltage Vsc of the
capacitor 52 continues to rise according to Formula (2).
[0073] At time t17a, the end-to-end voltage Vsc of the
capacitor 52 reaches the operation threshold voltage Vref
5 of the comparator 62 of the overcurrent determination
circuit 40. Then, the comparator 62 of the overcurrent
determination circuit 40 determines as an overcurrent, and
transmits the gate interruption signal Ssc for turning off
the semiconductor switching element 101 from the comparator
10 62 to the control circuit 11 of the drive circuit 10. When
receiving the gate interruption signal Ssc, the control
circuit 11 turns off the turn-on MOSFET 12 and turns on the
turn-off MOSFET 13. This causes a discharge current to
flow from the input capacitance of the semiconductor
15 switching element 101 to the second DC power supply 17 via
the wiring line G, the turn-off gate resistor 15, the turnoff MOSFET 13, and the negative power supply node 19, and
the gate voltage Vge of the semiconductor switching element
101 decreases. As the gate voltage Vge of the
20 semiconductor switching element 101 decreases, a shortcircuit current is interrupted at time t17a, and the
semiconductor switching element 101 can be protected from
the overcurrent.
[0074] In the power converters 500 and 500A described in
25 the first and second embodiments, a timing at which the npn
transistor 32 or the pnp transistor 35 is turned off after
the turn-on is started at time t0 is time t4, but is to be
time t3, which is the timing at which the collector voltage
Vce starts to change greatly, in the power converter 500B
30 described in the third embodiment.
[0075] As described above, according to the present
embodiment, the voltage dividing circuit 20B is configured
such that the capacitors 24a, 24b,..., and 24n are
30
connected to both ends of each of the resistance elements
21a, 21b,..., and 21n. Therefore, in addition to the
effects similar to those of the first and second
embodiments, the time constants R51 and C52 determined by
5 the resistance value R51 of the resistance element 51 and
the capacitance value C52 of the capacitor 52 of the
overcurrent determination circuit 40 can be made smaller
than those in the second embodiment, and overcurrent
detection and overcurrent protection can be performed more
10 quickly.
[0076] FIG. 8 is a block diagram illustrating a
configuration in which the power converter 500 according to
the above-described first embodiment is applied to a power
conversion system 700. Although FIG. 8 illustrates a case
15 where the present disclosure is applied to a three-phase
inverter, the present disclosure is not limited to a
specific power conversion system.
[0077] The power conversion system 700 illustrated in
FIG. 8 includes a power supply 400, the power converter
20 500, and a load 600. The power supply 400 is a DC power
supply, and supplies DC power to the power converter 500.
The power supply 400 can include various components, and
may include, for example, a DC system, a solar cell, and a
storage battery, or may include a rectifier circuit and an
25 AC/DC converter connected to an AC system. In addition,
the power supply 400 may include a DC/DC converter that
converts DC power outputted from the DC system into
predetermined power.
[0078] The power converter 500 is a three-phase inverter
30 connected between the power supply 400 and the load 600,
converts DC power supplied from the power supply 400 into
AC power, and supplies the AC power to the load 600. As
illustrated in FIG. 8, the power converter 500 includes a
31
main conversion circuit 110 that converts DC power into AC
power and outputs the AC power, and the drive controller
300 that outputs a control signal for controlling the main
conversion circuit 110 to the main conversion circuit.
5 [0079] The load 600 is a three-phase electric motor
driven by AC power supplied from the power converter 500.
Note that the load 600 is not limited to a specific
application, but is an electric motor mounted on various
electric devices, and is used as, for example, an electric
10 motor for a railway vehicle, a hybrid vehicle, an electric
vehicle, an elevator, or an air conditioner.
[0080] The main conversion circuit 110 converts DC power
supplied from the power supply 400 into AC power by
switching of the semiconductor switching element 101
15 described in the first embodiment, and supplies the AC
power to the load 600. When a short-circuit anomaly occurs
in the main conversion circuit 110 or the load 600, the
drive controller 300 of the power converter 500 detects an
overcurrent of the semiconductor switching element 101.
20 Since the drive controller 300 is obtained by applying the
present disclosure, it is possible to suppress a delay in
overcurrent detection of the semiconductor switching
element 101 due to an electrostatic capacitance being
parasitic in a wiring line.
25 [0081] Note that, although FIG. 8 illustrates an example
in which the first embodiment is applied to the power
converter 500, this similarly applies to a case where the
second and third embodiments are applied.
[0082] To the semiconductor switching element 101 and
30 the voltage dividing circuits 20, 20A, and 20B of the power
converters 500, 500A and 500B described in the first to
third embodiments a high voltage is applied. Therefore,
from the viewpoint of insulation, it is preferable to
32
dispose the voltage dividing circuits 20, 20A, and 20B and
the overcurrent determination circuit 40 to be separated at
a distance. According to the present disclosure, for
example, even in a case where the overcurrent determination
5 circuit 40 is provided on a substrate different from the
voltage dividing circuits 20, 20A, and 20B and the current
amplifier circuits 30 and 30A, and substrates of these are
separated at a distance, that is, the wiring lines V, C, G,
and S are separated at a distance, it is possible to
10 suppress a delay in overcurrent detection of the
semiconductor switching element 101 due to an electrostatic
capacitance being parasitic in a wiring line. In addition,
by changing a current amplification factor in accordance
with the distance between the substrates, it is possible to
15 reduce variation in delay due to a difference in the
distance between the substrates.
[0083] Further, in the configuration of the first to
third embodiments, since the overcurrent determination
circuit 40 is disposed on a substrate different from
20 substrates on which the voltage dividing circuits 20, 20A,
and 20B and the current amplifier circuits 30 and 30A are
disposed, the substrate can be freely disposed, and the
power converter 500 can be downsized.
[0084] Further, in the configuration of the first to
25 third embodiments, in a case where the semiconductor
switching element 101 is turned on and a voltage reduced by
the voltage dividing circuits 20, 20A, and 20B becomes
equal to or higher than an operation threshold voltage of
the current amplifier circuits 30 and 30A, the current
30 amplifier circuits 30 and 30A amplify a current flowing to
the voltage dividing circuits 20, 20A, and 20B. That is,
the overcurrent detection circuitry 200 operates only when
the semiconductor switching element 101 is in the on state.
33
With the circuit configuration of the present disclosure,
the overcurrent detection circuitry 200 can be easily
configured.
[0085] Note that, in the configuration of the first to
5 third embodiments, as an example, the overcurrent
determination circuit 40 includes the integration circuit
50 and the determination circuit 60, but the present
disclosure is not limited to this. Any circuit may be used
as long as it is determined whether or not the
10 semiconductor switching element is overcurrent on the basis
of the current. For example, the overcurrent determination
circuit 40 may be configured by replacing the capacitor 52
with a resistance element and arranging a control
microcontroller or the like at an output destination of the
15 comparator 62.
[0086] The circuit configuration of the drive circuit 10
of the first to third embodiments is an example, and is not
limited to this as long as the circuit controls on/off of
the semiconductor switching element 101. In addition, the
20 circuit configuration of the current amplifier circuits 30
and 30A is also an example, and is not limited to this as
long as the circuit amplifies and outputs currents
outputted from the voltage dividing circuits 20, 20A, and
20B.
25 [0087] In addition, in the first to third embodiments,
as a material constituting the semiconductor switching
element 101, not only silicon (Si) but also silicon carbide
(SiC), gallium nitride (GaN), gallium oxide (Ga2O3),
diamond, and the like, which are wide band gap
30 semiconductors, may be used. By making the semiconductor
switching element 101 as a self-arc-extinguishing
semiconductor device formed by a wide-gap semiconductor
having a wider band gap than silicon, it is possible to
34
achieve low loss and high-speed switching.
[0088] Although "directly connected" is written in the
present disclosure, it is not necessary to be directly
connected physically as long as the electrical
5 configuration is not changed, and for example, a resistance
element having a resistance value of zero may be
interposed.
[0089] Note that the configuration illustrated in the
above embodiment illustrates one example of the contents of
10 the present disclosure and can be combined with another
known technique, and it is also possible to omit and change
a part of the configuration without departing from the
subject matter of the present disclosure.
15 Reference Signs List
[0090] 10 drive circuit; 11 control circuit; 12 turnon MOSFET; 13 turn-off MOSFET; 14, 15 resistance element;
16 first DC power supply; 17 second DC power supply; 18
positive power supply node; 19 negative power supply node;
20 20, 20A, 20B voltage dividing circuit; 21a, 21b, 21n, 22,
23 resistance element; 24a, 24b, 24n capacitor; 30, 30A
current amplifier circuit; 31, 32 npn transistor; 33, 34
resistance element; 35 pnp transistor; 32a, 35a base;
32b, 35b emitter; 32c, 35c collector; 40 overcurrent
25 determination circuit; 50 integration circuit; 51
resistance element; 52 capacitor; 60 determination
circuit; 61 diode; 62 comparator; 63 DC power supply; 70
connection point; 100 semiconductor module; 101
semiconductor switching element; 102 diode; 103 first
30 electrode; 104 second electrode; 105 control electrode;
110 main conversion circuit; 200 overcurrent detection
circuitry; 300, 300A, 300B drive controller; 400 power
supply; 500, 500A, 500B power converter; 600 load; 700
35
power conversion system.

We Claim :
1. An overcurrent detection circuitry comprising:
a voltage dividing circuit adapted to reduce a voltage
applied to a first electrode and a second electrode of a
5 semiconductor switching element including the first
electrode, the second electrode, and a control electrode;
a current amplifier circuit adapted to amplify and
output a current outputted from the voltage dividing
circuit; and
10 an overcurrent determination circuit adapted to
determine whether or not the semiconductor switching
element is overcurrent, based on a current outputted from
the current amplifier circuit.
15 2. The overcurrent detection circuitry according to claim
1, wherein
the voltage dividing circuit is connected to the first
electrode side of the semiconductor switching element.
20 3. The overcurrent detection circuitry according to claim
1 or 2, wherein
the overcurrent determination circuit is provided on a
substrate different from substrates on which the voltage
dividing circuit and the current amplifier circuit are
25 provided.
4. The overcurrent detection circuitry according to any
one of claims 1 to 3, wherein
in a case where the semiconductor switching element is
30 turned on, and a voltage reduced by the voltage dividing
circuit becomes equal to or higher than an operation
threshold voltage of the current amplifier circuit, the
current amplifier circuit amplifies a current flowing in
37
the voltage dividing circuit.
5. The overcurrent detection circuitry according to any
one of claims 1 to 4, wherein,
5 when the semiconductor switching element is determined
to be an overcurrent, the overcurrent determination circuit
is adapted to transmit a gate interruption signal for
turning off the semiconductor switching element to a drive
circuit, which is adapted to control on/off of the
10 semiconductor switching element.
6. The overcurrent detection circuitry according to claim
5, wherein
the current amplifier circuit is directly connected to
15 a positive power supply voltage of the drive circuit.
7. The overcurrent detection circuitry according to claim
5 or 6, wherein the current amplifier circuit is directly
connected:
20 to the control electrode and the second electrode of
the semiconductor switching element; and
to the drive circuit.
8. The overcurrent detection circuitry according to any
25 one of claims 1 to 7, wherein
a resistance element and a capacitor are connected in
parallel in the voltage dividing circuit.
9. The overcurrent detection circuitry according to any
30 one of claims 1 to 8, wherein
the semiconductor switching element is a self-arcextinguishing semiconductor device formed by a wide-gap
semiconductor having a wider band gap than silicon.
38
10. The overcurrent detection circuitry according to claim
9, wherein the wide-gap semiconductor is any one of silicon
carbide, gallium nitride, and diamond.
5
11. A power converter comprising:
a main conversion circuit including the semiconductor
switching element;
a drive circuit adapted to control on/off of the
10 semiconductor switching element of the main conversion
circuit; and
the overcurrent detection circuitry according to any
one of claims 1 to 10.