PHASE SHIFTING DEVICE
FIELD OFTHE INVENTION
The present invention relates to a phase shifting device.
BACKGROUND
Signal processing devices such a s phase shifting devices are known.
Such signal processing devices typically receive a signal to be processed by the signal
processing device and provide a signal processed by the signal processing device. The
signal processing typically changes the received signal in some way to make it suitable
for onwards transmission. Such signal processing devices may be used in
telecommunications systems and may be required to operate at high frequencies. For
example, signal processing devices such as phase shifters are required to process
signals operating in the gigahertz region.
The choice of phase shifter for a particular application is influenced by many factors;
for example, the amount of phase shift obtainable from the device, the insertion losses
caused by the device and the power handling capability of the device. For lower
power handling capabilities, the variation in phase shift is obtained by using a varactor
and pin diode arrangement to achieve a variation of insertion phase. Although such
phase shifters provide acceptable performance for low power operations, they have
their shortfalls and these shortfalls are compounded in a typical telecommunications
system where each radio frequency system usually requires a great number of such
phase shifters.
Accordingly, it is desired to provide a n improved phase shifting device.
SUMMARY
According to a first aspect, there is provided a phase shifting device as claimed in
claim 1.
The first aspect recognises that the voltage tunability of the dielectric properties of
liquid crystals can be utilised in a phase shifting device and that a phase shifting device
based on liquid crystals may provide a convenient, controllable, accurate and low-cost
device. In principle, liquid crystals are anisotropic dielectric materials which means that
they exhibit different dielectric properties with regard to the direction of the applied
electric or magnetic field. However, the first aspect also recognises that operating
liquid crystal technology in phase shifting devices operating in the low gigahertz region
faces an immediate problem. Generally, the size of the device is comparable to the
wavelength at which the device operates and, as the frequency decreases, the
wavelength increases and so does the size of the radio frequency device. For
example, the free space wavelength at 60 gigahertz (millimetre wave frequency) is 5
mm, whilst the free space wavelength a t 2 gigahertz (S-band) is 150 mm. This infers that
a liquid crystal base phase shifting device operating at 2 gigahertz has a size that is
approximately 30 times greater than its equivalent at 60 gigahertz if direct scaling is
used. Accordingly, when using liquid crystal structures in the design of a reflective type
phase shifter, as shown in Figure , the reflective loads are realised using a liquid crystal
formed electrode that is in effect a resonant microstrip line. As the microstrip line is
formed on a liquid crystal substrate, its length a t a frequency of 60 gigahertz is in the
order of 2 mm. However, when implementing such a structure at lower frequencies (for
example, 2 gigahertz), the length of the microstrip line needs to be approximately 30
times longer. This results in increasingly long microstrip lines which can soon become a
prohibitive length. Hence, the first aspect recognises that it is desirable to provide a
phase shifting device that has a size which is comparable with existing devices without
compromising its performance.
The first aspect recognises that if, rather than using a microstrip structure, a lumped
element equivalent is instead used, then it is possible to exploit the advantages of a
liquid crystal structure but in a more compact form. It will be appreciated that the
characteristics of a microstrip line can be represented by a lumped element equivalent
made up of a network of inductors and capacitors. This enables only the capacitors to
be realised using a liquid crystal substrate while the inductors can be realised using
standard technologies such as, for example, surface mount. In this way, the equivalent
size of the device is significantly reduced and its dimensions may be effectively
determined by the length of the inductors.
Accordingly, a phase shifting device may be provided. The phase shifting device may
comprise a n input which receives a n input signal to be adjusted by the phase shifting
device. A coupling device may be provided which may couple the input with a n
output. The coupling device may also be coupled with at least one lumped equivalent
impedance transformer circuit. The input signal may then be received by the lumped
equivalent impedance transformer circuit. Liquid crystal variable capacitors may be
provided within the lumped equivalent impedance transformer circuit. The liquid
crystal variable capacitors may then adjust the input signal in response to a bias
voltage applied to the liquid crystal variable capacitors and provide that adjusted
input signal to the coupling device a s a n output signal. In this way, it can be seen that
rather than using a microstrip line a s an impedance transformer, a lumped equivalent
circuit may instead be provided.
It will be appreciated that a lumped element equivalent circuit may include a network
of discrete reactive devices providing the equivalent characteristics of a microstrip line.
Some of these reactive devices may be provided by liquid crystal variable capacitors
whose reactance is variable in response to a bias signal applied to those variable
capacitors. Using a lumped equivalent impedance transformer circuit provides a
variability to enable the input signal to be adjusted whilst also enabling a compact
circuit arrangement to be provided.
In one embodiment, the lumped equivalent impedance transformer circuit comprises a
half wavelength lumped equivalent impedance transformer circuit comprising a pair of
inductors in series coupled, at ends thereof, with first, second and third liquid crystal
variable capacitors operable to present both a variable impedance and a variable
effective electrical length to the hybrid coupler in response to the bias voltage to
provide an adjusted phase input signal a s the output signal. Accordingly, the lumped
equivalent impedance transformer circuit may comprise a network of three liquid
crystal variable capacitors coupled with two inductors to provide an equivalent circuit
of discrete components which is equivalent to a half wavelength microstrip line. By
providing a lumped equivalent circuit which is equivalent to a half wavelength
microstrip line, it is possible to readily adjust the phase of the input signal by biasing the
variable capacitors to change their capacitance. Implementing the half wavelength
microstrip line as a lumped equivalent circuit helps to de-couple any direct scaling
relationship between the size of the circuit and its operating frequency. In other words,
a multiple decrease in operating frequency no longer inevitably results in a
corresponding multiple increase in the length of components of the circuit.
In one embodiment, the first and the second liquid crystal variable capacitors have
matching capacitances.
In one embodiment, an absolute value of a reactance of the first and second liquid
crystal variable capacitors matches an absolute value of a reactance of each of the
pair of inductors.
In one embodiment, the second liquid crystal variable capacitor has a capacitance
which is double the capacitance of each of the first and the third liquid crystal variable
capacitors.
In one embodiment, the lumped equivalent impedance transformer circuit comprises
integer multiples of the half wavelength lumped equivalent impedance transformer
circuit.
In one embodiment, the lumped equivalent impedance transformer circuit comprises a
half wavelength lumped equivalent impedance transformer circuit comprising a pair of
liquid crystal variable capacitors in series coupled, at ends thereof, with first, second
and third inductors operable to present both a variable impedance and a variable
effective electrical length to the hybrid coupler in response to the bias voltage to
provide an adjusted phase input signal a s the output signal.
In one embodiment, the pair of liquid crystal variable capacitors have matching
capacitances.
In one embodiment, an absolute value of a reactance of the first and second inductors
matches an absolute value of each of the pair of liquid crystal variable capacitors.
In one embodiment, the second inductor has an inductance which is half the
inductance of each of the first and the third inductors.
In one embodiment, the lumped equivalent impedance transformer circuit comprises
integer multiples of the half wavelength lumped equivalent impedance transformer
circuit.
In one embodiment, the lumped equivalent impedance transformer circuit comprises
at least a pair of the half wavelength lumped equivalent impedance transformer
circuits. Accordingly, depending on the device implementation, it may be necessary
to provide a pair of half wavelength lumped equivalent impedance transformer circuits
in order to enable the coupling device to operate correctly and provide the required
output signal from the input signal.
In one embodiment, the lumped equivalent impedance transformer circuit comprises
at least a pair of the half wavelength lumped equivalent impedance transformer
circuits both coupled in parallel with the hybrid coupler.
In one embodiment, the lumped equivalent impedance transformer circuit comprises
at least a pair of the half wavelength lumped equivalent impedance transformer
circuits coupled by a quarter wave impedance transformer. Accordingly, each
lumped equivalent impedance transformer circuit may comprise two or more half
wavelength lumped equivalent impedance transformer circuits coupled together by a
quarter wave impedance transformer. It will be appreciated that by providing
additional half wavelength lumped equivalent impedance transformer circuits
improves the bandwidth of the device. Furthermore, a n increased phase shift is
possible. However, this can lead to increased insertion losses. This can be addressed by
reducing the amount of phase shift provided by each half wavelength lumped
equivalent impedance transformer circuit, but ensuring that the total phase shift for the
pair is greater than a predetermined amount such as, for example, 90° over as broad a
bandwidth as possible. The reduction of phase shift provided by each half wavelength
equivalent impedance transformer circuit has the consequence of a reduction in
length of the liquid crystal variable capacitors which can reduce device size.
In one embodiment, the lumped equivalent impedance transformer circuit comprises
at least a first pair of the half wavelength lumped equivalent impedance transformer
circuits coupled by a quarter wave impedance transformer and at least a second pair
of the half wavelength lumped equivalent impedance transformer circuits coupled by
a quarter wave impedance transformer, the first pair and the second pair both being
coupled in parallel with the hybrid coupler. Accordingly, first and second pairs of the
half wavelength lumped equivalent impedance transformer circuits may both be
coupled with the hybrid coupler in order to receive and output the appropriate signals.
In one embodiment, the liquid crystal variable capacitors comprise parallel plate liquid
crystal variable capacitors.
In one embodiment, the inductors comprise microstrip lines.
Further particular and preferred aspects of the present invention are set out in the
accompanying independent and dependent claims. Features of the dependent
claims may be combined with features of the independent claims a s appropriate and
in combinations other than those explicitly set out in the claims.
BRIEF DESCRIPTION OF THE DRAWINGS
Embodiments of the present invention will now be described further, with reference to
the drawings in which:
Figure 1 illustrates a reflective type phase shifter;
Figure 2 illustrates a liquid crystal structure;
Figure 3 illustrates a lumped equivalent circuit which corresponds to a half wavelength
resonant microstrip line;
Figures 4 to 6 illustrate alternative lumped equivalent circuits which corresponds to a
half wavelength resonant microstrip line;
Figure 7 illustrates a first phase shifter;
Figures 8a to 8c show a n example implementation of the equivalent lumped circuit
used a s a building block for the phase shifter of Figure 7;
Figure 9 shows a n example implementation of the phase shifter of Figure 7;
Figure 10 shows the differential phase shift of the phase shifter of Figure 7;
Figure 11shows the insertion loss of the phase shifter 100 of Figure 7;
Figure 12 shows the return loss of the phase shifter 100 of Figure 7;
Figures 13a and 13b show a n example implementation of a pair of equivalent lumped
circuits coupled by a quarter wavelength microstrip line transformer used as a building
block to provide a second phase shifter;
Figure 1 shows a n example implementation of the second phase shifter;
Figure 15 shows the differential phase shift of the phase shifter of Figure 14;
Figure 1 shows the insertion loss of the phase shifter 100 of Figure 14; and
Figure 17 shows the return loss of the phase shifter 100 of Figure 14.
DESCRIPTION OF THE EMBODIMENTS
OVERVIEW
Before discussing embodiments in any detail, a n overview of phase shifting devices
according to embodiments will now be described. As mentioned above,
embodiments recognize that phase shifting devices, particularly those operating at
high frequencies (such a s the gigahertz frequencies utilised by wireless
telecommunications equipment) can utilise resonant liquid crystal electrodes as
resonant microstrip lines in order to perform the required signal processing. However, a s
mentioned above, a problem with utilising liquid crystal structures in this way is that a s
the operating frequency of the devices reduces, the length of the resonant liquid
crystal electrodes needs to increase.
As mentioned above. Figure 1 illustrates a phase shifter implemented using a microstrip
line. The phase shifter receives a n input at one input of a hybrid coupler and outputs
the phase shifted output signal from a n output of the hybrid coupler. The input signal is
split and provided with a phase shift to both resonant liquid crystal electrode microstrip
lines. The length of the resonant liquid crystal electrode microstrip line will be
dependent on the frequency of the input signal to be processed. For a 2 gigahertz
signal, the length of the resonant liquid crystal electrode microstrip lines will need to be
around 60 mm. This length increases as the frequency decreases. Applying a bias
voltage to the resonant liquid crystal electrode microstrip lines will cause a phase shift in
the output signal.
Rather than using liquid crystal devices as resonant electrodes, embodiments instead
provide a lumped element equivalent circuit made of discrete reactive components
which has the same characteristics as a resonant microstrip line. The same effect as
the microstrip line can therefore be provided using the lumped equivalent circuit and
the characteristics of the circuit adjusted by adjusting characteristics of the discrete
components. In particular, such lumped equivalent circuits comprise a network of
inductors and capacitors, the capacitors being formed from liquid crystal structures to
provide a variable capacitor whose characteristics can be varied by applying a bias
voltage to the liquid crystal structure.
In this way, it can be seen that a phase shifting device which performs signal processing
using at least one resonant microstrip line may instead be implemented using discrete
devices forming a lumped equivalent circuit which enables changes to the input signal
to be made, simply by varying the bias applied to liquid crystal variable capacitors
provided as at least one of the discrete components within the lumped equivalent
circuit. By avoiding the use of actual resonant microstrip lines, the dimensions of the
device are influenced less by its operating frequency, which is dictated by the
frequency of the input signal. It will be appreciated that such a n approach provides
for a compact and scalable phase shifting device.
Before discussing the phase shifting device arrangements in more detail, a n overview of
liquid crystal devices will now be given. The molecules of the most commonly used
liquid crystal phase, nematic, can be treated as elongated rods which orientate
themselves alongside the direction of the applied electric or magnetic field. The
orientation of these elongated molecules with respect to the applied field gives rise to
dielectric anisotropy, as shown in Figure 2. Here, the molecules of a nematic liquid
crystal are contained within a system of two electrodes, conveniently deposited on a
substrate. In this illustrative example, the electrodes are treated with polyimide, which
acts a s a n alignment layer needed to ensure the orientation of the molecules of the
liquid crystal in a pre-defined direction in the absence of a n applied field. This
effectively defines the "ground" or "zero" state of the liquid crystal molecules,
macroscopically characterised by a relative dielectric constant of the liquid crystal
layer , where _L indicates that the direction of the RF electric field is perpendicular
to the direction of the liquid crystal molecules. As the bias voltage (a DC or low
frequency voltage) is increased, the molecules of the liquid crystal orientate themselves
in the direction of the applied field, giving rise to a different dielectric constant of the
liquid crystal layer , where Pindicates that the liquid crystal molecules and the RF
electric field are parallel to each other. Viewed in this way, the dielectric constant of
the liquid crystals is voltage tuneable and the degree of tunability is determined by the
difference between the two relative dielectric constants, referred to as the dielectric
anisotropy, A r = —. At RF frequencies : 2.7 and : 3.2 but it may vary
depending on the type of liquid crystal used. The two-electrode liquid crystal system
shown in Figure 2 is effectively a voltage tuneable parallel plate capacitor with the
C £ r capacitance ratio given by r = max = —- 1.2 .
mi r
LUMPED EQUIVALENT CIRCUITS
The building blocks for the signal processor are lumped equivalent circuits incorporating
liquid crystal variable capacitors which replace half wavelength resonant microstrip
lines used in various configurations. Four example lumped equivalent circuits will now
be described.
EXAMPLE 1
Figure 3 illustrates a lumped equivalent circuit 0 which corresponds to a half
wavelength resonant microstrip line 20. The lumped equivalent element circuit 10 is a
network of reactive discrete devices. In this example, point A illustrated on the network
0 corresponds with point A shown on the half wavelength microstrip device 20, a s
does point B.
The network 10 comprises two inductors and three liquid crystal variable capacitors.
The inductors are arranged in a series between points A and point B. A first capacitor is
provided with one of the inductors in parallel with respect to point A. Likewise, a
capacitor is provided in parallel with the second inductor with respect to point B. A
third capacitor is provided coupled to a node between the first and second inductor.
The inductors have matching inductances. The first and second capacitors have
matching capacitances. The capacitance of the third capacitor is twice that of the
first or second capacitances. The absolute value of the reactance of the first or the
second capacitors matches the absolute value of the reactance of the first or second
inductors.
The capacitors are implemented on a liquid crystal substrate. The inductors can be
implemented using a microstrip line or standard surface mount technology. This
enables the size of the equivalent lumped circuit 0 to be significantly reduced
compared to the resonant liquid crystal electrode microstrip line shown in Figure 1, with
the length of the equivalent lumped circuit 10 effectively being determined by the
length of the inductors.
The phase shift of the equivalent lumped circuit 10 can be achieved by altering the
bias voltage applied to the liquid crystal substrate in which the liquid crystal variable
capacitors are formed.
This equivalent lumped circuit 10 can be used in a liquid crystal based phase shifter,
together with its integer multiples, i.e. « - - , with «=7,2,5....
EXAMPLE 2
An alternative way of obtaining a half-wavelength approximation of the microstrip line
20 is illustrated in Figure 4, which shows an equivalent lumped circuit 10A. In this
arrangement, a short length of a microstrip line is used to represent the lumped
inductors of Figure 3.
This equivalent lumped circuit 10A can be used in a liquid crystal based phase shifter,
together with its integer multiples, i.e. - ^-, with n=l,2,3....
EXAMPLE 3
An alternative way of obtaining a half-wavelength approximation of the microstrip line
20 is illustrated in Figure 5, which shows a n equivalent lumped circuit 10B.
In this example, the equivalent lumped circuit 10B is equivalent to a third integer
multiple of a half-wavelength microstrip line, i.e. n s 's similar to the lumped
equivalent circuit 10 circuit of Figure 3, but with n=3).
This equivalent lumped circuit 10B can be used in a liquid crystal based phase shifter,
together with its integer multiples, i.e. n=l,2,3... .
EXAMPLE 4
An alternative way of obtaining a half-wavelength approximation of the microstrip line
20 is illustrated in Figure 6, which shows a n equivalent lumped circuit IOC. In this
arrangement, a short length of a microstrip line is used to represent the lumped
inductors of Figure 5.
This equivalent lumped circuit IOC can be used in a liquid crystal based phase shifter,
3together with its integer multiples, i.e. n - , n=l,2,3... .
PHASE SHIFTER
EXAMPLE - SINGLE LOAD
Figure 7 illustrates a voltage tuneable reflective circuit operating a s a phase shifter 100
in which a pair of the lumped equivalent circuits 10 show in Figure 3 are coupled with a
hybrid coupler 110 to replace the resonant liquid crystal electrode microstrip lines
shown in Figure 1. Although the lumped equivalent circuit of Figure 3 is used as a
building block for this phase shifter 100, it will be appreciated that other of the lumped
equivalent circuits could b e used.
Figures 8a to 8c show a n example implementation of the equivalent lumped circuit 10
which is used a s a building block for the phase shifter 100.
As can be seen in Figures 8a to 8c, the reflection loads are provided on a two-layer
substrate separated by a spacer layer 10 and deposited on a ground plane 120. A
cavity 130 into which a liquid crystal injected is formed by two strips with lengths of Lh
and Le and the top liquid crystal layer cover 140. The height H2 of the spacer layer 110 is
around 101 (as a Rogers duroid material is available in this thickness) but can be
increased up to about 200 without seriously affecting the behaviour of the liquid
crystal molecules.
The reflection load is printed on the bottom surface of the top liquid crystal layer cover
140 and its height should be as small a s possible. Its height H in this example is 50 m
(as a Rogers duroid material is available in this thickness). The height of the reflection
load can have implications. As the liquid crystal layer cover 140 sits directly above the
liquid crystal cavity 130, the choice of material for the reflection load becomes
important. Because the liquid crystal is, when viewed macroscopically, a tuneable
dielectric, its tunability is reduced by the presence of the liquid crystal cover layer 1 0.
This effect is minimised by choosing a low profile, low dielectric constant material for the
liquid crystal cover layer 0; in this case sr =3.48 and H 2 is 50 .
Figure 9 shows an example implementation of the phase shifter 100. As can be seen,
the tuneable variable capacitors are realised on the liquid crystal substrate. Their
dimensions are Lc= 7.25 mm, L.2C = 14.5 mm and W = 1.5 mm. These dimensions are
selected so that the equivalent characteristic impedance of the microstrip line Z c is
about 20 ohms on a grounded liquid crystal substrate with r =2. . The chosen
characteristic impedance Z c provides a compromise between the amount of phase
shift and insertion losses. The spacing between the capacitors Ls is determined by the
length of the surface mount inductors L and should be greater than or equal to around
1mm to allow a practical realisation of the reflective loads and to prevent coupling
between neighbouring capacitors. The length L h is determined by the size of the
coupler and, in this example, Lh is 2.5 mm. The length Le is not critical and in this design
is 1 mm. Accordingly, a device is provided having a n overall size of around 1 mm by
34 mm.
The simulated performance of the phase shifter 100 is shown in Figures 10 to 12. The
surface mount inductors used in the simulations were represented by the .S2P files
available from the AVX manufacturer's data in order to provide as realistic a
performance of the device as possible.
The performance of the phase shifter 100 was simulated for two cases; the case when
the applied bias voltage is 0 volts and the case when the applied bias voltage is 11
volts. Each voltage bias state is characterised by a set of dielectric properties of the
liquid crystal, the dielectric constant and the loss tangent. For the particular liquid
crystal used in the simulations (commonly referred to a s E7), the value of the relative
dielectric constants at 2 gigahertz, at voltage biases of 0 and 1, are known, but the
values of the loss tangents are not. However, as the values of the loss tangents at
higher frequencies are known (in the region of 30 to 60 gigahertz) they were used
instead of the loss tangents a t 2 gigahertz. This may have, as a consequence,
predicted higher losses than will realistically occur and may lead to a n understatement
of the performance of the device. However, this also forms the worst case scenario of
the insertion loss performance, should the loss tangents at 2 gigahertz not significantly
reduce from their values at 30 gigahertz. In this particular case, the loss tangents at 30
gigahertz are tan(£ )=0.04 (0 V) and tan( ) = 0.02 ( 1 1 V).
Figure 10 shows the differential phase shift of the phase shifter 100 a t 11 V
((peak) srp = 3.18) relative to 0 V [ r = 2.72 ) .
Figure 1 shows the insertion loss of the phase shifter 100 for bias voltage of a) 0 V and
b) 11 V.
Figure 2 shows the return loss of the phase shifter 100 for bias voltage of a ) 0 V and b)
11V.
As can be seen, the these Figures indicate that the phase shifter 100 achieves a 90°
phase shift over a bandwidth of 170 megahertz, with a maximum insertion loss of 5.7 dB.
EXAMPLE 2 - DOUBLE LOAD
Figures 13a and 13b show a n example implementation of a pair of the equivalent
lumped circuits 10' coupled by a quarter wavelength microstrip line transformer 15
which is used a s a building block to provide a phase shifter 100A which provides
increased bandwidth. Although the lumped equivalent circuit of Figure 3 is used as a
building block for this phase shifter 100A, it will be appreciated that other of the lumped
equivalent circuits could be used.
The advantage of this double load configuration is that it can, after some modifications
to the previous arrangement, allow broadband phase shift operation and similar
insertion losses a s the single load arrangement described above. Furthermore, in this
configuration, the phase shift obtained is doubled if the same reflective loads as that
mentioned above are used. However, this also doubles the insertion losses obtained.
Accordingly, to overcome the increased losses, the amount of phase shift provided by
each half wavelength equivalent lumped circuit of the double load is reduced.
However, the total phase shift provided by the double load is selected to be over 90°
for as broad a bandwidth a s possible. The reduction of the phase shift provided by
each equivalent lumped circuit 10' has the consequence of reducing the length of the
liquid crystal formed variable capacitors, a s now the characteristic impedance of the
approximated half wavelength microstrip line Zc is set to be around 75 ohms.
Figure 1 shows a n example implementation of the phase shifter 100A. The length of
the distributed capacitors are Lc = 2.3 mm, L2c = 4.6 mm, while the width W remains 1.5
mm. The separation between the loads LSP = 1.9 mm. The two loads are separated by
a quarter wavelength transformer microstrip 15 having a characteristic impedance Zc
of around 100 ohms which is needed for broadband, high phase shift operation. The
quarter wavelength transformer microstrip 15 is meandered onto the substrate a s
indicated in Figures 13a, 13b and 14. The overall dimensions of the phase shifter 100A in
this example are approximately 32 x 1 mm. All other dimensions are otherwise the
same a s the embodiment mentioned above.
The simulated performance of the phase shifter 100A is illustrated in Figures 15 to 7. As
with the embodiment mentioned above, higher loss tangents are assumed for the liquid
crystal used at 2 gigahertz.
Figure 15 shows the differential phase shift of the phase shifter 100A at 11 V
((peak) = 3.18) relative to OV [er =2 2) .
Figure 16 shows the insertion loss of the phase shifter 100A for bias voltage of a ) 0 V and
b) 11V.
Figure 17 shows the return loss of the phase shifter 100A for bias voltage of a ) 0 V and b)
11V.
As can be seen, the simulations indicate that the device achieves a 90° phase shift over
a bandwidth of approximately 370 megahertz, with a maximum insertion loss of 6.4 dB.
It will be appreciated that the reflection load can be made to have more than two
loads and the bandwidth may be increased even further. Also, in order to increase the
power handling capability of the liquid crystal based devices, a four-way hybrid
coupler can be used which would increase power handling by 3 dB. Furthermore, the
power handling of the device can be increased by extending the number of inductors
and capacitors within each equivalent circuit.
The functions of the various elements shown in the Figures, including any functional
blocks labelled as "processors" or "logic", may be provided through the use of
dedicated hardware a s well as hardware capable of executing software in association
with appropriate software. When provided by a processor, the functions may be
provided by a single dedicated processor, by a single shared processor, or by a
plurality of individual processors, some of which may be shared. Moreover, explicit use
of the term "processor" or "controller" or "logic" should not be construed to refer
exclusively to hardware capable of executing software, and may implicitly include,
without limitation, digital signal processor (DSP) hardware, network processor,
application specific integrated circuit (ASIC), field programmable gate array (FPGA),
read only memory (ROM) for storing software, random access memory (RAM), and non
volatile storage. Other hardware, conventional and/or custom, may also be included.
Similarly, any switches shown in the Figures are conceptual only. Their function may be
carried out through the operation of program logic, through dedicated logic, through
the interaction of program control and dedicated logic, or even manually, the
particular technique being selectable by the implementer a s more specifically
understood from the context.
It should be appreciated by those skilled in the art that any block diagrams herein
represent conceptual views of illustrative circuitry embodying the principles of the
invention. Similarly, it will be appreciated that any flow charts, flow diagrams, state
transition diagrams, pseudo code, and the like represent various processes which may
be substantially represented in computer readable medium and so executed by a
computer or processor, whether or not such computer or processor is explicitly shown.
The description and drawings merely illustrate the principles of the invention. It will thus
be appreciated that those skilled in the art will be able to devise various arrangements
that, although not explicitly described or shown herein, embody the principles of the
invention and are included within its spirit and scope. Furthermore, all examples recited
herein are principally intended expressly to be only for pedagogical purposes to aid the
reader in understanding the principles of the invention and the concepts contributed
by the inventor(s) to furthering the art, and are to be construed a s being without
limitation to such specifically recited examples and conditions. Moreover, all statements
herein reciting principles, aspects, and embodiments of the invention, as well as
specific examples thereof, are intended to encompass equivalents thereof.
CLAIMS
1. A phase shifting device (100; 100A), comprising:
an input operable to receive an input signal to be adjusted;
a hybrid coupler ( 10) coupling said input with an output; and
at least one reflective load (10; lOA-lOC; 10') comprising a lumped equivalent
impedance transformer circuit also coupled with said hybrid coupler to receive said
input signal, said lumped equivalent impedance transformer circuit having liquid crystal
variable capacitors operable to adjust said input signal in response to a bias voltage
applied thereto and to provide said adjusted input signal to said hybrid coupler as an
output signal.
2. The phase shifting device of claim , wherein said lumped equivalent
impedance transformer circuit comprises a half wavelength lumped equivalent
impedance transformer circuit comprising a pair of inductors in series coupled, at ends
thereof, with first, second and third liquid crystal variable capacitors operable to
present both a variable impedance and a variable effective electrical length to said
hybrid coupler in response to said bias voltage to provide an adjusted phase input
signal as said output signal.
3. The phase shifting device of claim 2, wherein said first and said second liquid
crystal variable capacitors have matching capacitances.
4. The phase shifting device of claim 2 or 3, wherein an absolute value of a
reactance of said first and second liquid crystal variable capacitors matches an
absolute value of a reactance of each of said pair of inductors.
5. The phase shifting device of any one of claims 2 to 4, wherein said second liquid
crystal variable capacitor has a capacitance which is double said capacitance of
each of said first and said third liquid crystal variable capacitors.
6. The phase shifting device of claim 1, wherein said lumped equivalent
impedance transformer circuit comprises a half wavelength lumped equivalent
impedance transformer circuit comprising a pair of liquid crystal variable capacitors in
series coupled, at ends thereof, with first, second and third inductors operable to
present both a variable impedance and a variable effective electrical length to said
hybrid coupler in response to said bias voltage to provide an adjusted phase input
signal as said output signal.
7. The phase shifting device of claim 6, wherein said pair of liquid crystal variable
capacitors have matching capacitances.
8. The phase shifting device of claim 6 or 7, wherein a n absolute value of a
reactance of said first and second inductors matches an absolute value of each of
said pair of liquid crystal variable capacitors.
9. The phase shifting device of any one of claims 6 to 8, wherein said second
inductor has an inductance which is half said inductance of each of said first and said
third inductors.
0 . The phase shifting device of any one of claims 2 to 9, wherein said lumped
equivalent impedance transformer circuit comprises at least a pair of said half
wavelength lumped equivalent impedance transformer circuits.
11. The phase shifting device of any one of claims 2 to 10, wherein said lumped
equivalent impedance transformer circuit comprises at least a pair of said half
wavelength lumped equivalent impedance transformer circuits both coupled in
parallel with said hybrid coupler.
12. The phase shifting device of any one of claims 2 to 11, wherein said lumped
equivalent impedance transformer circuit comprises at least a pair of said half
wavelength lumped equivalent impedance transformer circuits coupled by a quarter
wave impedance transformer (15).
13. The phase shifting device of any one of claims 2 to 11, wherein said lumped
equivalent impedance transformer circuit comprises at least a first pair of said half
wavelength lumped equivalent impedance transformer circuits coupled by a quarter
wave impedance transformer (15) and at least a second pair of said half wavelength
lumped equivalent impedance transformer circuits coupled by a quarter wave
impedance transformer, said first pair and said second pair both being coupled in
parallel with said hybrid coupler.
15. The phase shifting device of any preceding claim, wherein said liquid crystal
variable capacitors comprise parallel plate liquid crystal variable capacitors.