FORM 2
THE PATENTS ACT, 1970
(39 of 1970)
&
THE PATENTS RULES, 2003
COMPLETE SPECIFICATION
[See section 10, Rule 13]
POWER CONVERTING APPARATUS, MOTOR DRIVING APPARATUS, AND
AIR CONDITIONER;
MITSUBISHI ELECTRIC CORPORATION, A CORPORATION ORGANISED
AND EXISTING UNDER THE LAWS OF JAPAN, WHOSE ADDRESS IS 7-3,
MARUNOUCHI 2-CHOME, CHIYODA-KU, TOKYO 100-8310, JAPAN
THE FOLLOWING SPECIFICATION PARTICULARLY DESCRIBES THE
INVENTION AND THE MANNER IN WHICH IT IS TO BE PERFORMED.
2
DESCRIPTION
POWER CONVERTING APPARATUS, MOTOR DRIVING APPARATUS, AND
AIR CONDITIONER
5
Field
[0001] The present invention relates to a power
converting apparatus that converts an alternating-current
power supplied from an alternating-current power supply
10 into a direct-current power, and a motor driving apparatus
and an air conditioner that include the power converting
apparatus.
Background
15 [0002] A power supply current that is a current supplied
from a power supply includes a harmonic current. The
harmonic current is a frequency component with a frequency
higher than the frequency of a fundamental wave. In order
to reduce failures caused by a harmonic current,
20 international restrictions are imposed on electronic
devices generating harmonic currents. In compliance with
the restrictions, measures for reducing harmonic currents
included in power supply currents by chopping of an
alternating current (AC) or a direct current (DC) are taken
25 in converters.
[0003] Among such converters, bridgeless converters in
which a rectifier circuit is constituted by switching
elements have been actively examined as a technology for
reducing losses by using the AC chopping technology. A
30 direct-current power supply device, which is an example of
the bridgeless converters, described in Patent Literature 1
includes a first arm constituted by an upper diode and a
lower diode connected in series to each other, a second arm
3
constituted by an upper switching element and a lower
switching element connected in series to each other, and a
direct-current power supply for driving the second arm.
The direct-current power supply device described in Patent
Literature 1 also includes a first drive 5 circuit that uses
a voltage output from the direct-current power supply as a
power supply voltage to generate a driving signal for
driving the lower switching element of the second arm, a
bootstrap circuit that uses the voltage output from the
10 direct-current power supply to generate a voltage for
driving the upper switching element of the second arm, and
a second drive circuit that uses the voltage output from
the bootstrap circuit as a power supply voltage to generate
a driving signal for driving the upper switching element of
15 the second arm. Hereinafter, the drive circuits will be
referred to as driving circuits. In addition, hereinafter,
the upper switching element of the second arm will be
simply referred to as an upper switching element, and the
lower switching element of the second arm will be simply
20 referred to as a lower switching element.
[0004] The bootstrap circuit is constituted by a
resistor, a diode, and a capacitor. In the technology
described in Patent Literature 1, when the lower switching
element is turned ON, a closed circuit is formed by the
25 direct-current power supply, the bootstrap circuit, and the
lower switching element, and the capacitor of the bootstrap
circuit is thus charged by the direct-current power supply.
In this process, in addition to the voltage of the directcurrent
power supply, a forward voltage of the body diode
30 formed in the lower switching element of the second arm is
also applied to the capacitor. The capacitor voltage of
the charged capacitor is then used as the power supply
voltage for the second driving circuit, and a driving
4
signal for driving the upper switching element is thus
generated in the second driving circuit.
Citation List
5 Patent Literature
[0005] Patent Literature 1: Japanese Patent Application
Laid-open No. 2016-220378
Summary
10 Technical Problem
[0006] In a case where metal-oxide-semiconductor fieldeffect
transistors (MOSFETs) made of wide band gap (WBG)
semiconductors, for example, are used for the switching
elements, a potential barrier of a p-n junction of a WBG
15 semiconductor is higher than that of a silicon (Si)
semiconductor. Thus, a voltage at which a forward current
starts to flow in a body diode formed in a WBG MOSFET is a
value higher than a voltage at which a forward current
starts to flow in a body diode formed in a Si switching
20 element. It can thus be said that the forward current -
forward voltage characteristics of a body diode formed in a
WBG MOSFET is inferior to the forward current - forward
voltage characteristics of a body diode formed in a Si
switching element. In a case where a switching element in
25 which a voltage at which a forward current starts to flows
in a body diode is relatively high as described above is
used for a lower switching element of Patent Literature 1,
the capacitor voltage of the capacitor of the bootstrap
circuit, that is, the power supply voltage for the driving
30 circuits may be higher than a rated voltage of a driving
circuit. When a power supply voltage higher than the rated
voltage of a driving circuit is applied to the driving
circuit in this manner, there is a problem in that a
5
withstand voltage of the driving circuit decreases. The
withstand voltage used herein is a voltage that can be
applied to a driving circuit for a prescribed time without
causing breakdown of the driving circuit. In addition,
because the value of a driving signal 5 generated by a
driving circuit becomes larger as the power supply voltage
for the driving circuit is higher, there is a problem in
that a short circuit withstand of the upper switching
element decreases. The short circuit withstand is defined
10 as a time from when a short-circuit current starts to flow
into the upper switching element until the upper switching
element is damaged.
[0007] The present invention has been made in view of
the above, and an object thereof is to provide a power
15 converting apparatus capable of improving the reliability
by preventing or reducing an increase in a power supply
voltage for driving circuits for switching elements.
Solution to Problem
20 [0008] To solve the aforementioned problems and achieve
the object, a power converting apparatus according to the
present invention is a power converting apparatus for
converting an alternating-current power supplied from an
alternating-current power supply into a direct-current
25 power, and includes: a first line and a second line, each
of the first line and the second line being connected to
the alternating-current power supply; and a first reactor
disposed on the first line. The power converting apparatus
includes: a first arm including a first switching element,
30 a second switching element, and a third line having a first
connection point, the first switching element being
connected to the second switching element in series by the
third line, the first connection point being connected to
6
the first reactor by the first line. The power converting
apparatus includes: a second arm connected in parallel with
the first arm and including a third switching element, a
fourth switching element, and a fourth line having a second
connection point, the third switching 5 element being
connected to the fourth switching element in series by the
fourth line, the second connection point being connected to
the alternating-current power supply by the second line.
The power converting apparatus includes: a first capacitor
10 connected in parallel with the second arm; a first driving
circuit outputting a first driving signal for driving the
first switching element; a bootstrap circuit including a
second capacitor, the second capacitor applying a power
supply voltage for the first driving circuit to the first
15 driving circuit; and a diode adjusting the power supply
voltage, wherein a first voltage is lower than a second
voltage, the first voltage being a voltage at which a
forward current starts to flow in the diode, the second
voltage being a voltage at which a forward current starts
20 to flow in a body diode formed in the second switching
element.
Advantageous Effects of Invention
[0009] The power converting apparatus according to the
25 present invention produces an effect of being capable of
improving the reliability by preventing or reducing an
increase in the power supply voltage for the driving
circuits for the switching elements.
30 Brief Description of Drawings
[0010] FIG. 1 is a diagram illustrating an example of a
configuration of a power converting apparatus according to
a first embodiment.
7
FIG. 2 is a schematic cross-sectional view
illustrating an outline structure of a MOSFET that can be
used as switching elements illustrated in FIG. 1.
FIG. 3 is a first diagram illustrating a path of
current flowing in the power converting apparatus 5 according
to the first embodiment when the absolute value of a power
supply current is larger than a current threshold and a
power supply voltage polarity is positive.
FIG. 4 is a first diagram illustrating a path of
10 current flowing in the power converting apparatus according
to the first embodiment when the absolute value of the
power supply current is larger than the current threshold
and the power supply voltage polarity is negative.
FIG. 5 is a second diagram illustrating a path of
15 current flowing in the power converting apparatus according
to the first embodiment when the absolute value of the
power supply current is larger than the current threshold
and the power supply voltage polarity is positive.
FIG. 6 is a second diagram illustrating a path of
20 current flowing in the power converting apparatus according
to the first embodiment when the absolute value of the
power supply current is larger than the current threshold
and the power supply voltage polarity is negative.
FIG. 7 is a first diagram for explaining an operation
25 that causes a capacitor short circuit via an alternatingcurrent
power supply and a reactor in the power converting
apparatus according to the first embodiment.
FIG. 8 is a second diagram for explaining an operation
that causes a capacitor short circuit via the alternating30
current power supply and the reactor in the power
converting apparatus according to the first embodiment.
FIG. 9 is a first diagram illustrating a path of
current flowing in the power converting apparatus according
8
to the first embodiment when the absolute value of the
power supply current is smaller than the current threshold
and the power supply voltage polarity is positive.
FIG. 10 is a first diagram illustrating a path of
current flowing in the power converting apparatus 5 according
to the first embodiment when the absolute value of the
power supply current is smaller than the current threshold
and the power supply voltage polarity is negative.
FIG. 11 is a second diagram illustrating a path of
10 current flowing in the power converting apparatus according
to the first embodiment when the absolute value of the
power supply current is smaller than the current threshold
and the power supply voltage polarity is positive.
FIG. 12 is a second diagram illustrating a path of
15 current flowing in the power converting apparatus according
to the first embodiment when the absolute value of the
power supply current is smaller than the current threshold
and the power supply voltage polarity is negative.
FIG. 13 is a diagram illustrating an example of a
20 configuration of a control unit of the power converting
apparatus according to the first embodiment.
FIG. 14 is a chart illustrating an example of a power
supply voltage, and a power supply voltage phase estimation
value and a sinusoidal value calculated by a power supply
25 voltage phase calculating unit illustrated in FIG. 13.
FIG. 15 is a diagram illustrating an example of a
configuration of a first pulse generating unit of the power
converting apparatus according to the first embodiment.
FIG. 16 is a chart illustrating an example of a
30 reference ON-duty, a carrier wave, and a reference pulse
width modulation (PWM) signal in FIG. 15.
FIG. 17 is a chart illustrating an example of the
reference PWM signal, an inverted PWM signal, a first PWM
9
signal, and a second PWM signal in FIG. 15.
FIG. 18 is a flowchart illustrating an example of
procedures of a selecting process performed by a pulse
selector of the first pulse generating unit illustrated in
5 FIG. 15.
FIG. 19 is a schematic graph illustrating the relation
of currents flowing through a switching element and a body
diode illustrated in FIG. 1, the loss of the switching
element, and the loss of the body diode.
10 FIG. 20 is a flowchart illustrating an example of
procedures of a process performed by a second pulse
generating unit illustrated in FIG. 13.
FIG. 21 is a flowchart illustrating an example of
procedures for controlling the switching elements on the
15 basis of the power supply current by the second pulse
generating unit illustrated in FIG. 13.
FIG. 22 is a chart illustrating a first example of
signals, corresponding to one cycle of the power supply
voltage, generated in the power converting apparatus
20 according to the first embodiment.
FIG. 23 is a chart illustrating a second example of
signals, corresponding to one cycle of the power supply
voltage, generated in the power converting apparatus
according to the first embodiment.
25 FIG. 24 is a chart illustrating an example of signals
when the power converting apparatus according to the first
embodiment performs simple switching control.
FIG. 25 is a chart illustrating an example of signals
in passive states generated by the power converting
30 apparatus according to the first embodiment.
FIG. 26 is a diagram illustrating driving circuits and
bootstrap circuits included in the power converting
apparatus according to the first embodiment.
10
FIG. 27 is a diagram illustrating an example of a
configuration of a power converting apparatus according to
a first modification of the first embodiment.
FIG. 28 is a diagram illustrating an example of a
configuration of a power converting apparatus 5 according to
a second modification of the first embodiment.
FIG. 29 is a diagram illustrating an example of a
configuration of a power converting apparatus according to
a third modification of the first embodiment.
10 FIG. 30 is a diagram illustrating an example of a
configuration of a power converting apparatus according to
a second embodiment.
FIG. 31 is a diagram illustrating an example of a
hardware configuration implementing the control unit of the
15 first and second embodiments.
FIG. 32 is a diagram illustrating an example of a
configuration of a motor driving apparatus according to a
third embodiment.
FIG. 33 is a diagram illustrating an example of a
20 configuration of an air conditioner according to a fourth
embodiment.
Description of Embodiments
[0011] A power converting apparatus, a motor driving
25 apparatus, and an air conditioner according to certain
embodiments of the present invention will be described in
detail below with reference to the drawings. Note that the
present invention is not limited to the embodiments.
[0012] First Embodiment.
30 FIG. 1 is a diagram illustrating an example of a
configuration of a power converting apparatus according to
a first embodiment. A power converting apparatus 100
according to the first embodiment is a power supply device
11
having an AC-DC converting function for converting an
alternating-current power supplied form a single-phase
alternating-current power supply 1 into a direct-current
power and applying the direct-current power to a load 50.
Hereinafter, the single-phase alternating-5 current power
supply 1 may simply be referred to as an alternatingcurrent
power supply 1. As illustrated in FIG. 1, the
power converting apparatus 100 includes a reactor 2, which
is a first reactor, a bridge circuit 3, a smoothing
10 capacitor 4, which is a first capacitor, a power supply
voltage detecting unit 5, a power supply current detecting
unit 6, a bus voltage detecting unit 7, and a control unit
10.
[0013] The bridge circuit 3 includes a first arm 31,
15 which is a first circuit, and a second arm 32, which is a
second circuit. The first arm 31 includes a switching
element 311 and a switching element 312, which are
connected in series. A body diode 311a is formed in the
switching element 311. The body diode 311a is connected in
20 parallel between a drain and a source of the switching
element 311. A body diode 312a is formed in the switching
element 312. The body diode 312a is connected in parallel
between a drain and a source of the switching element 312.
The body diodes 311a and 312a are each used as a
25 freewheeling diode.
[0014] The second arm 32 incudes a switching element 321
and a switching element 322, which are connected in series.
The second arm 32 is connected in parallel with the first
arm 31. A body diode 321a is formed in the switching
30 element 321. The body diode 321a is connected in parallel
between a drain and a source of the switching element 321.
A body diode 322a is formed in the switching element 322.
The body diode 322a is connected in parallel between a
12
drain and a source of the switching element 322. The body
diodes 321a and 322a are each used as a freewheeling diode.
[0015] Specifically, the power converting apparatus 100
incudes a first line 501 and a second line 502, which are
each connected to the alternating-current 5 power supply 1,
and the reactor 2 disposed on the first line 501. In
addition, the first arm 31 includes the switching element
311, which is a first switching element, the switching
element 312, which is a second switching element, and a
10 third line 503 having a first connection point 506. The
switching element 311 is connected in series to the
switching element 312 by the third line 503. The first
line 501 is connected to the first connection point 506.
The first connection point 506 is connected to the
15 alternating-current power supply 1 via the first line 501
and the reactor 2.
[0016] The second arm 32 includes the switching element
321, which is a third switching element, the switching
element 322, which is a fourth switching element, and a
20 fourth line 504 having a second connection point 508. The
switching element 321 is connected in series to the
switching element 322 by the fourth line 504. The second
line 502 is connected to the second connection point 508.
The second connection point 508 is connected to the
25 alternating-current power supply 1 via the second line 502.
The smoothing capacitor 4, which is a capacitor, is
connected in parallel with the second arm 32.
[0017] MOSFETs formed of WBG semiconductors can be used
for the switching elements 311, 312, 321, and 322. For the
30 WBG semiconductors, gallium nitride (GaN) materials,
silicon carbide (SiC), diamond, or aluminum nitride is used.
Use of the WBG semiconductors for the switching elements
311, 312, 321, and 322 increases the withstand voltage
13
characteristics and also increases the allowable current
density, which allows miniaturization of modules. In
addition, because the WBG semiconductors have high heat
resistance, use of the WBG semiconductors for the switching
elements 311, 312, 321, and 322 allows 5 miniaturization of
radiating fins for radiating heat generated by the
switching elements.
[0018] The control unit 10 generates driving pulses for
causing the switching elements 311, 312, 321, and 322 of
10 the bridge circuit 3 to operate on the basis of signals
output from the power supply voltage detecting unit 5, the
power supply current detecting unit 6, and the bus voltage
detecting unit 7. The power supply voltage detecting unit
5 detects a power supply voltage Vs, which is a voltage
15 output from the alternating-current power supply 1, and
outputs an electrical signal indicating the detection
result to the control unit 10. The power supply current
detecting unit 6 detects a power supply current Is, which
is a current output from the alternating-current power
20 supply 1, and outputs an electrical signal indicating the
detection result to the control unit 10. The bus voltage
detecting unit 7 detects a bus voltage Vdc, and outputs the
detected bus voltage Vdc to the control unit 10. The bus
voltage Vdc is a voltage obtained by smoothing a voltage
25 output from the bridge circuit 3 by the smoothing capacitor
4.
[0019] Next, basic operation of the power converting
apparatus 100 according to the first embodiment will be
described. Hereinafter, the switching elements 311 and 321
30 connected to the positive side of the alternating-current
power supply 1, that is, a positive terminal of the
alternating-current power supply 1 may also be referred to
as upper switching elements. In addition, the switching
14
elements 312 and 322 connected to the negative side of the
alternating-current power supply 1, that is, a negative
terminal of the alternating-current power supply 1 may also
be referred to as lower switching elements.
[0020] In the first arm 31, the upper 5 switching element
and the lower switching element operate complementarily.
Specifically, when one of the upper switching element and
the lower switching element is ON, the other is OFF. The
switching elements 311 and 312 constituting the first arm
10 31 are driven by driving signals output from driving
circuits, which will be described later. The driving
circuits amplify PWM signals generated by the control unit
10, and output the amplified signals as driving signals.
The operations of turning the switching elements ON or OFF
15 in accordance with driving signals will hereinafter also be
referred to as switching operations.
[0021] The switching elements 321 and 322 constituting
the second arm 32 perform operations in accordance with
driving signals to be turned ON or OFF in a manner similar
20 to the switching elements 311 and 312. Basically, the
switching elements are turned ON or OFF depending on a
power supply voltage polarity that is the polarity of
voltage output from the alternating-current power supply 1.
Specifically, when the power supply voltage polarity is
25 positive, the switching element 322 is ON and the switching
element 321 is OFF, and when the power supply voltage
polarity is negative, the switching element 321 is ON and
the switching element 322 is OFF. In the first embodiment,
however, as will be described later, in order to prevent a
30 short circuit of the smoothing capacitor 4 via the
alternating-current power supply 1 and the reactor 2, the
switching element 322 and the switching element 321 are
both OFF when the absolute value of the power supply
15
current Is output from the alternating-current power supply
1 is equal to or smaller than a threshold. Alternatively,
in order to prevent a short circuit of the smoothing
capacitor 4 via the alternating-current power supply 1 and
the reactor 2, the switching element 312 5 and the switching
element 311 may both be OFF when the absolute value of the
power supply current Is output from the alternating-current
power supply 1 is equal to or smaller than a threshold.
Hereinafter, the threshold to be compared with the absolute
10 value of the power supply current Is will be referred to as
a current threshold. In addition, hereinafter, the short
circuit of the smoothing capacitor 4 will be referred to as
a capacitor short circuit. The capacitor short circuit is
a state in which the energy stored in the smoothing
15 capacitor 4 is released and the current is regenerated back
to the alternating-current power supply 1.
[0022] Next, the relation between the states of the
switching elements in the first embodiment and the path of
current flowing in the power converting apparatus 100
20 according to the first embodiment will be explained. Note
that the structure of the MOSFETs will be described with
reference to FIG. 2 before the explanation.
[0023] FIG. 2 is a schematic cross-sectional view
illustrating an outline structure of a MOSFET that can be
25 used as the switching elements illustrated in FIG. 1. FIG.
2 illustrates an n-type MOSFET as an example. In an n-type
MOSFET, a p-type semiconductor substrate 600 is used as
illustrated in FIG. 2. A source electrode S, a drain
electrode D, and a gate electrode G are formed on the
30 semiconductor substrate 600. High-concentration impurity
is introduced by ion implantation into portions in contact
with the source electrode S and the drain electrode D to
form n-type regions 601. In addition, an insulating oxide
16
layer 602 is formed between a portion of the semiconductor
substrate 600 where no n-type region 601 is formed and the
gate electrode G. Thus, the insulating oxide layer 602 is
present between the gate electrode G and a p-type region
603 of the semiconductor 5 substrate 600.
[0024] When a positive voltage is applied to the gate
electrode G, electrons are attracted to an interface
between the p-type region 603 and the insulating oxide
layer 602 of the semiconductor substrate 600, and the
10 interface is negatively charged. The electron density of a
portion where electrons have gathered becomes higher than a
hole density, and the portion becomes n-type. The portion
that has become n-type functions as a current path, and
will be referred to as a channel 604. The channel 604 is
15 an n-type channel in the example of FIG. 2. When the
MOSFET is controlled to be ON, more current flows to the
channel 604 than to a body diode formed in the p-type
region 603.
[0025] FIGS. 3 to 6 illustrate current paths in the
20 power converting apparatus 100 according to the first
embodiment when the absolute value of the power supply
current Is is larger than the current threshold.
[0026] FIG. 3 is a first diagram illustrating a path of
current flowing in the power converting apparatus according
25 to the first embodiment when the absolute value of the
power supply current is larger than the current threshold
and the power supply voltage polarity is positive. In FIG.
3, the power supply voltage polarity is positive, the
switching element 311 and the switching element 322 are ON,
30 and the switching element 312 and the switching element 321
are OFF. In this state, current flows in the order of the
alternating-current power supply 1, the reactor 2, the
switching element 311, the smoothing capacitor 4, the
17
switching element 322, and the alternating-current power
supply 1. Thus, in the first embodiment, a synchronous
rectification operation is performed in such a manner that
current flows through each of the channels of the switching
element 311 and the switching element 5 322 instead of
flowing through the body diode 311a and the body diode 322a.
[0027] FIG. 4 is a first diagram illustrating a path of
current flowing in the power converting apparatus according
to the first embodiment when the absolute value of the
10 power supply current is larger than the current threshold
and the power supply voltage polarity is negative. In FIG.
4, the power supply voltage polarity is negative, the
switching element 312 and the switching element 321 are ON,
and the switching element 311 and the switching element 322
15 are OFF. In this state, current flows in the order of the
alternating-current power supply 1, the switching element
321, the smoothing capacitor 4, the switching element 312,
the reactor 2, and the alternating-current power supply 1.
Thus, in the first embodiment, synchronous rectification
20 operation is performed in such a manner that current flows
through each of the channels of the switching element 321
and the switching element 312 instead of flowing through
the body diode 321a and the body diode 312a.
[0028] FIG. 5 is a second diagram illustrating a path of
25 current flowing in the power converting apparatus according
to the first embodiment when the absolute value of the
power supply current is larger than the current threshold
and the power supply voltage polarity is positive. In FIG.
5, the power supply voltage polarity is positive, the
30 switching element 312 and the switching element 322 are ON,
and the switching element 311 and the switching element 321
are OFF. In this state, current flows in the order of the
alternating-current power supply 1, the reactor 2, the
18
switching element 312, the switching element 322, and the
alternating-current power supply 1, and a power supply
short-circuit path that does not pass through the smoothing
capacitor 4 is thus formed. Thus, in the first embodiment,
the power supply short-circuit path is 5 formed in such a
manner that current flows through each of the channels of
the switching element 312 and the switching element 322
instead of flowing through the body diode 312a and the body
diode 322a.
10 [0029] FIG. 6 is a second diagram illustrating a path of
current flowing in the power converting apparatus according
to the first embodiment when the absolute value of the
power supply current is larger than the current threshold
and the power supply voltage polarity is negative. In FIG.
15 6, the power supply voltage polarity is negative, the
switching element 311 and the switching element 321 are ON,
and the switching element 312 and the switching element 322
are OFF. In this state, current flows in the order of the
alternating-current power supply 1, the switching element
20 321, the switching element 311, the reactor 2, and the
alternating-current power supply 1, and a power supply
short-circuit path that does not pass through the smoothing
capacitor 4 is formed. Thus, in the first embodiment, the
power supply short-circuit path is formed in such a manner
25 that current flows through each of the channels of the
switching element 311 and the switching element 321 instead
of flowing through the body diode 311a and the body diode
321a.
[0030] The control unit 10 can control the values of the
30 power supply current Is and the bus voltage Vdc by
controlling switching of the current paths described above.
[0031] When the switching element 311 and the switching
element 322 are turned ON while the power supply current Is
19
is not flowing, however, a capacitor short circuit via the
alternating-current power supply 1 and the reactor 2 occurs.
As a result, current flows in a direction opposite to the
normal direction, which may cause such problems as
degradation in power factor, increase 5 in harmonic
components, damage to an element due to overcurrent, or
increase in loss.
[0032] FIGS. 7 and 8 illustrate states in which a
capacitor short circuit via the alternating-current power
10 supply 1 and the reactor 2 occurs.
[0033] FIG. 7 is a first diagram for explaining an
operation that causes a capacitor short circuit via the
alternating-current power supply and the reactor in the
power converting apparatus according to the first
15 embodiment. FIG. 7 illustrates a state in which the power
supply voltage polarity is positive, and the power supply
current Is does not flow. Because the power supply voltage
polarity is positive, current should normally flow in the
order of the alternating-current power supply 1, the
20 reactor 2, the switching element 311, the smoothing
capacitor 4, the switching element 322, and the
alternating-current power supply 1 as illustrated in FIG. 3.
When the switching element 311 and the switching element
322 are turned ON while the power supply current Is is not
25 flowing, however, current flows in the direction opposite
to the normal direction and a capacitor short circuit thus
occurs as illustrated in FIG. 7. Thus, the energy stored
in the smoothing capacitor 4 is output to the alternatingcurrent
power supply 1.
30 [0034] FIG. 8 is a second diagram for explaining an
operation that causes a capacitor short circuit via the
alternating-current power supply and the reactor in the
power converting apparatus according to the first
20
embodiment. FIG. 8 illustrates a state in which the power
supply voltage polarity is negative, and the power supply
current Is does not flow. Because the power supply voltage
polarity is negative, current should normally flow in the
order of the alternating-current power 5 supply 1, the
switching element 321, the smoothing capacitor 4, the
switching element 312, the reactor 2, and the alternatingcurrent
power supply 1 as illustrated in FIG. 4. When the
switching element 312 and the switching element 321 are
10 turned ON in the case where the power supply current Is is
not flowing, however, current flows in the direction
opposite to the normal direction and a capacitor short
circuit occurs as illustrated in FIG. 8.
[0035] In order to prevent a capacitor short circuit,
15 the power converting apparatus 100 according to the first
embodiment permits the switching elements 321 and 322 to be
in the ON state when the absolute value of the power supply
current Is is equal to or larger than the current threshold,
and turns the switching elements 321 and 322 OFF when the
20 absolute value of the power supply current Is is smaller
than the threshold. This enables prevention of a capacitor
short circuit via the alternating-current power supply 1
and the reactor 2, and can achieve a highly reliable power
converting apparatus.
25 [0036] FIGS. 9 to 12 illustrate current paths in the
power converting apparatus 100 according to the first
embodiment when the absolute value of the power supply
current Is is smaller than the current threshold.
[0037] FIG. 9 is a first diagram illustrating a path of
30 current flowing in the power converting apparatus according
to the first embodiment when the absolute value of the
power supply current is smaller than the current threshold
and the power supply voltage polarity is positive. In FIG.
21
9, the power supply voltage polarity is positive, the
switching element 311 is ON, and the switching element 312,
the switching element 321, and the switching element 322
are OFF. In this case, the body diode 322a of the
switching element 322 functions as a freewheeling 5 diode,
and current flows in the order of the alternating-current
power supply 1, the reactor 2, the switching element 311,
the smoothing capacitor 4, the body diode 322a, and the
alternating-current power supply 1 as illustrated in FIG. 9.
10 Note that it is sufficient if the absolute value of the
power supply current Is is such a value that does not cause
malfunctions, and as the absolute value is smaller, the
synchronous rectification period is longer and conduction
loss can be reduced more effectively. In addition, when
15 the absolute value of the power supply current Is is such a
small value that does not require the synchronous
rectification operation, the switching element 311 may be
turned OFF. When the switching element 311 is turned OFF,
no gate driving power of the switching element 311 is
20 generated, which can reduce power consumption for
generating driving signals as compared with a case where
the synchronous rectification operation is performed. Note
that details of the driving circuits that generate driving
signals will be described later.
25 [0038] FIG. 10 is a first diagram illustrating a path of
current flowing in the power converting apparatus according
to the first embodiment when the absolute value of the
power supply current is smaller than the current threshold
and the power supply voltage polarity is negative. In FIG.
30 10, the power supply voltage polarity is negative, the
switching element 312 is ON, and the switching element 311,
the switching element 321, and the switching element 322
are OFF. In this case, the body diode 321a of the
22
switching element 321 functions as a freewheeling diode,
and current flows in the order of the alternating-current
power supply 1, the body diode 321a, the smoothing
capacitor 4, the switching element 312, the reactor 2, and
the alternating-current power supply 1 5 as illustrated in
FIG. 10. Note that it is sufficient if the absolute value
of the power supply current Is is such a value that does
not cause malfunctions, and as the absolute value is
smaller, the synchronous rectification period is longer and
10 conduction loss can be reduced more effectively. In
addition, when the absolute value of the power supply
current Is is such a small value that does not require the
synchronous rectification operation, the switching element
312 may be turned OFF. When the switching element 312 is
15 turned OFF, no gate driving power of the switching element
312 is generated, which can reduce power consumption for
generating driving signals as compared with a case where
the synchronous rectification operation is performed.
[0039] FIG. 11 is a second diagram illustrating a path
20 of current flowing in the power converting apparatus
according to the first embodiment when the absolute value
of the power supply current is smaller than the current
threshold and the power supply voltage polarity is positive.
In FIG. 11, the power supply voltage polarity is positive,
25 the switching element 312 is ON, and the switching element
311, the switching element 321, and the switching element
322 are OFF. In this case, the body diode 322a of the
switching element 322 functions as a freewheeling diode,
and current flows in the order of the alternating-current
30 power supply 1, the reactor 2, the switching element 312,
the body diode 322a, and the alternating-current power
supply 1 as illustrated in FIG. 11. Note that, in this
case, because a short-circuit current flows, even when the
23
absolute value of the power supply current Is is smaller
than the current threshold, the switching element 322 may
be turned ON at the same time when the switching element
312 is turned ON. In this case, because a drop voltage due
to an ON-resistance of the switching element 5 322 is smaller
than a forward voltage of the body diode 322a, the
conduction loss at the switching element 322 is reduced.
[0040] FIG. 12 is a second diagram illustrating a path
of current flowing in the power converting apparatus
10 according to the first embodiment when the absolute value
of the power supply current is smaller than the current
threshold and the power supply voltage polarity is negative.
In FIG. 12, the power supply voltage polarity is negative,
the switching element 311 is ON, and the switching element
15 312, the switching element 321, and the switching element
322 are OFF. In this case, the body diode 321a of the
switching element 321 functions as a freewheeling diode,
and current flows in the order of the alternating-current
power supply 1, the body diode 321a, the switching element
20 311, the reactor 2, and the alternating-current power
supply 1 as illustrated in FIG. 12. Note that, in this
case, because a short-circuit current flows, even when the
absolute value of the power supply current Is is smaller
than the current threshold, the switching element 321 may
25 be turned ON at the same time when the switching element
311 is turned ON. In this case, because a drop voltage due
to an ON-resistance of the switching element 321 is smaller
than a forward voltage of the body diode 321a, the
conduction loss at the switching element 321 is reduced.
30 [0041] Next, a configuration of the control unit 10 of
the power converting apparatus 100 according to the first
embodiment will be described. FIG. 13 is a diagram
illustrating an example of the configuration of the control
24
unit of the power converting apparatus according to the
first embodiment. As illustrated in FIG. 13, the control
unit 10 includes a power supply current command value
control unit 21, an ON-duty control unit 22, a power supply
voltage phase calculating unit 23, a first 5 pulse generating
unit 24, a second pulse generating unit 25, a current
command value calculating unit 26, and an instantaneous
value command value calculating unit 27.
[0042] The power supply current command value control
10 unit 21 calculates an effective current value command value
Is_rms* from a bus voltage Vdc detected by the bus voltage
detecting unit 7 and a bus voltage command value Vdc*. The
bus voltage command value Vdc* may be set in advance or may
be input from outside of the power converting apparatus 100.
15 The power supply current command value control unit 21
calculates the effective current value command value
Is_rms* by proportional-integral control based on a
difference between the bus voltage Vdc and the bus voltage
command value Vdc*.
20 [0043] The current command value calculating unit 26
converts the effective current value command value Is_rms*
into a sinusoidal command value, and outputs the sinusoidal
command value. The instantaneous value command value
calculating unit 27 calculates a power supply current
25 instantaneous value command value Is* by using the
effective current value command value Is_rms* calculated by
the current command value calculating unit 26, and a
sinusoidal value sinθ^
s calculated by the power supply
voltage phase calculating unit 23.
30 [0044] The ON-duty control unit 22 performs
proportional-integral control on a deviation between the
power supply current instantaneous value command value Is*
calculated by the instantaneous value command value
25
calculating unit 27 and the power supply current Is
detected by the power supply current detecting unit 6 to
calculate a reference ON-duty duty of the switching
elements 311 and 312.
[0045] The power supply voltage phase 5 calculating unit
23 calculates a power supply voltage phase estimation value
θ^
s and the sinusoidal value sinθ^
s by using the power
supply voltage Vs detected by the power supply voltage
detecting unit 5. FIG. 14 is a chart illustrating an
10 example of the power supply voltage, and the power supply
voltage phase estimation value and the sinusoidal value
calculated by the power supply voltage phase calculating
unit illustrated in FIG. 13. FIG. 14 illustrates the power
supply voltage Vs, the power supply voltage phase
estimation value θ^
s, and the sinusoidal value sinθ^
15 s in
this order from the top.
[0046] The power supply voltage phase calculating unit
23 linearly increases the power supply voltage phase
estimation value θ^
s, detects a timing at which the power
20 supply voltage Vs changes from the negative polarity to the
positive polarity, and resets the power supply voltage
phase estimation value θ^
s to 0 at the timing. As a result,
under an ideal condition with no control delay and no
detection delay, the power supply voltage phase estimation
value θ^
25 s becomes 360°, that is, 0° at the timing when the
power supply voltage Vs is switched from the negative
polarity to the positive polarity. The power supply
voltage phase calculating unit 23 calculates the sinusoidal
value sinθ^
s on the basis of the calculated power supply
voltage phase estimation value θ^
30 s. Note that, in a case of
implementing resetting of the power supply voltage phase
estimation value θ^
s by using an interrupt function of a
microcomputer, the power supply voltage phase calculating
26
unit 23 resets the power supply voltage phase estimation
value θ^
s by using a signal output from a zero crossing
detecting circuit as an interrupt signal. The zero
crossing detecting circuit is a circuit that detects a
timing at which the power supply voltage 5 Vs switches from
the negative polarity to the positive polarity. Note that
the method for calculating the power supply voltage phase
estimation value θ^
s is not limited to the example
described above, and any method may be used therefor.
10 [0047] FIG. 15 is a diagram illustrating an example of a
configuration of the first pulse generating unit of the
power converting apparatus according to the first
embodiment. The first pulse generating unit 24 includes a
carrier generating unit 241, a reference PWM generating
15 unit 242, a dead time generating unit 243, and a pulse
selector 244.
[0048] The carrier generating unit 241 generates a
carrier wave carry, which is a carrier signal. The carrier
wave carry is used for generation of a reference PWM signal
20 Scom. An example of the carrier wave carry can be a
triangular wave with a peak value “1” and a trough value
“0”. The reference PWM signal Scom is a signal that is a
reference of PWM signals used for driving the switching
elements 311, 312, 321, and 322. As described above, in
25 the first embodiment, complementary PWM control is assumed,
in which a reference PWM signal is used for driving one of
the switching elements of the first arm 31, and a PWM
signal complementary to the reference PWM signal is used
for the other of the switching elements of the first arm 31.
30 [0049] The reference PWM generating unit 242 compares
the magnitudes of the reference ON-duty duty calculated by
the ON-duty control unit 22 illustrated in FIG. 13 and the
carrier wave carry to generate the reference PWM signal
27
Scom. FIG. 16 is a chart illustrating an example of the
reference ON-duty, the carrier wave, and the reference PWM
signal in FIG. 15. As illustrated in FIG. 16, the
reference PWM generating unit 242 generates the reference
PWM signal Scom in such a manner that 5 the reference PWM
signal Scom has a value representing ON in the case where
reference ON-duty duty > carrier wave carry, and that the
reference PWM signal Scom has a value representing OFF in
the case where reference ON-duty duty ≤ carrier wave carry.
10 FIG. 16 illustrates a high active reference PWM signal Scom
as an example. The high active reference PWM signal Scom
is a signal with a high level representing ON and a low
level representing OFF. Note that the signal generated by
the reference PWM generating unit 242 is not limited to a
15 high active reference PWM signal Scom, and may be a low
active reference PWM signal Scom. The low active reference
PWM signal Scom is a signal with a high level representing
OFF and a low level representing ON.
[0050] The description refers back to FIG. 15, in which
20 the dead time generating unit 243 generates a first PWM
signal Sig1 and a second PWM signal Sig2, which are two
complementary signals, on the basis of the reference PWM
signal Scom, and outputs the first PWM signal Sig1 and the
second PWM signal Sig2. Specifically, the dead time
25 generating unit 243 generates an inverted PWM signal Scom'
that is a signal obtained by inverting the reference PWM
signal Scom. The dead time generating unit 243 then
generates the first PWM signal Sig1 and the second PWM
signal Sig2 by setting a dead time in the reference PWM
30 signal Scom and the inverted PWM signal Scom'.
[0051] Specifically, the dead time generating unit 243
generates the first PWM signal Sig1 and the second PWM
signal Sig2 such that the first PWM signal Sig1 and the
28
second PWM signal Sig2 both have a value representing OFF
during the dead time. In one example, the dead time
generating unit 243 makes the first PWM signal Sig1
identical to the reference PWM signal Scom. In addition,
the dead time generating unit 243 generates 5 the second PWM
signal Sig2 by changing a signal value of the inverted PWM
signal Scom' from a value representing ON to a value
representing OFF during the dead time.
[0052] In the case where the inverted PWM signal Scom'
10 is generated by inversion of the reference PWM signal Scom
and two switching elements constituting one arm are
respectively driven by the reference PWM signal Scom and
the inverted PWM signal Scom', there is, ideally, no period
during which two switching elements constituting one arm
15 are ON at the same time. Typically, however, a delay
occurs in a transition from an ON state to an OFF state,
and a delay occurs in a transition from an OFF state to an
ON state. Thus, the delays result in a period during which
two switching elements constituting one arm are ON at the
20 same time, and may cause short-circuit of the two switching
elements constituting one arm. The dead time is a period
set such that two switching elements constituting one arm
are not on at the same time even when a delay in a state
transition occurs. During the dead time, two PWM signals
25 for driving the two switching elements constituting one arm
are both set to a value representing OFF.
[0053] FIG. 17 is a chart illustrating an example of the
reference PWM signal, the inverted PWM signal, the first
PWM signal, and the second PWM signal in FIG. 15. FIG. 17
30 illustrates the reference PWM signal Scom, the inverted PWM
signal Scom', the first PWM signal Sig1, and the second PWM
signal Sig2 in this order from the top. In FIG. 17, when
the inverted PWM signal Scom' has a value representing ON,
29
the second PWM signal Sig2 has a value representing OFF
during a dead time td. Note that the method for generating
the dead time td described above is an example, the method
for generating the dead time td is not limited to the
above-described example, and any 5 method may be used
therefor.
[0054] The description refers back to FIG. 15, in which
the pulse selector 244 determines which of the driving
circuits for the switching element 311 and the switching
10 element 312 to transmit each of the first PWM signal Sig1
and the second PWM signal Sig2 output from the dead time
generating unit 243. FIG. 18 is a flowchart illustrating
an example of procedures of a selecting process performed
by the pulse selector of the first pulse generating unit
15 illustrated in FIG. 15. The pulse selector 244 first
determines whether or not the polarity of the power supply
voltage Vs is positive, that is Vs>0 (step S1). If the
polarity of the power supply voltage Vs is positive (step
S1: Yes), the pulse selector 244 transmits the first PWM
20 signal Sig1 as pulse_312A to the driving circuit for the
switching element 312, and transmits the second PWM signal
Sig2 as pulse_311A to the driving circuit for the switching
element 311 (step S2). This is because, when the power
supply voltage Vs is positive, the current path is switched
25 between the current path illustrated in FIG. 5 and the
current path illustrated in FIG. 3 by turning OFF or ON of
the switching element 311 and the switching element 312,
that is, the bus voltage Vdc and the power supply current
Is are controlled by switching operation of the switching
30 element 311 and the switching element 312.
[0055] If the polarity of the power supply voltage Vs is
negative (step S1: No), the pulse selector 244 transmits
the first PWM signal Sig1 as pulse_311A to the driving
30
circuit for the switching element 311, and transmits the
second PWM signal Sig2 as pulse_312A to the driving circuit
for the switching element 312 (step S3). This is because,
when the power supply voltage Vs is negative, the current
path is switched between the current path 5 illustrated in
FIG. 6 and the current path illustrated in FIG. 4 by
turning OFF or ON of the switching element 311 and the
switching element 312, that is, the bus voltage Vdc and the
power supply current Is are controlled by switching
10 operation of the switching element 311 and the switching
element 312. The pulse selector 244 repeats the abovedescribed
operation each time the polarity of the power
supply voltage Vs changes.
[0056] As described above, the first pulse generating
15 unit 24 generates pulse_311A that is a signal for driving
the switching element 311 and pulse_312A that is a signal
for driving the switching element 312.
[0057] As described above, because the switching element
311 and the switching element 312 are complementarily
20 controlled, the process of generating the inverted PWM
signal Scom' from the reference PWM signal Scom can be
achieved by using a simple signal inversion process. In
addition, the relation of driving pulse outputs in one
carrier can be made approximately the same regardless of
25 the power supply voltage polarity, and prevention of a
short circuit of the upper and lower arms can be easily
achieved. Stable control can be achieved by simple
processes.
[0058] In addition, in the power converting apparatus
30 100 according to the first embodiment, synchronous
rectification control by the switching elements 311 and 312
of the first arm 31 can be achieved. Thus, in the power
converting apparatus 100 according to the first embodiment,
31
loss can be reduced in a region in which the loss of a
switching element is smaller than the loss of a body diode,
that is, a region in which each of currents flowing through
the switching element and the body diode is small as
illustrated in FIG. 19. Thus, a highly-5 efficient system
can be achieved.
[0059] FIG. 19 is a schematic graph illustrating the
relation of currents flowing through a switching element
and a body diode illustrated in FIG. 1, the loss of the
10 switching element, and the loss of the body diode. The
horizontal axis in FIG. 19 represents a current flowing
through the switching element in the ON state, and a
current flowing through the body diode. The vertical axis
in FIG. 19 represents a loss caused when the current flows
15 through the switching element in the ON state and a loss
caused when the current flows through the body diode. A
solid line depicts the loss characteristics of the body
diode. The loss characteristics of the body diode indicate
the relation between the current flowing through the body
20 diode and the loss caused by the ON-resistance of the body
diode when the current flows. A dotted line depicts the
loss characteristics of the switching element in the ON
state. The loss characteristics indicate the relation
between the current flowing through the carrier of the
25 switching element and the loss caused by the ON-resistance
of the switching element when the current flows. A region
represented by a sign A is a region in which the currents
flowing through the switching element and the body diode
are small. A region represented by a sign B is a region in
30 which the currents flowing through the switching element
and the body diode are large. At the boundary between the
region A and the region B, the currents are equal to a
current value at which the value of the loss caused in the
32
switching element and the value of the loss caused in the
body diode are equal.
[0060] As illustrated in FIG. 19, in the region B in
which the loss of the switching element is higher than the
loss of the body diode, the complementary 5 operation is
stopped, so that an increase in the loss due to the
synchronous rectification control can be prevented or
reduced. Thus, by the control of switching between
performing and not performing the synchronous rectification
10 control depending on the power supply current Is, a highly
efficient system can be achieved in all load regions.
[0061] Note that optimal values according to a driving
condition are present for control parameters used for
computation by the power supply current command value
15 control unit 21 and the ON-duty control unit 22 illustrated
in FIG. 13. The driving condition is expressed by at least
one value of the power supply voltage Vs, the power supply
current Is, and the bus voltage Vdc. For example, it is
desirable that a proportional control gain in the ON-duty
20 control unit 22 change in inverse proportion to the bus
voltage Vdc. This is because, if the value of a control
parameter is constant with respect to a change in in the
driving condition, the control parameter will significantly
deviate from a value suitable for control, and as a result,
25 harmonics of the power supply current Is may increase,
pulsation of the bus voltage Vdc may increase, and powersupply
power factor may decrease. In order to prevent or
reduce such increase in pulsation of the bus voltage Vdc,
decrease in the power-supply power factor, and the like,
30 the power supply current command value control unit 21 and
the ON-duty control unit 22 may hold a calculation formula
or a table for implementing a desired circuit operation,
and adjust a control parameter on the basis of detected
33
information by using the calculation formula or the table.
The configuration to adjust a control parameter on the
basis of detected information makes the control parameter a
value suitable for control, which improves controllability.
Note that the detected information is at 5 least one of the
power supply voltage Vs, the power supply current Is, and
the bus voltage Vdc, or information from which these values
can be estimated, for example. An example of the
information from which the values can be estimated is power
10 information detected by a detector for detecting a power
supplied from the alternating-current power supply 1.
[0062] In addition, while the proportional-integral
control is presented as the computation method used in the
power supply current command value control unit 21 and the
15 ON-duty control unit 22 in the example described above, the
present invention is not limited to this computation method,
and other computation methods may be used and a derivative
term may be added to perform proportional-integralderivative
control. In addition, the computation methods
20 in the power supply current command value control unit 21
and the ON-duty control unit 22 need not be the same
computation method.
[0063] The description refers back to FIG. 13, in which
the second pulse generating unit 25 generates pulse_321A
25 that is a signal for driving the switching element 321 and
pulse_322A that is a signal for driving the switching
element 322 on the basis of the power supply voltage Vs
detected by the power supply voltage detecting unit 5 and
the power supply current Is detected by the power supply
30 current detecting unit 6, and outputs the pulse_321A and
the pulse_322A.
[0064] FIG. 20 is a flowchart illustrating an example of
procedures of a process performed by the second pulse
34
generating unit illustrated in FIG. 13. A basic operation
of the second pulse generating unit 25 is controlling the
ON or OFF states of the switching element 321 and the
switching element 322 depending on the polarity of the
power supply voltage Vs. As illustrated 5 in FIG. 20, the
second pulse generating unit 25 determines whether or not
the polarity of the power supply voltage Vs is positive,
that is, Vs>0 (step S11). If the polarity of the power
supply voltage Vs is positive (step S11: Yes), the second
10 pulse generating unit 25 generates and outputs pulse_321A
and pulse_322A to turn the switching element 321 OFF and
turn the switching element 322 ON (step S12).
[0065] If the polarity of the power supply voltage Vs is
negative (step S11: No), the second pulse generating unit
15 25 generates and outputs pulse_321A and pulse_322A to turn
the switching element 321 ON and turn the switching element
322 OFF (step S13). This enables the synchronous
rectification control, and a highly efficient system can be
achieved as described above.
20 [0066] As described above, however, when the switching
element 311 and the switching element 322 are turned ON
while the power supply current Is is not flowing, a
capacitor short circuit via the alternating-current power
supply 1 and the reactor 2 occurs. Thus, in addition to
25 control of the switching element 311 and the switching
element 322, the power converting apparatus 100 according
to the first embodiment controls the ON or OFF states of
the switching element 321 and the switching element 322 on
the basis of the power supply current Is.
30 [0067] FIG. 21 is a flowchart illustrating an example of
procedures for controlling the switching elements on the
basis of the power supply current by the second pulse
generating unit illustrated in FIG. 13. As illustrated in
35
FIG. 21, it is determined whether or not the absolute value
of the power supply current Is is larger than the current
threshold β (step S21). If the absolute value of the power
supply current Is is larger than the current threshold β
(step S21: Yes), the second pulse 5 generating unit 25
permits the switching element 321 and the switching element
322 to be ON (step S22). When the switching element 321
and the switching element 322 are permitted to be ON, the
ON and OFF states are controlled depending on the polarity
10 of the power supply voltage Vs illustrated in FIG. 20.
[0068] If the absolute value of the power supply current
Is is equal to or smaller than the current threshold β
(step S21: No), the second pulse generating unit 25 does
not permit the switching element 321 and the switching
15 element 322 to be ON (step S23). When the switching
element 321 and the switching element 322 are not permitted
to be ON, the switching element 321 and the switching
element 322 are controlled to be in the OFF state
regardless of the polarity of the power supply voltage Vs
20 illustrated in FIG. 20.
[0069] As a result of the control described above, when
a current larger than the current threshold β flows in the
forward direction in the body diodes of the switching
elements, the switching element 321 and the switching
25 element 322 are turned ON. This enables prevention of a
capacitor short circuit via the alternating-current power
supply 1 and the reactor 2. In addition, the second pulse
generating unit 25 may control the switching element 321
and the switching element 322 by using the polarity of the
30 power supply current Is, that is, the direction in which
the current flows, instead of ON-OFF control depending on
the polarity of the power supply voltage Vs.
[0070] In addition, instead of the process illustrated
36
in FIG. 21, whether or not to permit the switching element
321 and the switching element 322 to be ON may be
determined on the basis of the state of switching control.
When switching is not performed, no current flows in the
switching elements, and thus a timing to 5 enter such a state
is predicted so as not to permit the switching element 321
and the switching element 322 to be ON. Note that, in this
case, the synchronous rectification effect may not be
produced in a state in which passive full-wave
10 rectification, that is, a short-circuit path is not used,
but control can be simply built independently of detection
of a current or a voltage.
[0071] In addition, whether or not to permit the
switching element 321 and the switching element 322 to be
15 ON may be determined on the basis of a difference between
the power supply voltage Vs and the bus voltage Vdc instead
of the process illustrated in FIG. 21. Specifically, if
(power supply voltage - bus voltage) > 0, the switching
element 321 and the switching element 322 are permitted to
20 be ON, and if (power supply voltage - bus voltage) ≤ 0, the
switching element 321 and the switching element 322 are not
permitted to be ON.
[0072] Note that, in the example described above, the
second pulse generating unit 25 selects the switching
25 element to be turned ON from the switching element 321 and
the switching element 322 on the basis of the power supply
voltage polarity, and controls switching element 321 and
the switching element 322 on the basis of the power supply
current Is to prevent a capacitor short circuit. The
30 control, however, is not limited to this example, and the
first pulse generating unit 24 may determine whether or not
to permit the switching elements 311, 312, 321, and 322 to
be ON on the basis of the power supply current Is to
37
prevent a capacitor short circuit, and the second pulse
generating unit 25 may perform switching depending on the
power supply voltage polarity without performing control to
prevent a capacitor short circuit on the switching element
321 and the switching 5 element 322.
[0073] Specifically, in the case where the power supply
voltage Vs is positive, the first pulse generating unit 24
does not permit the switching element 311 to be ON when the
absolute value of the power supply current Is is equal to
10 or smaller than the current threshold β, and permits the
switching element 311 to be ON when the absolute value of
the power supply current Is is larger than the current
threshold β. In contrast, in the case where the power
supply voltage Vs is negative, the first pulse generating
15 unit 24 does not permit the switching element 312 to be ON
when the absolute value of the power supply current Is is
equal to or smaller than the current threshold β, and
permits the switching element 312 to be ON when the
absolute value of the power supply current Is is larger
20 than the current threshold β.
[0074] In addition, while the switching in each of the
arms in each power supply cycle is achieved by the method
of generating complementary PWM signals in the example
described above, the method of generating PWM signals is
25 not limited to this example. Specifically, the control
unit 10 may generate a signal pulse_312A for driving the
switching element 312 when the power supply voltage Vs is
positive, and generate a signal pulse_311A for driving the
switching element 311 when the power supply voltage Vs is
30 negative. In addition, in this case, the control unit 10
may generate PWM signals for driving the switching elements
311 and 312 on the basis of the relation of the power
supply current Is, the power supply voltage Vs, and the bus
38
voltage Vdc. This enables the switching elements 311 and
312 to be turned OFF before the timing at which the power
supply current Is becomes zero, and in this case, a
capacitor short circuit via the alternating-current power
supply 1 and the reactor 2 can be prevented 5 even when the
operations of the switching elements 321 and 322 are
controlled on the basis of the power supply voltage
polarity.
[0075] FIG. 22 is a chart illustrating a first example
10 of signals, corresponding to one cycle of the power supply
voltage, generated in the power converting apparatus
according to the first embodiment. FIG. 22 illustrates an
example of the signals generated by the process explained
with reference to FIG. 20. In FIG. 22, the horizontal axis
15 represents time, and the power supply voltage Vs, the power
supply current Is, a timer set value α and a carrier signal,
a signal for driving the switching element 311, a signal
for driving the switching element 312, a signal for driving
the switching element 321, and a signal for driving the
20 switching element 322 are illustrated in this order from
the top.
[0076] The timer set value α is a command value
associated with the reference ON-duty duty, and changes
with time in a stepwise manner. The timer set value α is a
25 period with the value of each step in the vertical axis
being constant. The reference ON-duty duty associated with
each timer set value α changing in such a stepwise manner
is compared with the carrier wave carry that is the carrier
signal, and the pulse widths of the switching elements 311
30 and 321 are thus determined. The reference ON-duty duty is
small near the zero crossing of the power supply voltage Vs,
and becomes larger as the power supply voltage Vs
approaches its peak value. Note that the dead time is not
39
illustrated in FIG. 22.
[0077] A current threshold (positive) on the positive
side is set to prevent excessive switching operations near
the zero crossing when the power supply current Is changes
from negative to positive. Similarly, a 5 current threshold
(negative) on the negative side is set to prevent excessive
switching operations near the zero crossing when the power
supply current Is changes from positive to negative.
[0078] FIG. 22 illustrates an example of operations for
10 complementarily performing PWM control on the switching
elements 311 and 312, in which the switching element 312 is
a master when the power supply voltage Vs has the positive
polarity and the switching element 311 is a master when the
power supply voltage Vs has the negative polarity. Thus, a
15 reference ON-duty duty of an arc shape that is convex
downward is used when the power supply voltage Vs has the
positive polarity, and a reference ON-duty duty of an arc
shape that is convex downward is also used when the power
supply voltage Vs has the negative polarity.
20 [0079] The switching elements 321 and 322 are switched
ON or OFF depending on the polarity of the power supply
voltage Vs, and are further turned OFF when the absolute
value of the power supply current Is is equal to or smaller
than the current threshold. Note that the power converting
25 apparatus 100 according to the first embodiment may have a
configuration in which the power supply current detecting
unit 6 has a filter or hysteresis to prevent excessive
switching operations near the current thresholds.
Alternatively, the power converting apparatus 100 according
30 to the first embodiment may have a configuration in which
the control unit 10 has a filter to the power supply
current Is or hysteresis to prevent excessive switching
operations near the current thresholds.
40
[0080] FIG. 23 is a chart illustrating a second example
of signals, corresponding to one cycle of the power supply
voltage, generated in the power converting apparatus
according to the first embodiment. In FIG. 23, in a manner
similar to FIG. 22, the horizontal axis 5 represents time,
and the power supply voltage Vs, the power supply current
Is, the timer set value α and the carrier signal, a signal
for driving the switching element 311, a signal for driving
the switching element 312, a signal for driving the
10 switching element 321, and a signal for driving the
switching element 322 are illustrated in this order from
the top.
[0081] FIG. 23 illustrates an example of operations for
complementarily performing PWM control on the switching
15 elements 311 and 312, in which the switching element 312 is
a master when the power supply voltage Vs has the positive
polarity and when the power supply voltage Vs has the
negative polarity. Thus, a reference ON-duty duty of an
arc shape that is convex downward is used when the power
20 supply voltage Vs has the positive polarity, and a
reference ON-duty duty of an arc shape that is convex
upward is used when the power supply voltage Vs has the
negative polarity. In the example of the operations in FIG.
23, the signal pulse_312A for driving the switching element
25 312 is generated when the power supply voltage Vs has the
positive polarity, and the signal pulse_311A for driving
the switching element 311 is generated when the power
supply voltage Vs has the negative polarity.
[0082] In addition, while the example in which the
30 switching elements are controlled by the carrier signals is
presented in FIG. 22 described above, the operations of the
first embodiment are also applicable to simple switching
control in which switching is performed once to several
41
times during a half cycle of the power supply cycle. FIG.
24 is a chart illustrating an example of signals when the
power converting apparatus according to the first
embodiment performs simple switching control. In FIG. 24,
the horizontal axis represents time, and 5 the power supply
voltage Vs, the power supply current Is, the absolute value
|Is| of the power supply current Is, a power supply
polarity signal, a power supply current signal, a signal
for driving the switching element 311, a signal for driving
10 the switching element 312, a signal for driving the
switching element 321, and a signal for driving the
switching element 322 are illustrated in this order from
the top. The power supply polarity signal is a binary
signal that changes with the polarity of the power supply
15 voltage Vs, and is used for controlling the switching
element operations of the switching elements 311 and 312.
The power supply current signal is a binary signal used for
controlling the switching element operations of the
switching elements 321 and 322.
20 [0083] In FIG. 24, three current thresholds are
illustrated. A current threshold on the positive side of
the power supply current Is is a threshold set for a
purpose similar to that of the current threshold (positive)
on the positive side described with reference to FIG. 22.
25 A current threshold on the negative side of the power
supply current Is is a threshold set for a purpose similar
to that of the current threshold (negative) on the negative
side described with reference to FIG. 22. A current
threshold set for the absolute value |Is| of the power
30 supply current Is is a threshold set for changing the value
of the power supply current signal.
[0084] The power supply polarity signal is generated by
detection of the zero crossing of the power supply voltage
42
Vs, and the power supply current signal is generated by
detection of the zero crossing of the power supply current
Is. In this case, when the absolute value |Is| of the
power supply current Is is equal to or smaller than the
current threshold, the power converting 5 apparatus 100
performs control such that the switching element 311 and
the switching element 321 are not ON at the same time and
such that the switching element 312 and the switching
element 322 are not ON at the same time. This enables
10 prevention of a capacitor short circuit.
[0085] In addition, even when the switching elements 311
and 312 are in a passive state in which no switching
operations are performed, the switching element 321 and the
switching element 322 are prevented from being turned ON
15 when the absolute value of the power supply current Is is
equal to or smaller than the current threshold, which
enables prevention of a capacitor short circuit.
[0086] FIG. 25 is a chart illustrating an example of
signals in passive states generated by the power converting
20 apparatus according to the first embodiment. In FIG. 25,
in a manner similar to FIG. 24, the horizontal axis
represents time, and the power supply voltage Vs, the power
supply current Is, the absolute value |Is| of the power
supply current Is, a power supply polarity signal, a power
25 supply current signal, a signal for driving the switching
element 311, a signal for driving the switching element 312,
a signal for driving the switching element 321, and a
signal for driving the switching element 322 are
illustrated in this order from the top. In this case as
30 well, when the absolute value of the power supply current
Is equal to or smaller than the current threshold, the
power converting apparatus 100 performs control such that
the switching element 311 and the switching element 321 are
43
not ON at the same time and such that the switching element
312 and the switching element 322 are not ON at the same
time. This enables prevention of a capacitor short circuit.
[0087] Next, driving circuits and bootstrap circuits for
the switching elements will be described 5 with reference to
FIGS. 26 to 29.
[0088] FIG. 26 is a diagram illustrating driving
circuits and bootstrap circuits included in the power
converting apparatus according to the first embodiment. As
10 illustrated in FIG. 26, the power converting apparatus 100
includes two direct-current voltage sources 300, four
driving circuits 311DC, 312DC, 321DC, and 322DC, and two
bootstrap circuits 401 and 402, in addition to the
configuration illustrated in FIG. 1. While the driving
15 circuits 311DC and 312DC share one direct-current voltage
source 300 and the driving circuits 321DC and 322DC share
the other direct-current voltage source 300 in the power
converting apparatus 100 of FIG. 26, one direct-current
voltage source 300 may be used instead of the two direct20
current voltage sources 300, and the four driving circuits
311DC, 312DC, 321DC, and 322DC may share the one directcurrent
voltage source 300.
[0089] The driving circuit 311DC that is a first driving
circuit converts pulse_311A from the control unit 10 into a
25 first driving signal for driving the switching element 311
by using a voltage output from the bootstrap circuit 401 as
the power supply voltage, and outputs the first driving
signal to the gate of the switching element 311. Details
of the configuration of the bootstrap circuit 401 will be
30 described later. The driving circuit 312DC that is a
second driving circuit converts pulse_312A from the control
unit 10 into a second driving signal for driving the
switching element 312 by using a voltage output from the
44
direct-current voltage source 300 as the power supply
voltage, and outputs the second driving signal to the gate
of the switching element 312.
[0090] The driving circuit 321DC converts pulse_321A
from the control unit 10 into a driving 5 signal for driving
the switching element 321 by using a voltage from the
bootstrap circuit 402 as the power supply voltage, and
outputs the driving signal to the gate of the switching
element 321. The driving circuit 322DC converts pulse_322A
10 from the control unit 10 into a driving signal for driving
the switching element 322 by using a voltage output from
the direct-current voltage source 300 as the power supply
voltage, and outputs the driving signal to the gate of the
switching element 322.
15 [0091] The bootstrap circuit 401 includes a boot
resistor 311R having one end connected to the directcurrent
voltage source 300, a boot diode 311D having an
anode connected to the other end of the boot resistor 311R,
a boot capacitor 311C that is a second capacitor having one
20 end connected to a cathode of the boot diode 311D and the
other end connected to the driving circuit 311DC, and a
gate voltage suppression diode 311D'.
[0092] An anode of the gate voltage suppression diode
311D' is connected to the cathode of the boot diode 311D
25 and one end of the boot capacitor 311C. A cathode of the
gate voltage suppression diode 311D' is connected to the
driving circuit 311DC. Assume that the value of a first
voltage that is a voltage at which a forward current starts
to flow in the gate voltage suppression diode 311D' is
30 lower than the value of a second voltage that is a voltage
at which a forward current starts to flow in the body diode
312a. Thus, assume that the forward current - forward
voltage characteristics of the gate voltage suppression
45
diode 311D' are superior to the forward current - forward
voltage characteristics of the body diode 312a. Note that,
a voltage at which a forward current starts to flow in a
diode is typically called a forward voltage.
[0093] The bootstrap circuit 402 has 5 a configuration
similar to that of the bootstrap circuit 401, and includes
a boot resistor 321R having one end connected to the
direct-current voltage source 300, a boot diode 321D having
an anode connected to the other end of the boot resistor
10 321R, a boot capacitor 321C having one end connected to a
cathode of the boot diode 321D and the other end connected
to the driving circuit 321DC, and a gate voltage
suppression diode 321D'.
[0094] An anode of the gate voltage suppression diode
15 321D' is connected to the cathode of the boot diode 321D
and one end of the boot capacitor 321C. A cathode of the
gate voltage suppression diode 321D' is connected to the
driving circuit 321DC. Assume that the value of a voltage
at which a forward current starts to flow in the gate
20 voltage suppression diode 321D' is lower than the value of
a voltage at which a forward current starts to flow in the
body diode 322a. Thus, assume that the forward current -
forward voltage characteristics of the gate voltage
suppression diode 321D' are superior to the forward current
25 - forward voltage characteristics of the body diode 322a.
The reason why the gate voltage suppression diode 311D' is
used will be described later. Note that, because the
bootstrap circuit 402 has a configuration similar to that
of the bootstrap circuit 401, details of the configuration
30 of the bootstrap circuit 402 will not be described.
[0095] In the bootstrap circuit 401 having such a
configuration, when the switching element 312 is turned ON,
a current flows through a path constituted by the direct46
current voltage source 300, the boot resistor 311R, the
boot diode 311D, the boot capacitor 311C, and the switching
element 312, and the boot capacitor 311C is charged. A
capacitor voltage Vc generated across the ends of the
charged boot capacitor 311C can be expressed 5 as Vc=Vdc+VBDVdr-
Vf. Vdc represents the voltage of the direct-current
voltage sources 300, VBD represents the forward voltage of
the body diode 312a, Vdr represents a drop voltage of the
boot resistor 311R, and Vf represents the forward voltage
10 of the boot diode 311D.
[0096] For example, when Vdc is 6.0 V, VBD is 3.0 V, Vdr
is 0.5 V, and Vf is 1.5 V, Vc is 7.0 V. In this case, when
the rated voltage of the driving circuit 311DC is 6.0 V,
the value of Vc is higher than the rated voltage of the
15 driving circuit 311DC. The reason why the value of Vc is
high is that the forward voltage of the body diode 312a is
also applied to the boot capacitor 311C in addition to the
voltage of the direct-current voltage source 300. The
forward voltage of the body diode 312a is a voltage at
20 which a forward current starts to flow in the body diode
312a. For example, in a case where a switching element
made of a WBG semiconductor in which the potential barrier
of a p-n junction is high is used as the switching element
312, the forward voltage of the body diode 312a of the
25 switching element 312 tends to be high. Note that the
switching element 312 in which the forward voltage of the
body diode 312a becomes high is not limited to a switching
element made of a WBG semiconductor, and a Si switching
element in which the forward voltage of a body diode tends
30 to be high so that the capacitor voltage Vc of the boot
capacitor 311C is higher than the rated voltage of the
driving circuit 311DC may also be applicable.
[0097] When the capacitor voltage Vc becomes higher than
47
the rated voltage of the driving circuit 311DC, the
withstand voltage of the driving circuit 311DC may decrease.
In addition, because the value of the driving signal
generated by the driving circuit 311DC becomes larger, the
short circuit withstand of the switching 5 element 311 may
decrease. In addition, when the switching element 311 is
driven by the driving circuit 311DC to which such a high
voltage is applied, the value of the driving signal
generated by the driving circuit 311DC becomes larger than
10 the value of the driving signal generated by the driving
circuit 312DC to which the voltage of the direct-current
voltage source 300 is applied. Thus, the value of the loss
of the switching element 311 in the ON state and the value
of the loss of the switching element 312 in the ON state
15 are different from each other, and imbalance in heat
generation between the switching element 311 and the
switching element 312 increases. When the imbalance in
heat generation increases and a junction temperature of a
semiconductor constituting one of the switching elements
20 exceeds a permissible value, there is a possibility that
normal operations can no longer be performed.
[0098] In the power converting apparatus 100 illustrated
in FIG. 26, the gate voltage suppression diode 311D' is
provided between the boot capacitor 311C and the driving
25 circuit 311DC. In other words, the boot capacitor 311C is
connected with the driving circuit 311DC via the gate
voltage suppression diode 311D'. Thus, the capacitor
voltage of the boot capacitor 311C is reduced by a certain
value by the gate voltage suppression diode 311D', and then
30 applied as the power supply voltage for the driving circuit
311DC to the driving circuit 311DC. In this manner, the
gate voltage suppression diode 311D' functions as a power
supply voltage adjusting element for adjusting the power
48
supply voltage for the driving circuit 311DC to be applied
from the boot capacitor 311C to the driving circuit 311DC.
[0099] For example, when Vdc is 6.0 V, VBD is 3.0 V, Vdr
is 0.5 V, Vf is 1.5 V, and VD is 1.0 V, the value of Vc is
expressed as Vc=Vdc+VBD-Vdr-Vf-VD and Vc=6.0 5 V is obtained. VD
represents a forward voltage of the gate voltage
suppression diode 311D', that is, a voltage at which a
forward current starts to flow in the gate voltage
suppression diode 311D'.
10 [0100] As described above, while a driving voltage equal
to Vc (7.0 V) is applied to the driving circuit 311DC when
the gate voltage suppression diode 311D' is not provided,
6.0 V is applied to the driving circuit 311DC when the gate
voltage suppression diode 311D' is provided. Thus, as a
15 result of providing the gate voltage suppression diode
311D', the power supply voltage for the driving circuit
311DC to be applied from the boot capacitor 311C to the
driving circuit 311DC can be reduced to the rated voltage
of the driving circuit 311DC. In addition, when Vdc, VBD,
20 Vdr, Vf, VD, etc. are set as described above, the power
supply voltage for the driving circuit 311DC becomes equal
to the voltage Vdc of the direct-current voltage source 300.
[0101] According to the power converting apparatus 100
according to the first embodiment, a decrease in the
25 withstand voltage of the driving circuit 311DC can be
prevented or reduced, and a decrease in the short circuit
withstand of the switching element 311 can be prevented or
reduced. In addition, because the power supply voltage for
the driving circuit 311DC can be adjusted to a value equal
30 to the power supply voltage for the driving circuit 312DC,
the imbalance in heat generation between the switching
element 311 and the switching element 312 can be reduced,
which improves the reliability of the power converting
49
apparatus 100.
[0102] In addition, according to the power converting
apparatus 100 according to the first embodiment, because
the power supply voltage for the driving circuit 311DC can
be adjusted to a value equal to the power 5 supply voltage
for the driving circuit 312DC, the driving circuit 311DC
and the driving circuit 312DC can be constituted by common
components. This improves the yield of components as
compared with a case where the driving circuit 311DC and
10 the driving circuit 312DC are produced from different
components. In addition, the manufacturing cost of the
driving circuit 311DC and the driving circuit 312DC is
reduced, and the volume of the components during
manufacture of the driving circuit 311DC and the driving
15 circuit 312DC can be reduced. Furthermore, replacement of
the driving circuit 311DC and the driving circuit 312DC in
repairing the power converting apparatus 100 is facilitated.
[0103] Note that, while the gate voltage suppression
diode 311D' is provided inside the bootstrap circuit 401 in
20 the power converting apparatus 100 illustrated in FIG. 26,
the gate voltage suppression diode 311D' may be produced
separately from the bootstrap circuit 401 and provided
between the bootstrap circuit 401 and the driving circuit
311DC. In the case where the gate voltage suppression
25 diode 311D' is provided inside the bootstrap circuit 401,
the bootstrap circuit 401 can be manufactured such that the
gate voltage suppression diode 311D', the boot capacitor
311C, and the like are formed integrally. This improves
the production efficiency of the power converting apparatus
30 100. In the case where the gate voltage suppression diode
311D' is produced separately from the bootstrap circuit 401
and provided between the bootstrap circuit 401 and the
driving circuit 311DC, a suitable gate voltage suppression
50
diode 311D' according to the value of the forward voltage
of the body diode 312a can be selected from a plurality of
gate voltage suppression diodes 311D' with different
forward voltages and can be mounted. This enables the
power supply voltage for the driving circuit 5 311DC to be
easily adjusted.
[0104] FIG. 27 is a diagram illustrating an example of a
configuration of a power converting apparatus according to
a first modification of the first embodiment. In a power
10 converting apparatus 100-1 illustrated in FIG. 27,
bootstrap circuits 401A and 402A are used instead of the
bootstrap circuits 401 and 402 illustrated in FIG. 26. In
the bootstrap circuit 401A, the gate voltage suppression
diode 311D' is not provided, and one end of the boot
15 capacitor 311C is directly connected to the driving circuit
311DC. In the bootstrap circuit 402A, the gate voltage
suppression diode 321D' is not provided, and one end of the
boot capacitor 321C is directly connected to the driving
circuit 321DC. In addition, in the power converting
20 apparatus 100-1, a gate voltage suppression diode 312RD is
connected in parallel with the switching element 312, and a
gate voltage suppression diode 322RD is connected in
parallel with the switching element 322.
[0105] An anode of the gate voltage suppression diode
25 312RD is connected to an anode of the body diode 312a, and
a cathode of the gate voltage suppression diode 312RD is
connected to a cathode of the body diode 312a. Assume that
the forward current - forward voltage characteristics of
the gate voltage suppression diode 312RD are superior to
30 the forward current - forward voltage characteristics of
the body diode 312a. For example, when the forward voltage
of the gate voltage suppression diode 312RD is 1.5 V and
the forward voltage of the body diode 312a is 3.0 V, the
51
boot capacitor 311C is charged by a voltage having a value
obtained by subtracting the drop voltage of the boot
resistor 311R and the forward voltage of the boot diode
311D from a sum of 1.5 V and the voltage of the directcurrent
voltage source 300. The capacitor 5 voltage of the
charged boot capacitor 311C has a value smaller than that
in the case where no gate voltage suppression diode 312RD
is used, and is used as the power supply voltage for the
driving circuit 311DC. As described above, the gate
10 voltage suppression diode 312RD functions as a capacitor
voltage adjusting element for adjusting a capacitor voltage
generated across the ends of the boot capacitor 311C.
[0106] An anode of the gate voltage suppression diode
322RD is connected to an anode of the body diode 322a, and
15 a cathode of the gate voltage suppression diode 322RD is
connected to a cathode of the body diode 322a. Assume that
the forward current - forward voltage characteristics of
the gate voltage suppression diode 322RD are superior to
the forward current - forward voltage characteristics of
20 the body diode 322a. The gate voltage suppression diode
322RD functions as a capacitor voltage adjusting element
for adjusting the capacitor voltage generated across the
ends of the boot capacitor 321C.
[0107] According to the power converting apparatus 100-1
25 illustrated in FIG. 27, an increase in the charging voltage
of the boot capacitor can be prevented or reduced, and an
increase in the loss due to the body diode during an
asynchronous rectification period of the zero crossing and
the dead time can be prevented or reduced.
30 [0108] FIG. 28 is a diagram illustrating an example of a
configuration of a power converting apparatus according to
a second modification of the first embodiment. In a power
converting apparatus 100-2 illustrated in FIG. 28, the
52
bootstrap circuit 402A illustrated in FIG. 27 is used
instead of the bootstrap circuit 402 illustrated in FIG. 26.
Thus, in the power converting apparatus 100-2, the gate
voltage suppression diode 311D' is used only in the first
5 arm.
[0109] In a power converting apparatus having a fullbridge
configuration like the power converting apparatus
100-2, no path for charging the boot capacitor via the body
diode occurs by synchronous rectification control based on
10 the power supply polarities of the switching elements 321
and 322. Thus, in the power converting apparatus 100-2,
the gate voltage suppression diode 311D' may be implemented
only in the first arm, which enables reduction in used
components.
15 [0110] FIG. 29 is a diagram illustrating an example of a
configuration of a power converting apparatus according to
a third modification of the first embodiment. In a power
converting apparatus 100-3 illustrated in FIG. 29, the gate
voltage suppression diode 322RD illustrated in FIG. 27 is
20 not provided. Thus, in the power converting apparatus 100-
3, the gate voltage suppression diode 312RD is used only in
the first arm. In the power converting apparatus 100-3, in
a manner similar to the power converting apparatus 100-2,
no path for charging the boot capacitor via the body diode
25 occurs by synchronous rectification control based on the
power supply polarities of the switching elements 321 and
322. Thus, in the power converting apparatus 100-3, the
gate voltage suppression diode 312RD may be implemented
only in the first arm, which enables reduction in used
30 components.
[0111] Note that, in the first embodiment, in a case
where the alternating-current power supply 1 is a
commercial power supply of 50 Hz or 60 Hz, the audible
53
frequency is in a range from 16 kHz to 20 kHz, that is, a
range from 266 to 400 times the frequency of the commercial
power supply. When the switching elements are driven with
such audible frequency, there is a problem of noise caused
by switching. Because switching elements 5 made of WBG
semiconductors can perform fast switching, switching
elements made of WBG semiconductors are suitable for
switching elements that can be switched at a frequency
higher than such audible frequency, such as a switching
10 frequency higher than 20 kHz.
[0112] In addition, in a case where switching elements
made of Si semiconductors are driven at a switching
frequency of several tens of kHz or higher, such as a
switching frequency higher than 20 kHz, the ratio of the
15 switching loss increases, and a measure for heat radiation
is essential. In the case of switching elements made of
WBG semiconductors, the switching loss is much smaller than
that in the case of switching elements made of Si
semiconductors even when the switching elements are driven
20 at a switching frequency higher than 20 kHz. Thus, use of
switching elements made of WBG semiconductors in the power
converting apparatus 100 eliminates the need for a measure
for heat radiation of switching elements or allows
miniaturization of members, such as radiating fins, used
25 for a measure for heat radiation of switching elements,
which enables reduction in size and weight of the power
converting apparatus 100. In addition, high-frequency
switching of switching elements made of WBG semiconductors
can be performed, which can make the inductance of the
30 reactor 2 relatively smaller. Thus, the reactor 2 can be
reduced in size. Note that the switching frequency is
preferably equal to or lower than 150 kHz so that the
primary component of the switching frequency is not
54
included in a range of measurement of noise terminal
voltage standard.
[0113] In addition, WBG semiconductors have a smaller
capacitance than Si semiconductors; therefore, a recovery
current caused by switching is low and the 5 occurrence of a
loss and noise caused by a recovery current can thus be
reduced. Thus, WBG semiconductors are suitable for highfrequency
switching.
[0114] In addition, even in a case where WBG
10 semiconductors are driven at a high frequency about 100 kHz,
an increase in a loss generated in the switching elements
is prevented or reduced; therefore, the loss reduction
effect produced by miniaturization of the reactor 2
increases. Thus, a highly efficient converter can be
15 achieved in a wide output band, that is, under a wide load
condition.
[0115] In addition, WBG semiconductors have a higher
heat resistance than Si semiconductors, and have a higher
permissible level of heat generation by switching due to
20 imbalance in the loss between arms. Because the first arm
31 is driven at a higher frequency than the second arm 32
and the switching loss and the heat generation of the first
arm 31 thus increase, WBG semiconductors are more suitable
for the first arm 31 with high heat generation than the
25 second arm 32.
[0116] Note that super junction (SJ)-MOSFETs may be used
for switching elements constituting an arm that performs
slow switching. Use of SJ-MOSFETs for an arm that performs
slow switching can reduce the disadvantages of SJ-MOSFETs,
30 which are high capacitance and high occurrence of recovery,
while making use of low ON-resistance that is an advantage
of SJ-MOSFETs. In addition, use of SJ-MOSFETs can reduce
the manufacturing cost of the arm that performs slow
55
switching as compared to use of switching elements made of
WBG semiconductors.
[0117] Note that the power converting apparatus 100
according to the first embodiment may be constituted by a
general-purpose intelligent power module 5 (IPM). Use of an
IPM enables the driving circuits for the switching elements
311, 312, 321, and 322 to be contained inside the IPM,
which can reduce the board area on which the reactor 2, the
bridge circuit 3, the smoothing capacitor 4, the power
10 supply voltage detecting unit 5, the power supply current
detecting unit 6, the bus voltage detecting unit 7, and the
control unit 10 are mounted. In addition, use of a
general-purpose IPM can prevent or reduce an increase in
cost.
15 [0118] Note that the power converting apparatus 100
according to the first embodiment only needs to obtain the
polarity of the power supply voltage Vs, and is not limited
to the configuration for determining the polarity of the
power supply voltage Vs by detecting a zero crossing point
20 of the power supply voltage Vs. In the case of detecting a
zero crossing point, in order to prevent erroneous
determination of the polarity near the zero crossing, the
power converting apparatus 100 turns the operations of the
first arm 31 and the second arm 32 OFF for a predetermined
25 period from the zero crossing point on the basis of the
power supply voltage phase estimation value θ^
s.
[0119] While the switching element 321 and the switching
element 322 are permitted to be in the ON state when the
absolute value of the power supply current Is is equal to
30 or larger than the current threshold in the power
converting apparatus 100 according to the first embodiment,
the configuration of the power converting apparatus 100 is
not limited thereto. The power converting apparatus 100
56
may estimate that a current flows in a body diode of a
switching element by using any of the power supply voltage
Vs, a voltage applied to the first arm 31, the bus voltage
Vdc, and a voltage applied across the ends of the switching
element to control the switching element 5 321 and the
switching element 322. In the case of estimating that a
current flows in a body diode of a switching element by
using any of the power supply voltage Vs, the voltage
applied to the first arm 31, and the bus voltage Vdc, there
10 are many factors of variation in determination and thus
attention should be given to estimation error. In addition,
in the case of estimating that a current flows in a body
diode of a switching element by using the voltage applied
across the switching element, a voltage detecting circuit
15 is required for each of the switching elements for which a
current flow is to be estimated.
[0120] While the example in which the synchronous
rectification control is performed by detecting the power
supply current Is is described in the first embodiment, the
20 power converting apparatus 100 according to the first
embodiment may have a configuration to perform synchronous
rectification control by detecting a current flowing
through a bus between the bridge circuit 3 and the
smoothing capacitor 4 instead of the power supply current
25 Is. In this case, because the current in a short-circuit
path cannot be detected, the synchronous rectification
control using a current threshold may shorten the period
during which the synchronous rectification operation can be
performed. Thus, in the case of performing the synchronous
30 rectification control by detecting a bus current, control
may be performed such that the switching element 321 or the
switching element 322 is turned ON depending on the
polarity even when the absolute value of the power supply
57
current Is is smaller than the threshold during the
operation with the short-circuit current as described above.
In this case, the synchronous rectification operation can
be performed for a long period; therefore, the conduction
loss of the switching element 321 or the 5 switching element
322 can be reduced.
[0121] Note that it is desirable that the first arm 31
be configured as a so-called 2-in-1 module in which the
switching elements 311 and 312 are provided in one package.
10 Similarly, it is desirable that the second arm 32 be
configured as a 2-in-1 module in which the switching
elements 321 and 322 are provided in one package. In a 2-
in-1 module, two switching elements having the same
switching characteristics are often mounted. When each of
15 the first arm 31 and the second arm 32 is configured as a
2-in-1 module, the imbalance in heat generation between the
switching element 311 and the switching element 312 is
reduced and further, the imbalance in heat generation
between the switching element 321 and the switching element
20 322 is reduced, as compared with the case where the
switching elements 311, 312, 321, and 322 are each
configured as one module.
[0122] As described above, according to the first
embodiment, because an increase in the power supply voltage
25 for the driving circuit 311DC can be prevented or reduced,
a decrease in the withstand voltage of the driving circuit
can be prevented or reduced, a decrease in the short
circuit withstand of the switching element can be prevented
or reduced, and further, the imbalance in heat generation
30 between the switching element 311 and the switching element
312 can be reduced. As a result, the reliability of the
power converting apparatus 100 can be improved. In
addition, because an increase in the power supply voltage
58
for the driving circuit 311DC can be prevented or reduced,
an isolated power supply for improving a dielectric
strength need not be additionally provided between the
bootstrap circuit 401 and the driving circuit 311DC, the
structure of the power converting 5 apparatus 100 is
simplified, and the manufacturing cost of the power
converting apparatus 100 can be reduced. In addition,
because the voltage of the direct-current voltage source
300 need not be reduced in order to prevent or reduce an
10 increase in the capacitor voltage, the power supply voltage
for the driving circuit 311DC can be adjusted to a value
equal to the power supply voltage for the driving circuit
312DC while the power supply voltage with which the driving
circuit 312DC can operate is ensured. Consequently, the
15 imbalance in heat generation between the switching element
311 and the switching element 312 can be reduced, and the
reliability of the power converting apparatus 100 is
improved. In addition, because the power supply voltage
for the driving circuit 311DC can be adjusted to a value
20 equal to the power supply voltage for the driving circuit
312DC, the loss caused when one of the power supply
voltages becomes higher than necessary during switching
operations is reduced, the power consumption of the power
converting apparatus 100 is reduced, and the efficiency of
25 the power converting apparatus 100 can be improved. In
addition, because an increase in the power supply voltage
for the driving circuit 311DC can be prevented or reduced
even when switching elements, such as WBG MOSFETs, with
inferior forward current - forward voltage characteristics
30 of the body diodes are used, the first embodiment is
suitable for the power converting apparatus 100 including
WBG MOSFETs, in particular, SiC MOSFETs. In addition, the
first embodiment is suitable for the power converting
59
apparatus 100 including switching elements, such as WBG
switching elements, having characteristics sensitive to a
gate driving voltage.
[0123] The sensitivity to the gate driving voltage will
be explained. A conduction loss and a 5 switching loss are
used for performance indices of a SiC MOSFET. The
conduction loss is determined by the ON-resistance and the
current value of a MOSFET, and the ON-resistance is known
to vary significantly depending on the gate driving voltage.
10 Typically, the ON-resistance exhibits a tendency to rapidly
increase when the gate driving voltage is low, and
converges to a specific value as the gate driving voltage
becomes higher. A semiconductor, however, has an element
withstand voltage; therefore, the gate driving voltage
15 cannot be increased unlimitedly. In such a case where the
ON-resistance converges to a specific value when the gate
driving voltage is 16 to 18 V, for example, the ONresistance
becomes twice the specific value when the gate
driving voltage is reduced to 10 V. The change in the ON20
resistance depending on the value of the gate driving
voltage in this manner is referred to as sensitivity to the
gate driving voltage in the present embodiment.
[0124] Second Embodiment.
While a switching element pair constituted by two
25 switching elements connected in series is provided in the
first arm 31 in the first embodiment, a configuration in
which n pairs of switching elements are connected in
parallel in the first arm 31 and synchronous control is
performed thereon will be described in a second embodiment.
30 Here, n is an integer not smaller than 2. FIG. 30 is a
diagram illustrating an example of a configuration of a
power converting apparatus according to the second
embodiment. In a power converting apparatus 100A according
60
to the second embodiment, the first arm 31 includes a
switching element 313 that is a fifth switching element and
a switching element 314 that is a sixth switching element.
The switching element 313 and the switching element 314 are
connected in series. A switching element 5 pair constituted
by the switching element 313 and the switching element 314
is connected in parallel with the switching element pair
constituted by the switching element 311 and the switching
element 312. A reactor 2 is connected at a connection
10 point of the switching element 313 and the switching
element 314. FIG. 30 illustrates an example of a
configuration in which synchronous control is performed by
using two arms.
[0125] When driving the first arm 31 in which two pairs
15 of switching elements are connected in parallel, the
control unit 10 drives two switching elements 311 and 313
constituting an upper arm simultaneously and drives two
switching elements 312 and 314 constituting a lower arm
simultaneously, among the two pairs of switching elements.
20 Note that simultaneously driving two switching elements
connected in parallel with each other will be referred to
as “parallel driving”.
[0126] The parallel driving of two pairs of switching
elements connected in parallel reduces the current flowing
25 in each of the switching elements to half of that in the
case of one pair of switching elements. As is clear from
the characteristics in FIG. 19, as the current is smaller,
the loss of the switching element is smaller, and the loss
occurring in the first arm 31 is thus reduced.
30 Consequently, the imbalance in heat generation between the
first arm 31 and the second arm 32 can further be reduced.
[0127] While an example of the configuration in which
two pairs of switching elements are connected in parallel
61
is illustrated in FIG. 30, the number of pairs of switching
elements is not limited to two, and may be n. In a case
where the first arm 31 is constituted by n pairs of
switching elements, the current flowing in one pair of
switching elements is reduced to one n-5 th, and the loss in
the first arm 31 can thus be further reduced. Note that
the imbalance in loss among n pairs of switching elements
connected in parallel need not be completely eliminated,
and the number of pairs of switching elements to be
10 connected in parallel may be selected within a range in
which the imbalance in loss is permitted.
[0128] In addition, simultaneous driving of two
switching elements connected in parallel in the first arm
31 is explained in the example of FIG. 30. Thus, in the
15 second embodiment, a synchronous control method of
simultaneously switching the switching elements connected
in parallel is employed. The method of controlling the
switching elements connected in parallel, however, is not
limited thereto, and so-called interleaved control in which
20 the phases of two switching elements connected in parallel
are shifted by 180° from each other for control may be used.
[0129] In the interleaved control, the phases when the
switching element 311 and the switching element 313
connected in parallel are turned ON are shifted by 180°
25 from each other for control, and the phases when the
switching element 312 and the switching element 314
connected in parallel are turned on are shifted by 180°
from each other for control. As a result, the two
switching elements connected in parallel are subjected to
30 interleaved driving.
[0130] Interleaved driving of the first arm 31
facilitates driving at higher frequency, and enables
reduction in size of the reactor 2 and reduction in the
62
reactor loss. Note that, in a case of being frequently
used in the passive state like the case with conditioners,
the reactor 2 need not be reduced in size, and the
configurations and operations of the first embodiment are
more effective in terms of harmonic wave 5 prevention and the
power-supply power factor.
[0131] While one reactor 2 is provided between the
alternating-current power supply 1 and the first arm 31 in
the first and second embodiments, the configurations of the
10 first and second embodiments are not limited thereto, and a
reactor may also be provided between the alternatingcurrent
power supply 1 and the second arm 32. Use of two
reactors in this manner can make the capacity of each
reactor smaller, which improves the design freedom of the
15 power converting apparatuses 100 and 100A as compared with
a case where one reactor with a large capacity is used.
[0132] A hardware configuration of the control unit 10
of the power converting apparatuses 100 and 100A according
to the first and second embodiments will now be described.
20 FIG. 31 is a diagram illustrating an example of the
hardware configuration for implementing the control unit of
the first and second embodiments. The control unit 10
described in the first and second embodiments is
implemented by a processor 201 and a memory 202.
25 [0133] The processor 201 is a central processing unit
(CPU; also referred to as a central processing device, a
processing device, a computing device, a microprocessor, a
microcomputer, a processor, or a digital signal processor
(DSP)), or a system large scale integration (LSI). The
30 memory 202 is a semiconductor memory such as a random
access memory (RAM), a read only memory (ROM), a flash
memory, an erasable programmable read only memory (EPROM),
or an electrically erasable programmable read only memory
63
(EEPROM: registered trademark). The semiconductor memory
may be a nonvolatile memory or may be a volatile memory.
Alternatively, the memory 202 may be a magnetic disk, a
flexible disk, an optical disk, a compact disk, a mini disc,
or a digital versatile disc 5 (DVD) instead of a
semiconductor memory.
[0134] The power supply current command value control
unit 21, the ON-duty control unit 22, the power supply
voltage phase calculating unit 23, the first pulse
10 generating unit 24, the second pulse generating unit 25,
the current command value calculating unit 26, and the
instantaneous value command value calculating unit 27
illustrated in FIG. 13 are implemented by the processor 201
and the memory 202 illustrated in FIG. 31. Specifically,
15 the respective units are implemented by the processor 201
by storing programs for operating as each of the power
supply current command value control unit 21, the ON-duty
control unit 22, the power supply voltage phase calculating
unit 23, the first pulse generating unit 24, the second
20 pulse generating unit 25, the current command value
calculating unit 26, and the instantaneous value command
value calculating unit 27 in the memory 202 and reading and
executing the programs stored in the memory 202.
[0135] Third Embodiment.
25 FIG. 32 is a diagram illustrating an example of a
configuration of a motor driving apparatus according to a
third embodiment. A motor driving apparatus 101 according
to a third embodiment drives a motor 42 that is a load.
The motor driving apparatus 101 includes the power
30 converting apparatus 100 of the first embodiment, an
inverter 41, a motor current detecting unit 44, and an
inverter controlling unit 43. The inverter 41 drives the
motor 42 by converting a direct-current power supplied from
64
the power converting apparatus 100 into an alternatingcurrent
power and outputting the alternating-current power
to the motor 42.
[0136] Note that the motor driving apparatus 101 may
include the power converting apparatus 5 100A of the second
embodiment instead of the power converting apparatus 100 of
the first embodiment. In addition, while the load of the
motor driving apparatus 101, that is, the device connected
to the inverter 41 is the motor 42 in the third embodiment,
10 the device connected to the inverter 41 may be any device,
other than the motor 42, to which an alternating-current
power is input.
[0137] The inverter 41 is a circuit including switching
elements, including insulated gate bipolar transistors
15 (IGBTs), in a three-phase bridge configuration or a twophase
bridge configuration. The switching elements
included in the inverter 41 are not limited to IGBTs, but
may be switching elements made of WBG semiconductors,
insulated gate controlled thyristors (IGCTs), field effect
20 transistors (FETs) or MOSFETs.
[0138] The motor current detecting unit 44 detects a
current flowing between the inverter 41 and the motor 42.
The inverter controlling unit 43 generates PWM signals for
driving the switching elements in the inverter 41 by using
25 a current detected by the motor current detecting unit 44
such that the motor 42 rotates at a rotating speed, and
outputs the generated PWM signals to the inverter 41. The
inverter controlling unit 43 is implemented by a processor
and a memory in a manner similar to the control unit 10.
30 Note that the inverter controlling unit 43 of the motor
driving apparatus 101 and the control unit 10 of the power
converting apparatus 100 may be implemented by one circuit.
[0139] In a case where the power converting apparatus
65
100 or 100A according to the first or second embodiment is
used in the motor driving apparatus 101, the bus voltage
Vdc necessary for controlling the bridge circuit 3
illustrated in FIG. 1 and FIG. 30 changes depending on the
operation state of the motor 42. 5 Typically, as the
rotating speed of the motor 42 is higher, the voltage
output from the inverter 41 need to be higher. The upper
limit of the voltage output from the inverter 41 is limited
by a voltage input to the inverter 41, that is, the bus
10 voltage Vdc that is output from the power converting
apparatus 100 or 100A. A region in which the voltage
output from the inverter 41 exceeds the upper limit limited
by the bus voltage Vdc and saturated is called an
overmodulation region.
15 [0140] In the motor driving apparatus 101 as described
above, the bus voltage Vdc need not be increased in a low
rotation range of the motor 42, that is, in a range in
which the overmodulation region is not reached. In
contrast, when the motor 42 rotates at high speed, the
20 overmodulation region can be shifted toward higher rotation
by increasing the bus voltage Vdc. As a result, the
operation range of the motor 42 can be expanded toward
higher rotation.
[0141] In addition, when the operation range of the
25 motor 42 need not be expanded, the number of coil turns
around a stator of the motor 42 can be increased by a
corresponding amount. In the low rotation region, the
increase in the number of coil turns makes the motor
voltage generated across the coil ends higher and lowers
30 the current flowing in the coil accordingly, which reduces
the loss caused by the switching operation of the switching
elements in the inverter 41. For producing both effects of
expansion of the operation range of the motor 42 and
66
improvement in the loss in the low rotation region, the
number of coil turns of the motor 42 is set to an
appropriate value.
[0142] According to the third embodiment, because the
power converting apparatus 100 or 100A 5 according to the
first or second embodiment is used, the effect of improving
the reliability of the motor driving apparatus 101 is
produced. In addition, because an increase in the
temperature of the motor driving apparatus 101 is prevented
10 or reduced as a result of applying switching elements made
of WBG semiconductors to the power converting apparatus 100
or 100A according to the first or second embodiment, the
capacity of cooling the components mounted on the motor
driving apparatus 101 can be ensured even when the motor
15 driving apparatus 101 is reduced in size. In addition,
high-frequency driving of switching elements made of WBG
semiconductors enables reduction in size and loss of the
reactor 2. Thus, as a result of applying switching
elements made of WBG semiconductors to the power converting
20 apparatus 100 or 100A according to the first or second
embodiment, an increase in weight of the motor driving
apparatus 101 can be prevented or reduced.
[0143] Fourth Embodiment.
FIG. 33 is a diagram illustrating an example of a
25 configuration of an air conditioner according to a fourth
embodiment. An air conditioner 700 according to the fourth
embodiment is an example of a refrigeration cycle system,
and includes the motor driving apparatus 101 according to
the third embodiment, and a motor 42. The air conditioner
30 700 also includes a compressor 81, a four-way valve 82, an
outdoor heat exchanger 83, an expansion valve 84, an indoor
heat exchanger 85, and refrigerant piping 86.
[0144] The air conditioner 700 may be a split air
67
conditioner in which an outdoor unit is separated from an
indoor unit, or may be an integrated air conditioner in
which the compressor 81, the indoor heat exchanger 85, and
the outdoor heat exchanger 83 are arranged in one housing.
[0145] The compressor 81 includes therein 5 a compression
mechanism 87 for compressing the refrigerant, and a motor
42 for causing the compression mechanism 87 to operate.
The motor 42 is driven by the motor driving apparatus 101.
In the air conditioner 700, a refrigeration cycle is
10 constituted by circulation of refrigerant through the
compressor 81, the four-way valve 82, the outdoor heat
exchanger 83, the expansion valve 84, the indoor heat
exchanger 85, and the refrigerant piping 86.
[0146] Note that the components of the air conditioner
15 700 can also be applied to such equipment as a refrigerator
or a freezer including a refrigeration cycle. In addition,
while the motor 42 is used for a driving source of the
compressor 81 in the fourth embodiment, the motor 42 may be
used as a driving source for driving each of an indoor unit
20 fan and an outdoor unit fan, which are not illustrated,
instead of the compressor 81. Alternatively, the motor 42
may be applied to a driving source for each of the indoor
unit fan, the outdoor unit fan, and the compressor 81, and
the three motors 42 may be driven by the motor driving
25 apparatus 101.
[0147] In addition, because the operation of the air
conditioner 700 under an intermediate condition in which
the power output is equal to or lower than half of a rated
power output, that is the operation of the air conditioner
30 700 in a low power output range is dominant throughout the
year, the contribution to the annual power consumption
under the intermediate condition is high. In addition, in
the air conditioner 700, the rotating speed of the motor 42
68
tends to be low, and the bus voltage required for driving
the motor 42 tends to be low. Thus, operation of the
switching elements used in the air conditioner 700 in a
passive state is effective in terms of system efficiency.
The power converting apparatus 100 capable 5 of reducing the
loss in a wide range of operation modes from the passive
state to the high-frequency switching state is therefore
useful for the air conditioner 700. Although the reactor 2
can be reduced in size with the interleaved control as
10 described above, the frequency of operation of the air
conditioner 700 under the intermediate condition is high
and thus the reactor 2 need not be reduced in size;
therefore, the configurations and operations of the power
converting apparatus 100 or 100A according to the first or
15 second embodiment are more effective in terms of harmonic
wave prevention and the power-supply power factor.
[0148] In addition, as described above, because the
switching loss in the case where switching elements made of
WBG semiconductors are driven at a high switching frequency
20 equal to or higher than 10 kHz is smaller than that of
switching elements made of Si semiconductors, application
of switching elements made of WBG semiconductors to the
power converting apparatus 100 or 100A according to the
first or second embodiment prevents or reduces an increase
25 in temperature of the motor driving apparatus 101. As a
result, the capacity of cooling the components mounted on
the motor driving apparatus 101 can be ensured even when
the outdoor unit fan is reduced in size. The power
converting apparatus 100 or 100A according to the first or
30 second embodiment is therefore suitable for use in the air
conditioner 700 that is highly efficient and has a high
power equal to or higher than 4.0 kW.
[0149] In addition, switching elements made of WBG
69
semiconductors can be driven at higher frequency than
switching elements made of Si semiconductors. Thus, highfrequency
driving enables reduction in size and in loss of
the reactor 2. Thus, as a result of applying switching
elements made of WBG semiconductors to the 5 power converting
apparatus 100 or 100A according to the first or second
embodiment, an increase in weight of the air conditioner
700 can be prevented or reduced.
[0150] In addition, according to the fourth embodiment,
10 high-frequency driving of the switching elements reduces
the switching loss, and the air conditioner 700 with a low
energy consumption rate and high efficiency can thus be
achieved.
[0151] The configurations presented in the embodiments
15 above are examples of the present invention, and can be
combined with other known technologies or can be partly
omitted or modified without departing from the scope of the
present invention.
20 Reference Signs List
[0152] 1 single-phase alternating-current power supply;
2 reactor; 3 bridge circuit; 4 smoothing capacitor; 5
power supply voltage detecting unit; 6 power supply
current detecting unit; 7 bus voltage detecting unit; 10
25 control unit; 11, 311, 312, 313, 314, 321, 322 switching
element; 21 power supply current command value control
unit; 22 on-duty control unit; 23 power supply voltage
phase calculating unit; 24 first pulse generating unit; 25
second pulse generating unit; 26 current command value
30 calculating unit; 27 instantaneous value command value
calculating unit; 31 first arm; 32 second arm; 41
inverter; 42 motor; 43 inverter controlling unit; 44
motor current detecting unit; 50 load; 81 compressor; 82
70
four-way valve; 83 outdoor heat exchanger; 84 expansion
valve; 85 indoor heat exchanger; 86 refrigerant piping;
87 compression mechanism; 100, 100-1, 100-2, 100-3, 100A
power converting apparatus; 101 motor driving apparatus;
201 processor; 202 memory; 241 carrier 5 generating unit;
242 reference PWM generating unit; 243 dead time
generating unit; 244 pulse selector; 300 direct-current
voltage source; 311C, 321C boot capacitor; 311D, 321D
boot diode; 311D', 312RD, 321D', 322RD gate voltage
10 suppression diode; 311DC, 312DC, 321DC, 322DC driving
circuit; 311R, 321R boot resistor; 311a, 312a, 321a, 322a
body diode; 312BD body diode voltage; 401, 401A, 402, 402A
bootstrap circuit; 501 first line; 502 second line; 503
third line; 504 fourth line; 506 first connection point;
15 508 second connection point; 600 semiconductor substrate;
601, 603 region; 602 insulating oxide layer; 604 channel;
700 air conditioner.
71
We Claim :
1. A power converting apparatus for converting an
alternating-current power supplied from an alternatingcurrent
power supply into a direct-current power, the power
converting apparatus 5 comprising:
a first line and a second line, each of the first line
and the second line being connected to the alternatingcurrent
power supply;
a first reactor disposed on the first line;
10 a first arm comprising a first switching element, a
second switching element, and a third line having a first
connection point, the first switching element being
connected to the second switching element in series by the
third line, the first connection point being connected to
15 the first reactor by the first line;
a second arm connected in parallel with the first arm
and comprising a third switching element, a fourth
switching element, and a fourth line having a second
connection point, the third switching element being
20 connected to the fourth switching element in series by the
fourth line, the second connection point being connected to
the alternating-current power supply by the second line;
a first capacitor connected in parallel with the
second arm;
25 a first driving circuit outputting a first driving
signal for driving the first switching element;
a bootstrap circuit comprising a second capacitor, the
second capacitor applying a power supply voltage for the
first driving circuit to the first driving circuit; and
30 a diode adjusting the power supply voltage, wherein a
first voltage is lower than a second voltage, the first
voltage being a voltage at which a forward current starts
to flow in the diode, the second voltage being a voltage at
72
which a forward current starts to flow in a body diode
formed in the second switching element.
2. The power converting apparatus according to claim 1,
wherein the diode is provided between the 5 second capacitor
and the first driving circuit.
3. The power converting apparatus according to claim 1,
wherein the diode is connected in parallel with the second
10 switching element.
4. The power converting apparatus according to any one of
claims 1 to 3, wherein the diode is provided in the
bootstrap circuit.
15
5. The power converting apparatus according to any one of
claims 1 to 4, further comprising:
a second driving circuit outputting a second driving
signal for driving the second switching element, wherein
20 the first voltage is set to a value at which a voltage
of the first driving signal is equal to a voltage of the
second driving signal.
6. The power converting apparatus according to any one of
25 claims 1 to 5, wherein a switching frequency of the first
arm is higher than a switching frequency of the second arm.
7. The power converting apparatus according to claim 6,
wherein the switching frequency of the first arm is higher
30 than 266 times of a frequency of the alternating-current
power supply.
8. The power converting apparatus according to claim 6,
73
wherein the switching frequency of the first arm is higher
than 16 kHz.
9. The power converting apparatus according to any one of
claims 1 to 8, wherein the first switching 5 element and the
second switching element are made of wide band gap
semiconductor.
10. The power converting apparatus according to claim 9,
10 wherein the wide band gap semiconductor is silicon carbide
or gallium nitride material.
11. The power converting apparatus according to claim 9 or
10, wherein the third switching element and the fourth
15 switching element are made of silicon carbide semiconductor.
12. The power converting apparatus according to claim 9 or
10, wherein the third switching element and the fourth
switching element are super junction metal-oxide20
semiconductor field-effect transistors.
13. The power converting apparatus according to any one of
claims 1 to 12, wherein at least one of the first arm and
the second arm is configured as a 2-in-1 module.
25
14. The power converting apparatus according to any one of
claims 1 to 13, further comprising:
a current detecting unit detecting a power supply
current output from the alternating-current power supply,
30 wherein
whether or not to permit the third switching element
and the fourth switching element to be ON is determined
depending on the power supply current.
74
15. The power converting apparatus according to claim 14,
wherein when the power supply current is equal to or
smaller than a threshold, the first switching element and
the second switching element are not permitted 5 to be ON,
and when the power supply current is larger than the
threshold, the first switching element and the second
switching element are permitted to be ON.
10 16. The power converting apparatus according to claim 14,
wherein when the power supply current is equal to or
smaller than a threshold, the third switching element and
the fourth switching element are not permitted to be ON,
and when the power supply current is larger than the
15 threshold, the third switching element and the fourth
switching element are permitted to be ON.
17. The power converting apparatus according to any one of
claims 1 to 16, wherein
20 the first arm includes a fifth switching element and a
sixth switching element, the fifth switching element being
connected in series to the sixth switching element,
the fifth switching element is connected in parallel
with the first switching element, and
25 the sixth switching element is connected in parallel
with the second switching element.
18. The power converting apparatus according to claim 17,
wherein
30 the first switching element and the fifth switching
element are driven simultaneously, and
the second switching element and the sixth switching
element are driven simultaneously.
75
19. A motor driving apparatus for driving a motor, the
motor driving apparatus comprising:
the power converting apparatus according to any one of
claims 5 1 to 18; and
an inverter converting a direct-current power output
from the power converting apparatus into an alternatingcurrent
power, and outputting the alternating-current power
to the motor.
10
20. An air conditioner comprising:
the motor; and
the motor driving apparatus according to claim 19.
15 21. The air conditioner according to claim 20, further
comprising: a fan driven by the motor.
22. The air conditioner according to claim 20 further
comprising: a compressor driven by the motor.