FORM 2
THE PATENTS ACT, 1970
(39 of 1970)
&
THE PATENTS RULES, 2003
COMPLETE SPECIFICATION
[See section 10, Rule 13]
ROTARY MACHINE CONTROL DEVICE;
MITSUBISHI ELECTRIC CORPORATION, A CORPORATION
ORGANISED AND EXISTING UNDER THE LAWS OF JAPAN, WHOSE
ADDRESS IS 7-3, MARUNOUCHI 2-CHOME, CHIYODA-KU, TOKYO
1008310, JAPAN
THE FOLLOWING SPECIFICATION PARTICULARLY DESCRIBES THE
INVENTION AND THE MANNER IN WHICH IT IS TO BE PERFORMED
2
DESCRIPTION
Field
[0001] The present disclosure relates to a rotary
machine control device that controls a rotary machine by5
obtaining rotor position information without using a
position sensor for detecting a rotor position.
Background
[0002] Driving a rotary machine in such a manner as to10
deliver sufficient performance of the rotary machine
requires information on the position of the rotor. For
this reason, a rotary machine is conventionally driven
using the position information detected by a position
sensor attached to the rotary machine. Meanwhile, for15
purposes such as further reduction of manufacturing cost of
the rotary machine, size reduction of the rotary machine,
and improvement in reliability of the rotary machine,
technology has recently been developed to drive a rotary
machine without use of a position sensor.20
[0003] One position sensorless control method for a
rotary machine is a technique of applying a high-frequency
signal to the rotary machine. For this technique, a stator
current is first detected upon application of a high-
frequency voltage to the rotary machine. Next, a high-25
frequency current having the same frequency component as
the frequency component of the high-frequency voltage is
extracted. Finally, the rotor position is estimated
utilizing the phenomenon where the inductance of a rotary
machine, i.e., the amplitude of the high-frequency current,30
varies at a frequency twice the electrical angle frequency
of the rotor position. Such method using a high-frequency
signal has the advantage of excellently estimating the
3
rotor position even when the rotary machine is operating at
a zero speed or in a low speed range, but has the
disadvantage of occurrence of torque pulsation and noise
due to the high-frequency voltage superimposed.
[0004] Another technique is also applicable. For5
example, Patent Literature 1 below discloses a technique
for estimating a rotor position from the stator voltage and
the stator current of a rotary machine without applying a
high-frequency signal. For this technique of Patent
Literature 1, first, a stator voltage and a stator current10
are input to an observer. Next, the observer estimates,
from components of the flux linkage, a component rotating
in synchronization with the rotor position. Finally, the
observer computes the rotor position from the phase of the
estimated value, and outputs the thus computed rotor15
position.
Citation List
Patent Literature
[0005] Patent Literature 1: Japanese Patent Application20
Laid-open No. 2006-230174
Summary of Invention
Problem to be solved by the Invention
[0006] The stator voltage for use in estimation of the25
rotor position in conventional technologies such as the
technology of Patent Literature 1 is not an actual voltage,
but a voltage command value that is a command value of the
voltage. The use of the voltage command value unavoidably
causes an error. The actual stator voltage, which is in a30
rectangular waveform, is applied to the rotary machine such
that a value generated by smoothing this stator voltage is
a voltage command value. For conventional technologies, an
4
error in the estimated value of the rotor position occurs
due to that voltage error, which in some case causes a
pulsation component. When a rotary machine is controlled
using an estimated value of the rotor position including
such estimation error, pulsation may occur in torque or5
power, which adversely affects a mechanical system or a
power system connected thereto.
[0007] The present disclosure has been made in view of
the foregoing, and it is an object of the present
disclosure to provide a rotary machine control device10
capable of reducing torque pulsation and power pulsation
resulting from an estimation error that may be included in
an estimated value of the rotor position.
Means to Solve the Problem15
[0008] To solve the problem and achieve the object
described above, a rotary machine control device according
to the present disclosure comprises a voltage applicator, a
current detector, a controller, a pulse width modulator, a
voltage integrator, and a position estimator. The voltage20
applicator is connected between a direct current power
supply and a rotary machine to apply a stator voltage of a
rectangular waveform to the rotary machine by switching on
and off a plurality of switching elements provided for each
phase. The current detector detects a stator current25
flowing between the voltage applicator and a stator winding
of the rotary machine. The controller computes a voltage
command value on the basis of the stator current and a
rotor position, the voltage command value being a command
value of the stator voltage, the stator voltage being a30
voltage applied to the stator winding, the rotor position
being information on a position of a rotor of the rotary
machine. The pulse width modulator generates a gate signal
5
such that a value obtained by smoothing the stator voltage
matches the voltage command value, and uses the gate signal
to control turning on and off each of the switching
elements. The voltage integrator computes a voltage
integration value by integrating the gate signal, the5
voltage integration value being an integrated value of the
stator voltage. The position estimator estimates the rotor
position on the basis of the voltage integration value and
the stator current.
10
Effects of the Invention
[0009] The rotary machine control device according to
the present disclosure provides the advantage of reducing
the torque pulsation and power pulsation resulting from the
estimation error that may be included in the estimated15
value of the rotor position.
Brief Description of Drawings
[0010] FIG. 1 is a diagram illustrating an example
configuration of a rotary machine control device according20
to a first embodiment.
FIG. 2 is a diagram illustrating an example
configuration of a main circuit of a three-phase inverter,
utilized as the voltage applicator of FIG. 1.
FIG. 3 is a first diagram for describing operation of25
the pulse width modulator (PWM modulator) illustrated in
FIG. 1.
FIG. 4 is a second diagram for describing operation of
the PWM modulator illustrated in FIG. 1.
FIG. 5 is a diagram illustrating an example30
configuration of the voltage integrator illustrated in FIG.
1.
FIG. 6 is a diagram illustrating an example
6
configuration of the position estimator illustrated in FIG.
1.
FIG. 7 is a first diagram for describing a
relationship between a switching period and a control
computation period in the first embodiment.5
FIG. 8 is a second diagram for describing the
relationship between the switching period and the control
computation period in the first embodiment.
FIG. 9 is a first diagram for describing an effect
produced when the PWM modulator illustrated in FIG. 110
switches a switching frequency.
FIG. 10 is a second diagram for describing the effect
produced when the PWM modulator illustrated in FIG. 1
switches the switching frequency.
FIG. 11 is a diagram illustrating an example15
configuration of a rotary machine control device according
to a second embodiment.
FIG. 12 is a diagram for describing operation of the
PWM modulator illustrated in FIG. 11.
FIG. 13 is a diagram illustrating an example20
configuration of the voltage integrator illustrated in FIG.
11.
FIG. 14 is a diagram for describing operation when the
voltage integrator in the second embodiment includes no
low-pass filter (LPF).25
FIG. 15 is a diagram for describing operation when the
voltage integrator in the second embodiment includes an LPF.
FIG. 16 is a diagram illustrating an example
configuration of the position estimator illustrated in FIG.
11.30
FIG. 17 is a diagram illustrating an example
configuration of a rotary machine control device according
to a third embodiment.
7
FIG. 18 is a diagram illustrating an example
configuration of the voltage integrator illustrated in FIG.
17.
FIG. 19 is a diagram illustrating an example
configuration of the position estimator illustrated in FIG.5
17.
FIG. 20 is a diagram illustrating a first example
hardware configuration for implementing each functionality
of the control devices according to the first through third
embodiments.10
FIG. 21 is a diagram illustrating a second example
hardware configuration for implementing each functionality
of the control devices according to the first through third
embodiments.
15
Description of Embodiments
[0011] A rotary machine control device according to
embodiments of the present disclosure will be described in
detail below with reference to the accompanying drawings.
[0012] First Embodiment.20
FIG. 1 is a diagram illustrating an example
configuration of a rotary machine control device
(hereinafter referred to simply as “control device” as
appropriate) 100 according to a first embodiment. The
control device 100 according to the first embodiment is25
configured to include a voltage applicator 3, a current
detector 4, a controller 5, a PWM modulator 6, a voltage
integrator 7, and a position estimator 8.
[0013] The voltage applicator 3 is connected between a
direct current (DC) power supply 1 and a rotary machine 2.30
The DC power supply 1 is a power supply source that
provides drive power for the rotary machine 2.
[0014] The rotary machine 2 is a three-phase electric
8
motor having inductance varying with the rotor position.
The rotary machine 2 includes a stator 2a and a rotor 2b
disposed inside the stator 2a. The stator 2a includes
stator windings of phase u, phase v, and phase w. The
rotary machine 2 also functions as a three-phase power5
generator depending on the operation mode. A synchronous
reluctance motor is herein described as being an example of
the rotary machine 2, but the rotary machine 2 can be a
motor other than a synchronous reluctance motor. Note that
the orientation of the rotor 2b that maximizes the10
inductance is herein defined as d-axis, and the orientation
of the rotor 2b that minimizes the inductance is herein
defined as q-axis. The rotor position is given with
respect to the d-axis.
[0015] The current detector 4 is disposed between the DC15
power supply 1 and the rotary machine 2. The current
detector 4 detects stator currents isu, isv, and isw. The
stator currents isu, isv, and isw flow between the voltage
applicator 3 and the stator windings of the rotary machine
2.20
[0016] The voltage applicator 3 switches on and off
multiple switching elements provided for each phase to
thereby apply a stator voltage of a rectangular waveform to
the rotary machine 2. The stator voltage is a voltage
applied to each of the stator windings of the rotary25
machine 2. The voltage applicator 3 is herein described as
being a three-phase inverter.
[0017] The controller 5 computes voltage command values
vsu*, vsv*, and vsw* on the basis of the rotor position and
the stator currents isu, isv, and isw detected by the current30
detector 4. The rotor position is information on the
position of the rotor 2b. The voltage command values vsu*,
vsv*, and vsw* are command values of the stator voltages for
9
driving the rotary machine 2. The stator voltages output
by the voltage applicator 3 are controlled through the
voltage command values vsu*, vsv*, and vsw*.
[0018] The PWM modulator 6 generates gate signals gu, gv,
and gw such that values obtained by smoothing the stator5
voltages each having a rectangular waveform output by the
voltage applicator 3 match the voltage command values vsu*,
vsv*, and vsw*, and uses these gate signals gu, gv, and gw to
control turning on and off each of the switching elements.
[0019] The voltage integrator 7 integrates the gate10
signals gu, gv, and gw to thereby compute voltage
integration values vsui, vsvi, and vswi. The voltage
integration values vsui, vsvi, and vswi are integrated values
of the stator voltages.
[0020] The position estimator 8 computes an estimated15
rotor position θ^r on the basis of the voltage integration
values vsui, vsvi, and vswi and the stator currents isu, isv,
and isw. The estimated rotor position θ^r is an estimated
value of the rotor position, the rotor position being
information on the position of the rotor 2b. Note that the20
estimated rotor position θ^r is herein given by a value
converted into an electrical angle.
[0021] FIG. 2 is a diagram illustrating an example
configuration of a main circuit of the three-phase inverter
used as the voltage applicator 3 of FIG. 1. In FIG. 2, a25
switching element 31 is the switching element on the
positive side of phase u, and a switching element 32 is the
switching element on the negative side of phase u.
Similarly, switching elements 33 and 34 are respectively
the switching elements on the positive and negative sides30
of phase v, and switching elements 35 and 36 are
respectively the switching elements on the positive and
negative sides of phase w. An example of each of the
10
switching elements 31 to 36 is the insulated gate bipolar
transistor (IGBT) in the illustration, but a switching
element other than an IGBT can also be used. An example of
switching element other than an IGBT is a metal oxide
semiconductor field effect transistor (MOSFET). A diode is5
connected to both ends of each switching element in inverse
parallel. The term “inverse parallel” refers to a
connection form in which the anode of a diode is connected
to the emitter of an IGBT while the cathode of the diode is
connected to the collector of the IGBT.10
[0022] An operation of the controller 5 will next be
specifically described. The controller 5 is configured to
include a current command value computing element 501, a
three-phase to two-phase converter 502, a rotational
coordinate converter 503, a d-q current controller 504, a15
rotational coordinate reverse converter 505, and a two-
phase to three-phase converter 506. The controller 5
receives a torque command value T*. The controller 5
computes the voltage command values vsu*, vsv*, and vsw* such
that the rotary machine 2 outputs a torque corresponding to20
the torque command value T*.
[0023] The current command value computing element 501
computes current command values isd* and isq*. The current
command values isd* and isq* are each a command value of the
stator current necessary for the rotary machine 2 to output25
a torque corresponding to the torque command value T*. The
current command values isd* and isq* are each a computed
value in a rotational coordinate system that rotates in
synchronization with the rotational speed of the rotary
machine 2. Note that the current command values isd* and30
isq* are computed to minimize the current root-mean-square
(RMS) value for that torque. That is, the current command
values isd* and isq* are computed to minimize copper loss of
11
the rotary machine 2 for that torque.
[0024] The three-phase to two-phase converter 502
converts, by three-phase to two-phase conversion, the
stator currents isu, isv, and isw in a three-phase coordinate
system into stator currents isα and isβ in a two-phase5
coordinate system. The two-phase coordinate system that is
a static coordinate system. Note that this conversion
process is herein performed using a transformation matrix
C32 shown in Equation (1) below.
[0025] Formula 1:10
[0026] The rotational coordinate converter 503 converts
the stator currents isα and isβ in the two-phase coordinate
system into stator currents isd and isq in a rotational
coordinate system, by rotational coordinate conversion15
using the estimated rotor position θ^r. Note that this
conversion process is herein performed using a
transformation matrix Cdq(θr) shown in Equation (2) below.
[0027] Formula 2:
20
[0028] The d-q current controller 504 provides control
such that the stator currents isd and isq match the current
command values isd* and isq*, and computes voltage command
values vsd* and vsq* in the rotational coordinate system.
This control can be made using proportional-integral25
control. Note that control other than proportional-
integral control can be used.
[0029] The rotational coordinate reverse converter 505
12
uses the estimated rotor position θ^r to convert the
voltage command values vsd* and vsq* in the rotational
coordinate system into voltage command values vsα* and vsβ*
in the two-phase coordinate system by rotational coordinate
inverse conversion. Note that this inverse conversion5
process is herein performed using an inverse transformation
matrix Cdq–1(θ^r) shown in Equation (3) below.
[0030] Formula 3:
[0031] The two-phase to three-phase converter 50610
converts the voltage command values vsα* and vsβ* in the
two-phase coordinate system into the voltage command values
vsu*, vsv*, and vsw* in the three-phase coordinate system by
two-phase to three-phase conversion. Note that this
conversion process is herein performed using a15
transformation matrix C23 shown in Equation (4) below.
[0032] Formula 4:
[0033] FIG. 3 is a first diagram for describing
operation of the PWM modulator 6 illustrated in FIG. 1.20
FIG. 3 illustrates a waveform of phase u as an example
waveform of one of the three phases.
[0034] In FIG. 3, the top portion illustrates a waveform
of the phase-u voltage command value vsu*, and a waveform
of a carrier signal “c”. The phase-u voltage command value25
vsu* is the voltage command value of phase u, and the
carrier signal “c” is a triangular wave. The upper middle
13
portion illustrates a waveform of a phase-u upper gate
signal gup. The phase-u upper gate signal gup is the gate
signal on the upper side of phase u. The lower middle
portion illustrates a waveform of a phase-u lower gate
signal gun. The phase-u lower gate signal gun is the gate5
signal on the lower side of phase u. The bottom portion
illustrates a waveform of a phase-u voltage vsu. The
phase-u voltage vsu is the stator voltage of phase u. The
voltage vdc, which is the supply voltage, is the voltage of
the DC power supply 1. In this case, as illustrated in FIG.10
3, a voltage of vdc/2, i.e., half the supply voltage vdc, is
the step width of a phase voltage. The phase-u voltage
command value vsu* and the phase-u voltage vsu vary within a
range of ±vdc/2.
[0035] The PWM modulator 6 compares the phase-u voltage15
command value vsu* with the carrier signal c. When the
phase-u voltage command value vsu* is greater than the
value of the carrier signal c, the PWM modulator 6 sets the
phase-u upper gate signal gup to H-level, and sets the
phase-u lower gate signal gun to L-level. Alternatively,20
when the phase-u voltage command value vsu* is lower than
or equal to the value of the carrier signal c, the PWM
modulator 6 sets the phase-u upper gate signal gup to L-
level, and sets the phase-u lower gate signal gun to H-
level. In this respect, H represents “High” and L25
represents “Low”. When the phase-u upper gate signal gup=H
and the phase-u lower gate signal gun=L, the PWM modulator
6 turns on the switching element 31 on the positive side of
phase u, and turns off the switching element 32 on the
negative side of phase u, in the voltage applicator 3.30
Alternatively, when the phase-u upper gate signal gup=L and
the phase-u lower gate signal gun=H, the PWM modulator 6
turns off the switching element 31 on the positive side of
14
phase u, and turns on the switching element 32 on the
negative side of phase u, in the voltage applicator 3.
Operations for phase v and for phase w are similar to the
operation for phase u.
[0036] The phase-u voltage vsu actually output is a5
voltage value that is an average phase-u voltage command
value vsu* over a switching period Tsw. Note that the
switching period Tsw is equal to the carrier period, i.e.,
the period of the carrier signal c. In the case of
switching on and off switching elements on the positive and10
negative sides, a dead time, during which the switching
elements on both of these sides are in off states, is set
in general to prevent those switching elements from
becoming ON states simultaneously. However, FIG. 3 omits
illustration of such a dead time. In addition, for15
simplicity of illustration, the neutral point voltage,
which is the average value of the three-phase voltage, is
not illustrated in the waveform of the phase-u voltage vsu
illustrated in FIG. 3.
[0037] The first embodiment employs a technique of20
synchronizing a switching frequency fsw with an integer
multiple of a fundamental wave frequency fs of the
rotational speed of the rotary machine 2. The switching
frequency fsw is the reciprocal of the switching period Tsw.
This synchronization technique reduces a low-order harmonic25
component even when the switching frequency fsw is not
sufficiently higher than the fundamental wave frequency fs.
This enables supplying the rotary machine 2 with a stator
voltage having low distortion and a stator current having
low distortion. Note that the switching frequency fsw not30
sufficiently high refers to, for example, the switching
frequency fsw in the range from the fundamental wave
frequency fs to 15 times the fundamental wave frequency fs.
15
[0038] FIG. 4 is a second diagram for describing
operation of the PWM modulator 6 illustrated in FIG. 1.
FIG. 4 illustrates a relationship between the fundamental
wave frequency fs and the switching frequency fsw. As
illustrated in FIG. 4, the switching frequency fsw is5
varied depending on the fundamental wave frequency fs. The
frequency fsmax is the maximum fundamental wave frequency.
The frequency fswmax is the maximum switching frequency.
The example illustrated in FIG. 4 is where: the switching
frequency fsw is 15 times the fundamental wave frequency fs10
when the fundamental wave frequency fs exceeds fsmax/20 but
is not more than fsmax/15; and nine times the fundamental
wave frequency fs when the fundamental wave frequency fs
exceeds fsmax/15 but is not more than fsmax/9. In addition,
the switching frequency fsw is three times the fundamental15
wave frequency fs when the fundamental wave frequency fs
exceeds fsmax/9 but is not more than fsmax/3; and the
switching frequency fsw is equal to the fundamental wave
frequency fs when the fundamental wave frequency fs is
higher than fsmax/3. Note that the frequency fswmax is simply20
used when the fundamental wave frequency fs is less than or
equal to fsmax/20. In addition, assuming that a carrier
signal common to the three phases is used, the switching
frequency fsw is one, three, nine, or 15 times the
fundamental wave frequency.25
[0039] FIG. 5 is a diagram illustrating an example
configuration of the voltage integrator 7 illustrated in
FIG. 1. FIG. 5 illustrates a configuration including
integrators 701, 703, and 705, and multipliers 702, 704,
and 706. As one of a positive-side gate signal and a30
negative-side gate signal is an inverted signal of the
other, FIG. 5 illustrates only components relating to the
positive-side gate signals.
16
[0040] An operation for phase u will first be described.
The positive-side gate signal gup is input to the
integrator 701. The integrator 701 integrates the
positive-side gate signal gup, taking a high level H as +1
and a low level L as –1. An integration time, which is the5
time period for performing integration, is a time period
from a time earlier than the current time “t” by Tgi to the
current time “t”. After integration of the gate signal,
the multiplier 702 multiplies the integrated value from the
integrator 701 by vdc/2, i.e., a half of the supply voltage10
vdc, and outputs the product as the phase-u voltage
integration value vsui. This enables computing an
integrated value of the instantaneous value of the phase-u
voltage vsu of a rectangular waveform. The integrators 703
and 705 and the multipliers 704 and 706 perform similar15
operations for the other phases, i.e., for phase v and
phase w. As a result, the phase-v voltage integration
value vsvi is output from the multiplier 704, and the phase-
w voltage integration value vswi is output from the
multiplier 706. Note that these voltage integration values20
vsui, vsvi, and vswi for their respective phases will be
designated as voltage integration value vsxi when these
values are not distinguished by phase.
[0041] As the voltage integration value is a computed
integral of the instantaneous value of the stator voltage25
vs of a rectangular waveform, the voltage integration value
vsxi is computed at a computation frequency set to a value
sufficiently higher than the switching frequency fsw of the
gate signal generated by the PWM modulator 6. In order
that the computation frequency is sufficiently high, the30
computation frequency is preferably 25 or more times the
switching frequency fsw, and more preferably 100 or more
times the switching frequency fsw. In addition, a burden
17
of this integration operation itself is small. Even with a
high computation frequency, thus, a computing device bears
a small computation burden as compared to the amount of
calculation necessary to control the rotary machine 2. The
value of the supply voltage vdc for use in this operation5
can be a detection value, a rated value, or an expected
value during use.
[0042] A principle for the position estimator 8 to
estimate the rotor position and the rotational speed will
next be described. A rotary machine model that formulates10
characteristics of the rotary machine 2 is expressed by
Equations (5) and (6) below in a two-phase coordinate
system.
[0043] Formula 5:
15
Formula 6:
[0044] In the equations above, vsαβ is the stator voltage,
isαβ is the stator current, ψsαβ is the flux linkage, and Rs
is the winding resistance. The superscript “αβ” indicates20
that the value is a value in a two-phase coordinate system.
[0045] In addition, the inductance of the rotary machine
2 varies depending on the rotor position. Equation (6)
above is expressed using an inductance averaged component
Lsavg and an inductance variable component Lsvar, the25
inductance averaged component Lsavg representing an
inductance component invariable regardless of the rotor
position, the inductance variable component Lsvar
representing an inductance component that varies at a
frequency twice the electrical angle frequency of the rotor30
position. Such inductance averaged component Lsavg and
18
inductance variable component Lsvar are expressed by
Equations (7) and (8) below using inductance Lsd in the d-
axis direction and inductance Lsq in the q-axis direction.
[0046] Formula 7:
5
Formula 8:
[0047] According to the rotary machine model expressed
by Expressions (5) and (6) above, a product of the
inductance Lsq in the q-axis direction and the stator10
current isαβ is subtracted from the flux linkage ψsαβ to
thereby extract a d-axis-based active flux ψafdαβ, as shown
by Equation (9) below.
[0048] Formula 9:
15
[0049] The d-axis-based active flux ψafdαβ, which is a
component of the flux linkage ψsαβ, rotates in
synchronization with the rotor position.
[0050] In addition, the stator current isαβ can be
expressed by Equation (10) below using the current RMS20
value Iph thereof and a conduction angle φi, which is the
angular difference from the rotor position.
[0051] Formula 10:
[0052] Substitution of Expressions (6) and (10) above25
into the right-hand side of Equation (9) above results in
Equation (11) below, Equation (11) expressing the d-axis-
based active flux ψafdαβ in the two-phase coordinate system.
[0053] Formula 11:
19
[0054] As shown by Equation (11) above, the active flux
ψafdαβ is a component produced by multiplication of the
inductance variable component Lsvar by the stator current isd.
In addition, since the active flux ψafdαβ of Equation (11)5
above is based on the d-axis direction, the rotor position
can be estimated by inputting this to a publicly known
observer.
[0055] Note that instead of Equation (9) above, a q-
axis-based active flux ψafqαβ expressed by Equation (12)10
below can also be used. The q-axis-based active flux ψafqαβ
is a component obtained by subtracting a product of the
inductance Lsd in the d-axis direction and the stator
current isαβ from the flux linkage ψsαβ.
[0056] Formula 12:15
[0057] As in the d-axis base, substitution of
Expressions (6) and (10) above into the right-hand side of
Equation (12) above results in Equation (13) below,
Equation (13) expressing the q-axis-based active flux ψafqαβ
20
in the two-phase coordinate system.
[0058] Formula 13:
[0059] Since the q-axis-based active flux ψafqαβ
expressed by Equation (13) above is based on the q-axis25
direction, the rotor position can be estimated by inputting
this to a publicly known observer.
[0060] In the present embodiment, the d-axis-based
active flux ψafdαβ is input to the observer disclosed in
foregoing Patent Literature 1 to thereby estimate the rotor30
20
position. Note that the rotor position can be estimated
using an element other than the observer disclosed in
Patent Literature 1.
[0061] The observer expressed by Equation (14) in Patent
Literature 1 can be expressed by Equation (14) below using5
parameters used herein.
[0062] Formula 14:
[0063] In Equation (14) above, ψ^safddq is an estimated
value of the d-axis-based active flux. This observer is10
expressed in a rotational coordinate system synchronized
with the estimated rotor position. The superscript “dq”
indicates that the value is a value in the rotational
coordinate system. In addition, ωr in Equation (14) above
represents the rotational angular velocity, and ωs15
represents the rotational angular velocity in the
rotational coordinate system. Moreover, the symbol J in
Equation (14) above is a transformation matrix expressed by
Equation (15) below.
[0064] Formula 15:20
[0065] Setting an observer gain in the observer
expressed by Equation (14) above in accordance with the
teachings of Patent Literature 1 obtains an estimated value
of the d-axis-based active flux ψafdαβ. In addition, since25
the d-axis-based active flux ψafdαβ is synchronized with the
rotor position θr, as shown by Equation (11) above,
computation of arc tangent of the two components in
Equation (11) above estimates the rotor position.
[0066] Moreover, both sides of Equation (14) above is30
divided by “s” to thereby obtain Equation (16) below.
21
[0067] Formula 16:
[0068] Note that Equation (14) above, which is an
equation expressed using an observer, is basically
expressed integrating a term including the stator voltage5
vsdq and the stator current isdq. In addition, the first
term on the right-hand side of Equation (16) that is a
modified Equation (14) above is an integral value of the
stator voltage vsdq. In view of this, the first embodiment
uses, for this part, the voltage integration value vsxi10
computed by the voltage integrator 7. Note that the
conventional technology described in Patent Literature 1
uses a voltage command value vsdq* as the stator voltage vsdq.
[0069] FIG. 6 is a diagram illustrating an example
configuration of the position estimator 8 illustrated in15
FIG. 1. The position estimator 8 can be configured to
include three-phase to two-phase converters 801 and 803,
rotational coordinate converters 802 and 804, and an
observer 805.
[0070] The three-phase to two-phase converter 80120
converts the voltage integration values vsui, vsvi, and vswi
output from the voltage integrator 7, into voltage
integration values vsαi and vsβi in the two-phase coordinate
system by three-phase to two-phase conversion. The
rotational coordinate converter 802 converts the voltage25
integration values vsαi and vsβi in the two-phase coordinate
system, into voltage integration values vsdi and vsqi in the
rotational coordinate system by rotational coordinate
conversion using the estimated rotor position θ^r. Note
that the estimated rotor position θ^r is feedback of an30
output of the observer 805, i.e., feedback of the estimated
22
rotor position θ^r that is an output of the position
estimator 8.
[0071] In addition, the three-phase to two-phase
converter 803 converts the stator currents isu, isv, and isw
in the three-phase coordinate system, into the stator5
currents isα and isβ in the two-phase coordinate system by
three-phase to two-phase conversion. The rotational
coordinate converter 804 converts the stator currents isα
and isβ in the two-phase coordinate system, into the stator
currents isd and isq in the rotational coordinate system by10
rotational coordinate conversion using the estimated rotor
position θ^r.
[0072] Using the above-mentioned observer, the observer
805 computes the estimated rotor position θ^r and an
estimated rotational angular velocity ω^r. The estimated15
rotational angular velocity ω^r is an estimated value of
the rotational angular velocity. Note that Patent
Literature 1 describes that the rotor position and the
rotational angular velocity are estimated through a phase
synchronizer as well as through the observer. The observer20
805 herein is defined as having functionality of a phase
synchronizer, too. In addition, the position estimator 8
is conveniently described as receiving values in the three-
phase coordinate system, but the received values are not
limited thereto. The position estimator 8 can receive25
values in the rotational coordinate system from the
controller 5 or from the voltage integrator 7.
[0073] Moreover, assuming that the computation period of
processing of computing the flux linkage ψs, using the
observer is denoted by Tpsi1, this computation period Tpsi130
is not an integer multiple of half of the switching period
Tsw. In addition, assuming that the computation period of
processing of computing the estimated rotor position θ^r
23
after the processing of computation of the flux linkage ψs
is denoted by Tpsi2, this computation period Tpsi2 is also
not an integer multiple of half of the switching period Tsw.
Moreover, assuming that the integration time, which is the
time period of processing of integrating the voltage5
integration value vsxi, is denoted by Tgi, this integration
time Tgi is also not an integer multiple of half of the
switching period Tsw.
[0074] A relationship between the switching period Tsw
and a control computation period Tpsi will next be described10
with reference to the drawings of FIGs. 7 and 8. FIG. 7 is
a first diagram for describing a relationship between the
switching period Tsw and the control computation period Tpsi
in the first embodiment. FIG. 8 is a second diagram for
describing the relationship between the switching period15
Tsw and the control computation period Tpsi in the first
embodiment. Note that the description it is herein made
assuming that the computation period Tpsi1 of the flux
linkage ψs, the computation period Tpsi2 of the estimated
rotor position θ^r, and the integration time Tgi of the20
voltage integration value vsxi are equal to one another, and
that the control computation period Tpsi is also equal to
each of the computation period Tpsi1 of the flux linkage ψs,
the computation period Tpsi2 of the estimated rotor position
θ^r, and the integration time Tgi of the voltage integration25
value vsxi.
[0075] Control operation of a rotary machine generally
uses a command value instead of a detection value, as a
value of the stator voltage. When the control computation
period Tpsi is an integer multiple of half of the switching30
period Tsw, the voltage command value and the value
obtained by smoothing the actual voltage will be equal to
each other for each control computation period Tpsi. Note
24
that even when the average value of an actual voltage is
used in the smoothing operation, the voltage command value
will be almost equal to the average value.
[0076] The switching period Tsw and the control
computation period Tpsi are illustrated as having a5
relationship of Tpsi=1×(Tsw/2) in FIG. 7, and as having a
relationship of Tpsi=3×(Tsw/2) in FIG. 8. The top portion
of each figure illustrates a waveform of the phase-u
voltage command value vsu* and a waveform of the carrier
signal c. The bottom portion of each figure illustrates a10
waveform of the phase-u voltage vsu. The phase-u voltage
command value vsu* has a sinusoidal waveform.
[0077] FIGs. 7 and 8 both show that the averaged phase-u
voltage vsu for the control computation period Tpsi is
almost equal to the phase-u voltage command value vsu*. It15
is also shown that when the control computation period Tpsi
is not an integer multiple of Tsw/2, the phase-u voltage vsu
smoothed for each control computation period Tpsi will not
match the phase-u voltage command value vsu*. Meanwhile,
the voltage integration value vsxi is computed at a20
computation frequency sufficiently higher than the
switching frequency fsw, in which case the voltage
integration value extracted at any time is an accurate
voltage integration value vsxi obtained in the integration
time Tgi immediately before the extraction. In addition,25
although computation processing of the voltage integration
value vsxi is performed at a high computation frequency, the
burden of computation will be small because this
computation processing is a mere integration operation and
allows for the extraction of the voltage integration value30
vsxi at any time.
[0078] As described above, the switching frequency fsw
is set to an integer multiple of the fundamental wave
25
frequency fs. In this respect, the fundamental wave
frequency fs of the rotary machine 2 is not constant, but
varies continuously. For this reason, the carrier
frequency, which is equivalent to the switching frequency
fsw, is required to vary in real time in accordance with5
the change in the fundamental wave frequency fs. In order
to make the control computation period Tpsi equal to an
integer multiple of half of the switching period Tsw under
a typical control on a rotary machine, the control
computation period Tpsi will need to vary sequentially in10
real time. To this end, the control computation period Tpsi
should be changed during which the computation is conducted
with the variable period, which will result in a high
amount of control computation and a complex control design
as well.15
[0079] In view of this, in the first embodiment, the
control computation period Tpsi is set to a fixed value, and
is not sequentially adjusted to an integer multiple of half
of the switching period Tsw. This reduces the amount of
computation in control computation, eliminates the need for20
a computing device such as a costly microprocessor, and
enables a relatively simple control design. This results
in the voltage command value vs* including an error
relative to the actual voltage.
[0080] As described above, when the control computation25
period Tpsi is not adjusted to an integer multiple of half
of the switching period Tsw, the stator voltages can
include an error. When the stator voltages include an
error, the flux linkage ψs computed using these values will
also include an error. In addition, the computation of the30
flux linkage ψs, which is basically integration processing,
has a particularly large effect on near-DC components
ranging from a direct current to a low-frequency component.
26
In addition, an error in a near-DC component in each of the
stator voltages will appear as an error near the
fundamental wave frequency fs when converted into those in
the rotational coordinate system rotating at the
fundamental wave frequency fs in synchronization with the5
rotor position. An error near the fundamental wave
frequency fs also occurs in the estimated rotor position θ^r
because the position estimation is performed using the flux
linkage ψs in the rotational coordinate system, more
accurately, the d-axis-based active flux. Thus, use of the10
estimated rotor position θ^r having a pulsating error in
controlling the rotary machine 2 will cause the torque and
the power to pulsate. In contrast, the position estimator
8 of the first embodiment uses the voltage integration
values vsxi instead of the voltage command values vsu*, vsv*,15
and vsw* in computing the flux linkage ψs and the estimated
rotor position θ^r. As a result, it becomes possible to
perform position estimation, removing the pulsation of
torque and power that results from an error near the
fundamental wave frequency fs.20
[0081] An effect produced when the PWM modulator 6
switches the switching frequency fsw will next be described
with reference to FIGs. 9 and 10. FIG. 9 is a first
diagram for describing an effect produced when the PWM
modulator 6 illustrated in FIG. 1 switches the switching25
frequency fsw. FIG. 10 is a second diagram for describing
the effect produced when the PWM modulator 6 illustrated in
FIG. 1 switches the switching frequency fsw.
[0082] FIG. 9 illustrates waveforms of the flux linkage
ψs when the fundamental wave components of the stator30
voltage vs have the same amplitudes under the conditions
that: the switching frequency fsw is 15fs; and the switching
frequency fsw is 9fs such that the waveforms of the flux
27
linkage ψs are compared phase-by-phase. As illustrated in
FIG. 9, it is seen that even when the fundamental wave
components are the same, the instantaneous values of the
two waveforms differ from each other. When the switching
frequency fsw is switched with the instantaneous values5
differing from each other, the difference present at the
time of the switching of the switching frequency fsw still
remains as an error in the DC component after the switching
of the switching frequency fsw because the flux linkage ψs
is basically an integration value of the stator voltage vs.10
Similarly, FIG. 10 illustrates waveforms of the flux
linkage ψs when the fundamental wave components of the
stator voltage vs have the same amplitudes under the
conditions that: the switching frequency fsw is 9fs; and the
switching frequency fsw is 3fs. It is also seen from FIG.15
10 that even when the fundamental wave components are the
same, the instantaneous values of the two waveforms differ
from each other. Accordingly, when the switching frequency
is switched, the difference present at the time of the
switching of the switching frequency still remains as an20
error in the DC component after the switching of the
switching frequency.
[0083] The conventional technology described in Patent
Literature 1 uses a command value as the stator voltage in
computing the magnetic flux, and estimating the rotor25
position. The stator voltage command value is determined
in consideration of only the fundamental wave component
without consideration of the instantaneous value. As a
result, an error in a near-DC component as described above
occurs when the switching frequency is switched. An error30
in the DC component in the three-phase coordinate system
appears as an error in a near-fundamental-wave-frequency
component in the rotational coordinate system. The rotor
28
position estimated using this causes an error near the
fundamental wave frequency. This will result in a torque
pulsation and a power pulsation in the rotary machine 2.
In contrast, instead of using the voltage command value vs*,
the position estimator 8 of the first embodiment uses the5
voltage integration value vsxi, i.e., an integral of the
instantaneous value, in the processing of computation of
position estimation. It thus becomes possible to estimate
the position, removing the pulsation of torque and of power
resulting from an error near the fundamental wave frequency10
fs.
[0084] Advantages of the foregoing control computation
according to the first embodiment will next be summarized.
First, in the first embodiment, the switching frequency is
synchronized with an integer multiple of the fundamental15
wave frequency fs of the rotary machine 2. This enables
supplying the rotary machine 2 with a stator voltage having
low distortion and a stator current having low distortion
even at a low switching frequency. In addition, in the
first embodiment, neither the computation period Tpsi1 of20
the flux linkage ψs or the computation period Tpsi2 of the
estimated rotor position θ^r used in the observer 805 is
sequentially adjusted to an integer multiple of half of the
switching period Tsw. This reduces the amount of
computation in control computation, eliminates the need for25
a computing device such as a costly microprocessor, and
enables a relatively simple control design. With such
configuration, it becomes possible to use the voltage
integration value vsxi to thereby estimate the rotor
position, reducing an error near the fundamental wave30
frequency fs and pulsation. This results in an
unprecedented outstanding advantage that the control device
100 including no position sensor and providing a low level
29
of torque pulsation and a low level of power pulsation can
be made without need for a costly microprocessor.
[0085] As described above, according to the rotary
machine control device according to the first embodiment,
the voltage integrator integrates a gate signal to thereby5
compute a voltage integration value, and the position
estimator estimates the rotor position on the basis of the
voltage command values and the stator currents. This can
reduce torque pulsation and power pulsation resulting from
an estimation error that can be included in an estimated10
value of the rotor position.
[0086] Note that the rotary machine control device
according to the first embodiment can include a voltage
detector for detecting the stator voltage output by the
voltage applicator. In this case, the voltage integration15
value can be obtained by integration of the detection value
of the stator voltage detected by the voltage detector,
instead of integration of the gate signal. Such operation
also provides an advantage of reduction in torque pulsation
and power pulsation.20
[0087] To obtain an estimated value of the rotor
position, a component rotating in synchronization with the
rotor position is estimated from components of the flux
linkage, and the estimated value of the rotor position is
computed from the phase of that estimated value. The flux25
linkage for use in computing the estimated value of the
rotor position can be computed using the voltage
integration values. Integration operation for the flux
linkage may produce an offset component and cause an error
and a pulsation in the estimated value, but use of the30
technique of the first embodiment makes it possible to
reduce the error and the pulsation that can be included in
the estimated value.
30
[0088] Moreover, according to the rotary machine control
device according to the first embodiment, the PWM modulator
synchronizes the switching frequency of switching on and
off the switching elements with an integer multiple of the
fundamental wave frequency of the rotational speed of the5
rotary machine. This enables supplying the rotary machine
with a stator voltage having low distortion and a stator
current having low distortion. Furthermore, synchronous
PWM provides a large voltage error between the voltage
command value, i.e., the command value of the stator10
voltage, and the actual voltage, but use of this technique
enables the accurate computation of the stator voltage.
This makes it possible to reduce the error and the
pulsation that can be included in the estimated value of
the rotor position.15
[0089] Note that the rotary machine control device
according to the first embodiment can offer the advantages
thereof when the computation period for which the rotor
position is estimated is not an integer multiple of half of
the switching period. When the computation period for20
estimating the rotor position is not adjusted to an integer
multiple of half of the switching period, the estimated
value of the rotor position may include an error, but use
of the technique of the first embodiment makes it possible
to reduce such error.25
[0090] In addition, the rotary machine control device
according to the first embodiment can offer the advantages
thereof when the computation period for which the flux
linkage is computed is not an integer multiple of half of
the switching period. When the computation period for30
computing the flux linkage is not adjusted to an integer
multiple of half of the switching period, the stator
voltage and the stator current may include an error, but
31
the technique of the first embodiment makes it possible to
reduce such error.
[0091] Note that in the rotary machine control device
according to the first embodiment, the computation
frequency, which is the frequency at which the voltage5
integration value is computed, is preferably 25 or more
times the switching frequency. Such setting of the
frequency enables the accurate computation of the voltage
integration value.
[0092] In addition, in the rotary machine control device10
according to the first embodiment, it is desirable that the
switching frequency for the voltage applicator to switch on
and off the switching elements be switched depending on the
fundamental wave frequency of the rotational speed of the
rotary machine. A large voltage error will occur between15
the voltage command value and the stator voltage upon
switching of the switching frequency, but use of this
technique makes it possible to reduce such error.
[0093] Second Embodiment.
FIG. 11 is a diagram illustrating an example20
configuration of a rotary machine control device 100A
according to a second embodiment. Comparison of the
control device 100A according to the second embodiment with
the control device 100 illustrated in FIG. 1 reveals that
FIG. 11 replaces the PWM modulator 6 with a PWM modulator 9,25
the voltage integrator 7 with a voltage integrator 10, and
the position estimator 8 with a position estimator 11. The
other part of the configuration is the same as or
equivalent to the corresponding part of the configuration
of the control device 100. The same or equivalent30
components are designated by like reference characters, and
duplicate description will be omitted.
[0094] The PWM modulator 9 generates the gate signals gu,
32
gv, and gw similarly to the PWM modulator 6 of the first
embodiment, but varies the switching frequency fsw in a
manner different than the PWM modulator 6 does.
Specifically, the PWM modulator 9 varies the switching
frequency fsw depending on the amplitude of the stator5
voltage vs rather than depending on the fundamental wave
frequency fs. Since the magnitude of the flux linkage ψs
for producing torque of the rotary machine 2 does not vary
with the rotational speed, the stator voltage vs of the
rotary machine 2 can be considered as being proportional to10
the fundamental wave frequency fs.
[0095] FIG. 12 is a diagram for describing operation of
the PWM modulator 9 illustrated in FIG. 11. FIG. 12
illustrates a relationship between an amplitude vsdq* of the
voltage command value vs* and the switching frequency fsw.15
In the second embodiment, as illustrated in FIG. 12, the
switching frequency fsw is varied depending on the
amplitude vsdq*. As in FIG. 4, the value vsdqmax is the
maximum amplitude of the amplitude vsdq*, and the frequency
fswmax is the maximum switching wave frequency. The example20
of FIG. 12 is where: the switching frequency fsw is 15
times the fundamental wave frequency fs when the amplitude
vsdq* exceeds vsdqmax/20 but is not more than vsdqmax/15,
inclusive; and nine times the fundamental wave frequency fs
when the amplitude vsdq* exceeds vsdqmax/15 but is not more25
than vsdqmax/9. In addition, the switching frequency fsw is
three times the fundamental wave frequency fs when the
amplitude vsdq* exceeds vsdqmax/9 but is not more than
vsdqmax/3; and the switching frequency fsw is equal to the
fundamental wave frequency fs when the amplitude vsdq* is30
higher than vsdqmax/3. Note that the frequency fswmax is
simply used when the amplitude vsdq* is less than or equal
to vsdqmax/20. In addition, assuming that a carrier signal
33
common to the three phases is used, the switching frequency
fsw is set to be one, three, nine, or 15 times the
fundamental wave frequency. Note that the amplitude vsdq*
can be calculated using Equation (17) below.
[0096] Formula 17:5
[0097] FIG. 13 is a diagram illustrating an example
configuration of the voltage integrator 10 illustrated in
FIG. 11. FIG. 13 illustrates a configuration including
LPFs 1001, 1004, and 1007, integrators 1002, 1005, and 1008,10
and multipliers 1003, 1006, and 1009. As one of a
positive-side gate signal and a negative-side gate signal
is an inverted signal of the other, FIG. 13 illustrates
only components relating to the positive-side gate signals.
[0098] An operation for phase u will first be described.15
The positive-side gate signal gup passes through the LPF
1001, and is subsequently input to the integrator 1002.
The LPF 1001 cuts off high-frequency components of the gate
signal gup to smooth the waveform of the gate signal gup.
The integrator 1002 integrates the gate signal gup, taking20
a high level H as “+1” and a low level L as “–1”. The
integration time Tgi, which is the time period for
performing integration, is a time period from a time
earlier than the current time “t” by Tgi to the current
time “t”. After integration of the gate signal gup, the25
multiplier 1003 multiplies the integrated value from the
integrator 1002 by vdc/2, i.e., a half of the supply
voltage vdc, and outputs the product as the phase-u voltage
integration value vsui. This enables computing an
integrated value of the instantaneous value of the phase-u30
voltage vsu of a rectangular waveform. The integrators
1005 and 1008 and the multipliers 1006 and 1009 perform
34
similar operations for the other phases, i.e., for phase v
and phase w. As a result, the phase-v voltage integration
value vsvi is output from the multiplier 1006, and the
phase-w voltage integration value vswi is output from the
multiplier 1009.5
[0099] As the voltage integration value is a computed
integral of the instantaneous value of the stator voltage
vs of a rectangular waveform, the voltage integration value
vsxi is computed at a computation frequency set to a value
sufficiently higher than the switching frequency fsw of the10
gate signal generated by the PWM modulator 9. In order
that the computation frequency is sufficiently high, the
computation frequency is preferably 25 or more times the
switching frequency fsw, and more preferably 100 or more
times the switching frequency fsw. In addition, a burden15
of this integration operation itself is small. Even with a
high computation frequency, thus, a computing device bears
a small computation burden as compared to the amount of
calculation necessary to control the rotary machine 2. The
value of the supply voltage vdc for use in this operation20
can be a detection value, a rated value, or an expected
value during use.
[0100] The LPFs 1001, 1004, and 1007 can each be a
first-order LPF whose transfer function is expressed by,
for example, Equation (18) below.25
[0101] Formula 18:
[0102] In Equation (18) above, ωc is a cut-off angular
frequency. The LPFs 1001, 1004, and 1007 can each be
implemented by a digital filter or an analog filter.30
Alternatively, the LPFs 1001, 1004, and 1007 can each be a
second-order filter or one that uses a moving average.
35
[0103] Advantages in use of the LPFs 1001, 1004, and
1007 will next be described with reference to FIGs. 14 and
15. The LPFs 1001, 1004, and 1007 are each hereinafter
collectively referred to simply as “LPF”.
[0104] FIG. 14 is a diagram for describing operation5
when the voltage integrator 10 in the second embodiment
includes no LPF. In FIG. 14, the broken line represents a
waveform of the gate signal gup, a circle symbol represents
a sampling point, and the solid line represents a post-
sampling waveform of the gate signal gup. In addition, FIG.10
15 is a diagram for describing operation when the voltage
integrator 10 in the second embodiment includes an LPF. In
FIG. 15, the broken line represents a waveform of the gate
signal gup, the dash dot line represents a waveform of the
gate signal gup having passed through the LPF, a circle15
symbol represents a sampling point, and the solid line
represents a post-sampling waveform of the gate signal gup
having passed through the LPF.
[0105] In the case of FIG. 14, hatched portions A1 and
A2, which are the portions having their areas enclosed by20
the broken line and the solid line, represent the
integration error. Also in the case of FIG. 15, the
portions having their areas enclosed by the broken line and
the solid line together represent the integration error.
Referring to the waveforms of FIG. 15, the post-sampling25
waveform slightly deviates from the waveform of the gate
signal gup having passed through the LPF. Referring to the
waveforms of FIG. 14, the post-sampling waveform greatly
deviates from the waveform of the gate signal gup. In
addition, as the areas of the portions A1 and A2 in FIG. 1430
have the same signs, the sum of the areas of the portions
A1 and A2 represents the integration error for one period
of the gate signal gup. In contrast, in the case of FIG.
36
15, the area of hatched portions B1 and B2 and the area of
a hatched portion B3 have different signs from each other,
and thus the difference between B3 and the sum of B1 and B2
represents the integration error for one period of the gate
signal gup. This demonstrates that use of an LPF makes it5
possible to accurately compute the voltage integration
value vsxi. The use of an LPF makes it possible to more
precisely compute the voltage integration value vsxi even at
a low sampling frequency, i.e., a low computation frequency
at which the voltage integration value vsxi is computed.10
[0106] FIG. 16 is a diagram illustrating an example
configuration of the position estimator 11 illustrated in
FIG. 11. The position estimator 11 can be configured to
include three-phase to two-phase converters 1101 and 1102,
a rotational coordinate converter 1103, a first computing15
element 1104, a first estimator 1105, a second computing
element 1106, and a third computing element 1107.
[0107] The three-phase to two-phase converter 1101
converts, by three-phase to two-phase conversion, the
voltage integration values vsui, vsvi, and vswi in the three-20
phase coordinate system computed by the voltage integrator
10, into the voltage integration values vsαi and vsβi in the
two-phase coordinate system. Similarly, the three-phase to
two-phase converter 1102 converts the stator currents isu,
isv, and isw in the three-phase coordinate system into the25
stator currents isα and isβ in the two-phase coordinate
systems, by three-phase to two-phase conversion. The
rotational coordinate converter 1103 converts the stator
currents isα and isβ in the two-phase coordinate system into
the stator currents isd and isq in the rotational coordinate30
system, by rotational coordinate conversion using the
estimated rotor position θ^r.
[0108] Operational details of the first computing
37
element 1104 and of the first estimator 1105 will next be
described. The first computing element 1104 computes a
flux linkage inductance variable component, and the first
estimator 1105 estimates the flux linkage inductance
variable component.5
[0109] First, the flux linkage ψsαβ of the rotary machine
2 in the two-phase coordinate system is obtained by
Equation (19) below.
[0110] Formula 19:
10
[0111] In addition, the integration operation of
Equation (19) above is expressed by the transfer function
expressed by Equation (20) below.
[0112] Formula 20:
15
[0113] In general, when the flux linkage is computed by
integration, the initial value is usually unknown. Thus,
when the flux linkage is to be computed in a three-phase
coordinate system and in a two-phase coordinate system,
both of which are static coordinate systems, a high-pass20
filter (HPF) having a cut-off frequency sufficiently lower
than the fundamental wave frequency component is utilized.
This technique, that is, the technique of computing the
flux linkage in a static coordinate system, using
integration and an HPF, is herein referred to as25
“incomplete integration”. The transfer function of the
high-pass filter to be used in this incomplete integration
can be expressed by Equation (21) below, where ωhpf
represents the cut-off frequency.
[0114] Formula 21:30
38
[0115] Applying the HPF expressed by Equation (21) above
to Equation (20) above results in Equation (22) below.
[0116] Formula 22:
5
[0117] Equation (22) above is an expression that
expresses a flux linkage ψshpfαβ when an HPF is employed. In
addition, Equation (22) above is modified to result in
Equation (23) below.
[0118] Formula 23:10
[0119] Note that the first term on the right-hand side
of Equation (23) above is the integral value of the stator
voltage vsαβ. In view of this, the second embodiment uses,
for this part, the voltage integration value vsxi computed15
by the voltage integrator 10. Note that the conventional
technology described in Patent Literature 1 uses a voltage
command value vsαβ* as the stator voltage vsαβ.
[0120] Position sensorless control of a synchronous
reluctance motor allows the use of the technique utilizing20
the incomplete integration in computing the flux linkage.
The technique utilizing the incomplete integration provides
a smaller burden of computation than when an observer.
This allows the use of a computing device such as a less
costly microprocessor.25
[0121] In addition, in the second embodiment, the
computation period Tpsi1 for which the flux linkage is
computed using the incomplete integration is not an integer
multiple of half of the switching period Tsw, and the
computation period Tpsi2 for which the estimated rotor30
39
position θ^r is subsequently computed is not an integer
multiple of half of the switching period Tsw, either.
[0122] The flux linkage ψsαβ of the rotary machine 2 is
expressed by Equation (6) above in the two-phase coordinate
system. Rotational coordinate conversion of this flux5
linkage ψsαβ, using the estimated rotor position θ^r results
in Equation (24) below.
[0123] Formula 24:
[0124] In Equation (24) above, the first term is a term10
including the inductance averaged component Lsavg. The
inductance averaged component Lsavg is invariable regardless
of the rotor position. The second term is a term including
the inductance variable component Lsvar. The inductance
variable component Lsvar varies at a frequency twice the15
rotor position.
[0125] The first computing element 1104 obtains a
component corresponding to the second term of Equation (24)
above by calculation. Specifically, that component is
computed using Equation (25) below that is the modification20
to Equation (24) above.
[0126] Formula 25:
[0127] The first term on the right-hand side of Equation
(25) above is obtained by rotational coordinate conversion25
of the flux linkage ψshpfαβ shown by Equation (23) above. In
addition, the second term on the right-hand side of
Equation (25) above represents the first term of Equation
(24) above. FIG. 16 illustrates an example configuration
of the first computing element 1104, but the configuration30
is not limited to this example.
40
[0128] Meanwhile, the first estimator 1105 directly
estimates a component corresponding to the second term of
Equation (24) above. FIG. 16 illustrates an example
configuration of the first estimator 1105. The following
description is made as to a reason why the first estimator5
1105 can be simple, as shown in FIG. 16.
[0129] First, assuming that the second term of Equation
(24) above is an estimated value of the flux linkage
inductance variable component in the rotational coordinate
system, ψ^svardq representing this estimated value can be10
expressed as Equation (26) below.
[0130] Formula 26:
[0131] Approximating the estimated rotor position θ^r to
a value nearly equal to the true value θr of the rotor15
position in Equation (26) above, Equation (26) above can be
simplified as Equation (27) below. Note that FIG. 6
illustrates a configuration of the controller representing
this Equation (27).
[0132] Formula 27:20
[0133] Operational details of the second computing
element 1106 and of the third computing element 1107 will
next be described.
[0134] First, the cross product of the estimated value25
ψ^svardq of the flux linkage inductance variable component
and the computed value ψsvar,calcdq is expressed by Equation
(28) below.
[0135] Formula 28:
30
41
[0136] Approximating the estimated rotor position θ^r to
a value nearly equal to the true value θr of the rotor
position, that is, approximating the estimated rotor
position θ^r to provide θ^r≈θr, in Equation (28) above, the
estimation error “–(θ^r–θr)” of the rotor position can be5
computed using Equation (29) below.
[0137] Formula 29:
[0138] As described above, the second computing element
1106 computes the estimation error “–(θ^r–θr)” of the rotor10
position on the basis of the computed value obtained by
Equation (25) above and the estimated value obtained by
Equation (27) above.
[0139] The estimation error “–(θ^r–θr)” of the rotor
position computed by the second computing element 1106 is15
input to the third computing element 1107. The third
computing element 1107 performs proportional-integral (PI)
control on the estimation error “–(θ^r–θr)” of the rotor
position, and then integrates the result of the PI control
to cause the estimation error “–(θ^r–θr)” of the rotor20
position to converge to zero. The third computing element
1107 thus computes the estimated rotor position θ^r. The
third computing element 1107 also computes the estimated
rotational angular velocity ω^r in the process of causing
the estimation error “–(θ^r–θr)” of the rotor position to25
converge to zero.
[0140] As described above, the rotary machine control
device according to the second embodiment is designed such
that the technique for estimating the rotor position using
a voltage integration value, i.e., an integrated value of30
the stator voltage, is applicable to the configuration for
42
computing the flux linkage in a static coordinate system.
For position sensorless control of the rotary machine that
is a synchronous reluctance motor, the flux linkage is
computed by integration performed in a static coordinate
system. This makes it more likely to generate an offset5
component, and produce an error and pulsation in the
estimated value. Accordingly, the technique of the second
embodiment can be suitably used when a synchronous
reluctance motor is controlled without a position sensor.
[0141] In addition, in the rotary machine control device10
according to the second embodiment, it is desirable that
the voltage integration value be computed after the gate
signal or the detection value of the stator voltage passes
through a low-range filter. Such operation makes it
possible to more precisely compute the voltage integration15
value as well as to reduce the computation burden of
computing the voltage integration value.
[0142] In addition, in the rotary machine control device
according to the second embodiment, it is desirable that
the switching frequency for the voltage applicator to20
switch on and off the switching elements be switched
depending on the amplitude of the voltage command value. A
large voltage error will occur between the voltage command
value and the stator voltage upon switching of the
switching frequency, but use of this technique can reduce25
such error.
[0143] Advantages of the foregoing control computation
according to the second embodiment will next be summarized.
First, in the second embodiment, neither the computation
period Tpsi1 of the flux linkage ψs computed using Equation30
(23) above using the incomplete integration, nor the
computation period Tpsi2 of the estimated rotor position θ^r
is an integer multiple of half of the switching period Tsw.
43
In this case, the voltage command value vs* does not match
the value obtained by smoothing the actual voltage. As a
result, the voltage command value vs* includes an error
relative to the actual voltage. An error therefore also
occurs in the flux linkage ψs computed using a conventional5
technology. Moreover, in the second embodiment, the flux
linkage ψs is computed using the incomplete integration
rather than using an observer for making the flux linkage
ψs converge to the true value. This will result in a large
error in the flux linkage ψs, and also relatively slow10
convergence to the true value. In addition, since the flux
linkage is computed basically on the basis of integration
processing, such error will be large in components ranging
from DC to a low-frequency component. This will generate a
large error near the fundamental wave frequency fs in the15
rotational coordinate system, and will also generate a
large error in the estimated rotor position θ^r near the
fundamental wave frequency fs. To address this problem,
the position estimator 11 of the second embodiment uses the
voltage integration value vsxi rather than the voltage20
command value vs* in computing the flux linkage ψs and the
estimated rotor position θ^r. This makes it possible to
estimate the rotor position, reducing the error and
pulsation at frequencies near the fundamental wave
frequency. In addition, the position estimator 11 of the25
second embodiment computes the flux linkage, using
incomplete integration without using an observer. This
makes it possible to further reduce the computation burden
as compared to the first embodiment. Thus, use of the
technique of the second embodiment provides an30
unprecedented outstanding advantage that the control device
100A including no position sensor and providing a low level
of torque pulsation and a low level of power pulsation can
44
be made without the need for a costly microprocessor.
[0144] Third Embodiment.
FIG. 17 is a diagram illustrating an example
configuration of a rotary machine control device 100B
according to a third embodiment. Comparison of the control5
device 100B according to the third embodiment with the
control device 100 illustrated in FIG. 1 reveals that FIG.
17 replaces the voltage integrator 7 with a voltage
integrator 13, and the position estimator 8 with a position
estimator 14. In addition, the control device 100B further10
includes a voltage detector 12. The voltage detector 12
detects the stator voltage vs of a rectangular waveform
applied by the voltage applicator 3 to the rotary machine 2.
The other part of the configuration is the same as or
equivalent to the corresponding part of the configuration15
of the control device 100. The same or equivalent
components are designated by like reference characters, and
duplicate description will be omitted.
[0145] FIG. 18 is a diagram illustrating an example
configuration of the voltage integrator 13 illustrated in20
FIG. 17. FIG. 18 illustrates a configuration including
integrators 1301, 1302, and 1303. As one of a positive-
side gate signal and a negative-side gate signal is an
inverted signal of the other, FIG. 18 illustrates only
components relating to the positive-side gate signals.25
[0146] An operation for phase u will first be described.
The phase-u voltage vsu, which is a detection value of the
voltage detector 12, is input to the integrator 1301. The
integrator 1301 integrates the phase-u voltage vsu. The
integration time Tgi, which is the time period for30
performing integration, is a time period from a time
earlier than the current time “t” by Tgi to the current
time “t”. The integrators 1302 and 1303 perform similar
45
operations for the other phases, i.e., for phase v and
phase w. As a result, the phase-u voltage integration
value vsui is output from the integrator 1301, the phase-v
voltage integration value vsvi is output from the integrator
1302, and the phase-w voltage integration value vswi is5
output from the integrator 1303.
[0147] As the voltage integration value is a computed
integral of the instantaneous value of the stator voltage
vs of a rectangular waveform, the voltage integration value
vsxi is computed at a computation frequency set to a value10
sufficiently higher than the switching frequency fsw of the
gate signal generated by the PWM modulator 6. In order
that the computation frequency is sufficiently high, the
computation frequency is preferably 25 or more times the
switching frequency fsw, and more preferably 100 or more15
times the switching frequency fsw. In addition, a burden
of this integration operation itself is small. Even with a
high computation frequency, a computing device bears a
small computation burden as compared to the amount of
calculation necessary to control the rotary machine 2. The20
value of the supply voltage vdc for use in this operation
can be a detection value, a rated value, or an expected
value during use.
[0148] In the third embodiment, without using
integration, the flux linkage is computed to estimate the25
rotor position and the rotational speed. Here, a principle
for the position estimator 14 to estimate the rotor
position and the rotational speed will first be described.
A rotary machine model that formulates characteristics of
the rotary machine 2 in the rotational coordinate system is30
expressed by Equations (30) and (31) below.
[0149] Formula 30:
46
Formula 31:
[0150] Note that the symbol J in Equation (30) above is
the transformation matrix in Equation (15) above.5
[0151] In addition, since the inductance value is
obtained by calculation in the third embodiment, Equation
(31) above is expressed as Equation (32) below.
[0152] Formula 32:
10
[0153] In Equation (32) above, Lsd,calc represents a d-
axis inductance obtained by calculation, and Lsq,calc
represents a q-axis inductance obtained by calculation.
[0154] In addition, to calculate the induced voltage
ωrJψsdq that is the third term on the right-hand side of15
Equation (30) above, this induced voltage is denoted by
vemf,calc. In this respect, neglecting the derivative term
in Equation (30) above, i.e., the second term on the right-
hand side of Equation (30) above, the induced voltage
vemf,calc, which is a calculated value, can be computed using20
Equation (33) below from the stator voltage vsdq and the
stator current isdq.
[0155] Formula 33:
[0156] The voltage integration value vsxi computed by the25
voltage integrator 13 is used as the stator voltage vsdq of
Equation (33) above, and a detection value is used as the
stator current isdq of Equation (33). As the voltage
integration value vsxi is an integral of the stator voltage
vs of a rectangular waveform for the integration time Tgi,30
47
the voltage integration value vsxi is divided by Tgi to
thereby accurately obtain an average value of the stator
voltage vs over the integration time Tgi. The third
embodiment is based on the assumption that the integration
time Tgi is equal to the computation period Tpsi2 of the5
estimated rotor position θ^r. Note that the conventional
technology described in Patent Literature 1 uses the
voltage command value vsdq* as the stator voltage vsdq.
[0157] A flux linkage ψs,calcdq is computed using Equation
(32) above. Using Equation (34) below, an estimated10
induced voltage v^emf, which is an estimated value of an
induced voltage vemf, can be obtained from the thus computed
flux linkage ψs,calcdq and the estimated rotational angular
velocity ω^r.
[0158] Formula 34:15
[0159] The value computed by Equation (33) above and the
value estimated by Equation (34) above are compared with
each other. Proportional-integral control is performed in
such a manner that the difference between the values thus20
computed by Equations (33) and (34) converges to zero,
thereby obtaining the estimated rotational angular velocity
ω^r, i.e., the estimation of the rotational speed ωr, as
discussed above.
[0160] In addition, the induced voltage vemf,calc25
calculated using Equation (33) above is divided by the
estimated rotational angular velocity ω^r to thereby
provide a calculated value of the flux linkage ψs. This
calculated value is divided by the stator current is to
thereby provide a calculated value of the inductance value.30
[0161] As shown by Equation (6) above, the inductance
value varies depending on the true rotor position θr. In
48
addition, as shown by Equation (24) above, the inductance
value varies depending on the difference between the true
rotor position θr and the estimated rotor position θ^r.
Thus, checking the calculated value of the inductance value
against these inductance variation characteristics enables5
estimating the rotor position. Specifically, the flux
linkage ψsdq, which includes the flux linkage inductance
variable component generated by a product of the inductance
variable component Lsvar and the stator current is, is
divided by the stator current isdq to calculate the10
inductance value for obtaining the estimated rotor position
θ^r from the inductance variation characteristics dependent
on the rotor position.
[0162] A configuration and operation of the position
estimator 14 will next be described. FIG. 19 is a diagram15
illustrating an example configuration of the position
estimator 14 illustrated in FIG. 17. The position
estimator 14 can be configured to include a divider 1401,
three-phase to two-phase converters 1402 and 1404,
rotational coordinate converters 1403 and 1405, and a20
velocity-and-angle computing element 1406.
[0163] The divider 1401 divides the three-phase voltage
integration values vsui, vsvi, and vswi by the integration
time Tgi to thereby compute stator voltage average values
vsua, vsva, and vswa. The stator voltage average values vsua,25
vsva, and vswa are each an average value of the stator
voltage vs over the integration time Tgi. The three-phase
to two-phase converter 1402 converts the stator voltage
average values vsua, vsva, and vswa in the three-phase
coordinate system into stator voltage average values vsαa30
and vsβa in the two-phase coordinate system by three-phase
to two-phase conversion. The three-phase to two-phase
converter 1404 converts the stator currents isu, isv, and isw
49
in the three-phase coordinate system into the stator
currents isα and isβ in the two-phase coordinate system by
three-phase to two-phase conversion. The rotational
coordinate converter 1403 converts the stator voltage
average values vsαa and vsβa in the two-phase coordinate5
system into stator voltage average values vsda and vsqa in
the rotational coordinate system by rotational coordinate
conversion, using the estimated rotor position θ^r. The
rotational coordinate converter 1405 converts the stator
currents isα and isβ in the two-phase coordinate system into10
the stator currents isd and isq in the rotational coordinate
system by rotational coordinate conversion using the
estimated rotor position θ^r. Note that the estimated
rotor position θ^r, which is one of the outputs of the
velocity-and-angle computing element 1406, is fed back for15
use. The velocity-and-angle computing element 1406
computes the estimated rotor position θ^r and the estimated
rotational angular velocity ω^r in accordance with the
foregoing description.
[0164] The technique of the third embodiment allows a20
longer computation period of the flux linkage than those in
the case of the use of the observer and the incomplete
integration because the technique of the third embodiment
uses neither an observer nor incomplete integration in
computing the flux linkage. This longer computation period25
provides a lower computation burden, thereby making it
possible to use a computing device such as a less costly
microprocessor. The third embodiment is based on the
assumption that the control computation period Tpsi for
computing the estimated rotor position θ^r and the30
estimated rotational angular velocity ω^r is equal to the
integration time Tgi for computing the voltage integration
value vsxi.
50
[0165] In addition, as in the first and second
embodiments, the integration time Tgi for computing the
voltage integration value vsxi is not an integer multiple of
half of the switching period Tsw. Under this condition, a
conventional technology using a voltage command value vs*5
causes a mismatch between the voltage command value vs* and
the value obtained by smoothing the actual voltage. As a
result, the voltage command value vs* includes an error
relative to the actual voltage. An error therefore also
occurs in the estimated rotor position θ^r and the10
estimated rotational angular velocity ω^r computed using
these values.
[0166] In the rotary machine 2, an error in a lower
frequency will cause a larger oscillating component in the
magnetic flux and the torque. A near-DC component in a15
range from DC to a low frequency in a static coordinate
system appears as an error near the fundamental wave
frequency fs in the rotational coordinate system. In
contrast, the position estimator 14 of the third embodiment
uses the stator voltage integration value, thereby making20
it possible to accurately obtain an average stator voltage
over the integration interval thereof, i.e., over the
computation period of position estimation. This enables
estimating the position, removing the pulsation of torque
and power resulting from an error near the fundamental wave25
frequency fs.
[0167] As described above, the position estimator 14 of
the third embodiment, which is configured to compute the
flux linkage with a small computation burden thereon
without using an observer or incomplete integration, uses30
the voltage integration value vsxi to thereby estimate the
position, removing the pulsation of torque and power
resulting from an error near the fundamental wave frequency
51
fs. Use of the technique of the third embodiment
accordingly provides an unprecedented outstanding advantage
that the control device 100B including no position sensor
and providing a low level of torque pulsation and a low
level of power pulsation can be made without the need for a5
costly microprocessor.
[0168] A configuration of hardware in the control
devices 100, 100A, and 100B according to the first through
third embodiments described above will next be described
with reference to FIGs. 20 and 21. FIG. 20 is a diagram10
illustrating a first example hardware configuration for
implementing each functionality of the control devices 100,
100A, and 100B according to the first through third
embodiments. FIG. 21 is a diagram illustrating a second
example configuration for implementing each functionality15
of the control devices 100, 100A, and 100B according to the
first through third embodiments. Note that the term each
functionality of the control devices 100, 100A, and 100B
refers to the functionality of each of the controller 5,
the PWM modulators 6 and 9, the voltage integrators 7, 10,20
and 13, and the position estimators 8, 11, and 14 included
in the control devices 100, 100A, and 100B.
[0169] Each functionality of the controller 5, the PWM
modulators 6 and 9, the voltage integrators 7, 10, and 13,
and the position estimators 8, 11, and 14 can be25
implemented using a processing circuitry. FIG. 20
illustrates a dedicated processing circuitry 15, which
replaces the controller 5, the PWM modulators 6 and 9, the
voltage integrators 7, 10, and 13, and the position
estimators 8, 11, and 14 in the first through third30
embodiments. When a dedicated hardware element is used,
the dedicated processing circuitry 15 is a single circuit,
a set of multiple circuits, an application specific
52
integrated circuit (ASIC), a field-programmable gate array
(FPGA), or a combination thereof. The functionalities of
the controller 5, the PWM modulators 6 and 9, the voltage
integrators 7, 10, and 13, and the position estimators 8,
11, and 14 may each be implemented in a processing5
circuitry, or may together be implemented in a processing
circuitry.
[0170] In addition, FIG. 21 illustrates a processor 16
and a storage device 17, which together replace the
controller 5, the PWM modulators 6 and 9, the voltage10
integrators 7, 10, and 13, and the position estimators 8,
11, and 14 in the configurations of the first through third
embodiments. The processor 16 may be computing means such
as a computing unit, a microprocessor, a microcomputer, a
central processing unit (CPU), or a digital signal15
processor (DSP). In addition, examples of the storage
device 17 include non-volatile and volatile semiconductor
memories such as a random access memory (RAM), a read-only
memory (ROM), a flash memory, an erasable programmable ROM
(EPROM), and an electrically erasable programmable ROM20
(EEPROM) (registered trademark).
[0171] When the processor 16 and the storage device 17
are used, each functionality of the controller 5, the PWM
modulators 6 and 9, the voltage integrators 7, 10, and 13,
and the position estimators 8, 11, and 14 is implemented in25
software, firmware, or a combination thereof. The software
or firmware is described in the form of programs, and is
stored in the storage device 17. The processor 16 reads
and executes a program stored in the storage device 17. In
addition, it can also be said that these programs cause a30
computer to perform a procedure and a method of each
functionality of the controller 5, the PWM modulators 6 and
9, the voltage integrators 7, 10, and 13, and the position
53
estimators 8, 11, and 14. The storage device 17 may be,
for example, a non-volatile or volatile semiconductor
memory such as a ROM, an EPROM, or an EEPROM; a flexible
disk, an optical disk, a compact disc, a digital versatile
disc (DVD), or the like.5
[0172] Each functionality of the controller 5, the PWM
modulators 6 and 9, the voltage integrators 7, 10, and 13,
and the position estimators 8, 11, and 14 may be
implemented partially in hardware, and partially in
software or firmware. For example, an implementation may10
be used in which the functionality of the PWM modulators 6
and 9 and the voltage integrators 7, 10, and 13 is
implemented in a dedicated hardware element, and the
functionality of the controller 5 and the position
estimators 8, 11, and 14 is implemented in a combination of15
the processor 16 and the storage device 17.
[0173] Note that the second and third embodiments of the
present description has been described in which the rotary
machine 2 is a synchronous reluctance motor by way of
example. However, the rotary machine 2 may be an induction20
motor or a permanent magnet motor. When the rotary machine
2 is an induction motor, the technique disclosed in, for
example, Japanese Patent Application Laid-open No. H11-4599
can be used. Alternatively, when the rotary machine 2 is a
permanent magnet motor, the technique disclosed in, for25
example, WO 2002/091558 A can be used. Note that the
technique in the third embodiment uses, in part, the
technique described in Japanese Patent Application Laid-
open No. 2002-165475. Thus, refer to this patent literature
for unmentioned details with respect to the third30
embodiment.
[0174] In addition, the present description has
described the voltage applicator 3 in the context of a
54
three-phase two-level inverter, but the voltage applicator
3 is not limited thereto. The voltage applicator 3 may be
an inverter having another number of phases, or may be a
multi-level inverter such as a three-level inverter or a
five-level inverter. Use of any of these inverters can5
also provide the rotary machine control device according to
the present disclosure.
[0175] Moreover, the present description has described
that the switching frequency ranges from one to 15 times
the fundamental wave frequency fs as an example of10
switching frequency. In general, when, for example, a
carrier signal common to three phases is used, a switching
frequency that is a multiple-of-three times (e.g., three
times, six times, nine times, 12 times, and 15 times) the
fundamental wave frequency fs is used in addition to a15
switching frequency equal to the fundamental wave frequency
fs. Meanwhile, when a fixed switching pattern is used
without using a carrier signal, any integer multiple may be
used.
[0176] Furthermore, the present description has20
described that the stator current for the torque of the
rotary machine 2 is set to minimize the current RMS value,
but the stator current is not limited thereto. The stator
current for the torque of the rotary machine 2 may be set
to minimize the flux linkage or to maximize the efficiency25
of the voltage applicator 3 or of the rotary machine 2.
[0177] The configurations described in the foregoing
embodiments are merely examples. These configurations may
be combined with a known other technology, and
configurations of different embodiments may be combined30
together. Moreover, part of such configurations may be
omitted and/or modified without departing from the spirit
thereof.
55
Reference Signs List
[0178] 1 DC power supply; 2 rotary machine; 2a
stator; 2b rotor; 3 voltage applicator; 4 current
detector; 5 controller; 6, 9 PWM modulator; 7, 10, 135
voltage integrator; 8, 11, 14 position estimator; 12
voltage detector; 15 dedicated processing circuitry; 16
processor; 17 storage device; 31-36 switching element;
100, 100A, 100B control device; 501 current command value
computing element; 502, 801, 803, 1101, 1102, 1402, 140410
three-phase to two-phase converter; 503, 802, 804, 1103,
1403, 1405 rotational coordinate converter; 504 d-q
current controller; 505 rotational coordinate reverse
converter; 506 two-phase to three-phase converter; 701,
703, 705, 1002, 1005, 1008, 1301, 1302, 1303 integrator;15
702, 704, 706, 1003, 1006, 1009 multiplier; 805 observer;
1001, 1004, 1007 LPF; 1104 first computing element; 1105
first estimator; 1106 second computing element; 1107
third computing element; 1401 divider; 1406 velocity-and-
angle computing element.20
56
We Claim:
[Claim 1] A rotary machine control device comprising:
a voltage applicator connected between a direct
current power supply and a rotary machine to apply a stator
voltage of a rectangular waveform to the rotary machine by5
switching on and off a plurality of switching elements
provided for each phase;
a current detector to detect a stator current flowing
between the voltage applicator and a stator winding of the
rotary machine;10
a controller to compute a voltage command value on the
basis of the stator current and a rotor position, the
voltage command value being a command value of the stator
voltage, the stator voltage being a voltage applied to the
stator winding, the rotor position being information on a15
position of a rotor of the rotary machine;
a pulse width modulator to generate a gate signal such
that a value obtained by smoothing the stator voltage
matches the voltage command value, and to use the gate
signal to control turning on and off each of the switching20
elements;
a voltage integrator to compute a voltage integration
value by integrating the gate signal, the voltage
integration value being an integrated value of the stator
voltage; and25
a position estimator to estimate the rotor position on
the basis of the voltage integration value and the stator
current.
[Claim 2] The rotary machine control device according to30
claim 1, comprising:
a voltage detector to detect the stator voltage,
wherein
57
the voltage integrator computes the voltage
integration value by integrating a detection value of the
stator voltage detected by the voltage detector, instead of
integrating the gate signal.
5
[Claim 3] The rotary machine control device according to
claim 1 or 2, wherein
an estimated value of the rotor position is computed
on the basis of a flux linkage of the rotary machine, and
the flux linkage is computed using the voltage10
integration value.
[Claim 4] The rotary machine control device according to
claim 3, wherein
a computation period for which the flux linkage is15
computed is not an integer multiple of half of a switching
period, the switching period being a reciprocal of a
switching frequency of switching on and off each of the
switching elements.
20
[Claim 5] The rotary machine control device according to
any one of claims 1 to 4, wherein
an estimated value of the rotor position is computed
on the basis of a flux linkage of the rotary machine, and
the flux linkage is computed using the voltage25
integration value in a static coordinate system.
[Claim 6] The rotary machine control device according to
any one of claims 1 to 5, wherein
the pulse width modulator synchronizes a switching30
frequency of switching on and off each of the switching
elements with an integer multiple of a fundamental wave
frequency of a rotational speed of the rotary machine.
58
[Claim 7] The rotary machine control device according to
any one of claims 1 to 6, wherein
the pulse width modulator switches a switching
frequency of switching on and off each of the switching5
elements, depending on a fundamental wave frequency of a
rotational speed of the rotary machine or depending on an
amplitude of the voltage command value.
[Claim 8] The rotary machine control device according to10
any one of claims 1 to 7, wherein
a computation period for which the rotor position is
estimated is not an integer multiple of half of a switching
period, the switching period being a reciprocal of a
switching frequency of switching on and off each of the15
switching elements.
[Claim 9] The rotary machine control device according to
claim 8, wherein
a computation frequency is 25 or more times the20
switching frequency, the computation frequency being a
frequency at which the voltage integration value is
computed.
[Claim 10] The rotary machine control device according25
to any one of claims 1 to 9, wherein
the voltage integration value is computed after the
gate signal or a detection value of the stator voltage
passes through a low-pass filter.
30
[Claim 11] The rotary machine control device according
to any one of claims 1 to 10, wherein
the rotary machine has inductance including an
59
inductance variable component that varies with the rotor
position, and
the position estimator estimates the rotor position on
the basis of a flux linkage inductance variable component
generated by a product of the inductance variable component5
and the stator current.
[Claim 12] The rotary machine control device according
to claim 11, wherein
the inductance of the rotary machine includes an10
averaged component and a variable component, the averaged
component being invariable regardless of the rotor position,
the variable component varying at a frequency twice an
electrical angle frequency of the rotor position, and
the flux linkage inductance variable component is15
generated by a product of the variable component and the
stator current.