DESCRIPTION [TITLE OF INVENTION] SIGNAL GENERATION METHOD AND SIGNAL GENERATION DEVICE 5 [Technical Field] [OOO 11 (CROSS-REFERENCE TO RELATED APPLICATIONS) This application is based on application No. 2010-276447 filed December 10,2010 in Japan, the contents of which are hereby incorporated by reference. 10 The present invention relates to a signal generation method and a signal generation apparatus for communication using multiple antennas. [Background Art] [0002] 15 A MIMO (Multiple-Input, Multiple-Output) system is an example of a conventional communication system using multiple antennas. In multi-antenna communication, of which the MIMO system is typical, multiple transmission signals are each modulated, and each modulated signal is simultaneously transmitted from a different antenna in order to increase the transmission speed of the data. 20 [0003] Fig. 23 illustrates a sample configuration of a transmission and reception device having two transmit antennas and two receive antennas, and using two transmit modulated signals (transmit streams). In the transmission device, encoded data are interleaved, the interleaved data are modulated, and frequency conversion 25 and the like are performed to generate transmission signals, which are then transmitted from antennas. In this case, the scheme for simultaneously transmitting different modulated signals fiom different transmit antennas at the same time and on a common frequency is a spatial multiplexing MIMO system. 1 [0004] In this context, Patent Literature 1 suggests using a transmission device provided with a different interleaving pattern for each transmit antenna. That is, the transmission device from Fig. 23 should use two distinct interleaving patterns 5 performed by two interleavers (n, and nb). AS for the reception device, Non-Patent Literature 1 and Non-Patent Literature 2 describe improving reception quality by iteratively using soft values for the detection scheme (by the MIMO detector of Fig. 10 As it happens, models of actual propagation environments in wireless communications include NLOS (Non Line-Of-Sight), typified by a Rayleigh fading environment is representative, and LOS (Line-Of-Sight), typified by a Rician fading environment. When the transmission device transmits a single modulated signal, and the reception device performs maximal ratio combination on the signals 15 received by a plurality of antennas and then demodulates and decodes the resulting signals, excellent reception quality can be achieved in a LOS environment, in particular in an environment where the Rician factor is large. The Rician factor represents the received power of direct waves relative to the received power of scattered waves. However, depending on the transmission system (e.g., a spatial . 20 multiplexing MIMO system), a problem occurs in that the reception quality deteriorates as the Rician factor increases (see Non-Patent Literature 3). Figs. 24A and 24B illustrate an example of simulation results of the BER (Bit Error Rate) characteristics (vertical axis: BER, horizontal axis: SNR (signal-to-noise ratio) for data encoded with LDPC (low-density parity-check) codes 25 and transmitted over a 2 x 2 (two transmit antennas, two receive antennas) spatial multiplexing MIMO system in a Rayleigh fading environment and in a Rician fading environment with Rician factors of K = 3, 10, and 16 dB. Fig. 24A gives the Max-Log approximation-based log-likelihood ratio (Max-log APP) BER 2 characteristics without iterative detection (see Non-Patent Literature 1 and Non-Patent Literature 2), while Fig. 24B gives the Max-log APP BER characteristic with iterative detection (see Non-Patent Literature 1 and Non-Patent Literature 2) (number of iterations: five). Figs. 24A and 24B clearly indicate that, regardless of 5 whether or not iterative detection is performed, reception quality degrades in the spatial multiplexing MIMO system as the Rician factor increases. Thus, the problem of reception quality degradation upon stabilization of the propagation environment in the spatial multiplexing MIMO system, which does not occur in a conventional single-modulation signal system, is unique to the spatial multiplexing 10 MIMO system. [0006] Broadcast or multicast communication is a service applied to various propagation environments. The radio wave propagation environment between the broadcaster and the receivers belonging to the users is often a LOS environment. 15 When using a spatial multiplexing MIMO system having the above problem for broadcast or multicast communication, a situation may occur in which the received electric field strength is high at the reception device, but in which degradation in reception quality makes service reception difficult. In other words, in order to use a spatial multiplexing MIMO system in broadcast or multicast communication in 20 both the NLOS environment and the LOS environment, a MIMO system that offers a certain degree of reception quality is desirable. [0007] Non-Patent Literature 8 describes a scheme for selecting a codebook used in precoding (i.e. a precoding matrix, also referred to as a precoding weight matrix) 25 based on feedback information fiom a communication party. However, Non-Patent Literature 8 does not at all disclose a scheme for precoding in an environment in which feedback information cannot be acquired fiom the other party, such as in the above broadcast or multicast communication. 3 [0008] On the other hand, Non-Patent Literature 4 discloses a scheme for switching the precoding matrix over time. This scheme is applicable when no feedback information is available. Non-Patent Literature 4 discloses using a unitary matrix 5 as the precoding matrix, and switching the unitary matrix at random, but does not at all disclose a scheme applicable to degradation of reception quality in the above-described LOS environment. Non-Patent Literature 4 simply recites hopping between precoding matrices at random. Obviously, Non-Patent Literature 4 makes no mention whatsoever of a precoding method, or a structure of a precoding 10 matrix, for remedying degradation of reception quality in a LOS environment. [Citation List] patent Literature] [0009] [Patent Literature 11 15 International Patent Application Publication No. W020051050885 [Non-Patent Literature] [OO lo] won-Patent Literature 11 "Achieving near-capacity on a multiple-antenna channel" IEEE Transaction on communic@ions, vo1.51, no.3, pp.389-399, March 2003 [Non-Patent Literature 21 "Performance analysis and design optimization of LDPC-coded MIMO OFDM systems" IEEE Trans. Signal Processing, vo1.52, 110.2, pp.348-361, Feb. 2004 Don-Patent Literature 31 "BER performance evaluation in 2x2 MIMO spatial multiplexing systems under Rician fading channels" IEICE Trans. Fundamentals, vol.E9 1 -A, no. 10, pp.2798-2807, Oct. 2008 4 won-Patent Literature 41 "Turbo space-time codes with time varying linear transformations" IEEE Trans. Wireless communications, ~01.6, 110.2, pp.486-493, Feb. 2007 Won-Patent Literature 51 5 "Likelihood function for QR-MLD suitable for soft-decision turbo decoding and its performance" IEICE Trans. Commun., vol.E88-B, no.1, pp.47-57, Jan. 2004 won-Patent Literature 61 "A tutorial on 'Parallel concatenated (Turbo) coding', 'Turbo (iterative) decoding' and related topics" IEICE, Technical Report IT98-5 1 10 won-Patent Literature 71 "Advanced signal processing for PLCs: Wavelet-OFDM" Proc. of IEEE International symposium on ISPLC 2008, pp. 187-192,2008 won-Patent Literature 81 D. J. Love and R W. Heath Jr., "Limited feedback unitary precoding for 15 spatial multiplexing systems" IEEE Trans. Inf. Theory, vo1.5 1, no.8, pp.2967-2976, Aug. 2005 won-Patent Literature 91 DVB Document A122, Framing structure, channel coding and modulation for a second generation digital terrestrial television broadcasting system (DVB-T2), 20 June 2008 won-Patent Literature 101 L. Vangelista, N. Benvenuto, and S. Tomasin "Key technologies for next-generation terrestrial digital television standard DVB-T2," IEEE Commun. Magazine, vo1.47, no.10, pp.146-153, Oct. 2009 25 won-Patent Literature 1 11 T. Ohgane, T. Nishimura, and Y. Ogawa, "Application of space division multiplexing and those performance in a MIMO channel" IEICE Trans. Commun., vol.E88-B, no.5, pp. 1843-185 1, May 2005 5 won-Patent Literature 123 R G. Gallager "Low-density parity-check codes," IRE Trans. Inform. Theory, IT-8, pp.21-28, 1962 won-Patent Literature 131 D. J. C. Mackay, "Good error-correcting codes based on very sparse matrices," IEEE Trans. Inform. Theory, vo1.45,no.2, pp.399-43 1, March 1999. won-Patent Literature 141 ETSI EN 302 307, "Second generation framing structure, channel coding and modulation systems for broadcasting, interactive services, news gathering and other broadband satellite applications" v. 1.1.2, June 2006 won-Patent Literature 151 Y.-L. Ueng, and C.-C. Cheng "A fast-convergence decoding method and memory-efficient VLSI decoder architecture for irregular LDPC codes in the IEEE 802.16e standards" IEEE VTC-2007 Fall, pp. 1255-1259 [Non-Patent Literature 161 S. M. Alarnouti "A simple transmit diversity technique for wireless communications" IEEE J. Select. Areas Commun., vol. 16,no.8, pp. 145 1 - 1458, Oct 1998 won-Patent Literature 171 V. Tarokh, H. Jafrkhani, and A. R Calderbank "Space-time block coding for wireless communications: Performance results" IEEE J. Select. Areas Commun., vo1.17, no.3, no.3, pp.451-460, March 1999 [Summary of Invention] [Technical Problem] [OOll] An object of the present invention is to provide a MIMO system that improves reception quality in a LOS environment. [Solution to Problem] 6 [OO 121 The present invention provides a signal generation scheme for generating, fi-om a plurality of baseband signals, a plurality of signals for transmission on a common fiequency band and at a common time, comprising the steps of: performing 5 a change of phase on each of a first baseband signal sl generated fiom a first set of bits and a second baseband signal s2 generated fiom a second set of bits, thus generating a first post-phase change baseband signal sl' and a second post-phase change baseband signal s2'; and applying weighting to the first post-phase change baseband signal slr and to the second post-phase change baseband signal s2' 10 according to a predetermined matrix F, thus generating a first weighted signal zl and a second weighted signal 22 as the plurality of signals for transmission on the common fiequency band and at the common time, wherein the first weighted signal zl and the second weighted signal 22 satisfy the relation: (21, z21T = F(slt, ~ 2 'a)nd~ the change of phase is performed on the first baseband signal sl and the second 15 baseband signal s2 using a phase modification value sequentially selected from among N phase modification value candidates, N being an integer equal to or greater than two and each of the N phase modification value candidates being selected at least once within a predetermined period. [00 131 20 Also, the present invention provides a signal generation apparatus for generating, fiom a plurality of baseband signals, a plurality of signals for transmission on a common fi-equency band and at a common time, comprising: a phase changer performing a change of phase on each of a first baseband signal sl generated fiom a first set of bits and a second baseband signal s2 generated fi-om a 25 second set of bits, thus generating a first post-phase change baseband signal sl' and a second post-phase change baseband signal s2'; and a weighting unit applying weighting to the first post-phase change baseband signal sl' and to the second post-phase change baseband signal s2' according to a predetermined matrix F, thus 7 generating a first weighted signal zl and a second weighted signal 22 as the plurality of signals for transmission on the common frequency band and at the common time, wherein the first weighted signal zl and the second weighted signal 22 satisfy the relation: (zl, 22)' = F(sl', s2'lT and the change of phase is performed on the first 5 baseband signal sl and the second baseband signal s2 using a phase modification value sequentially selected from among N phase modification value candidates, N being an integer equal to or greater than two and each of the N phase modification value candidates being selected at least once within a predetermined period. [00 141 10 [Advantageous Effects of Invention] [00 151 According to the above structure, the present invention provides a signal generation scheme and signal generation apparatus that remedy degradation of reception quality in a LOS environment, thereby providing high-quality service to 15 LOS users during broadcast or multicast communication. [Brief Description of Drawings] [00 1 61 Fig. 1 illustrates an example of a transmission and reception device in a spatial multiplexing MIMO system. 20 Fig. 2 illustrates a sample frame configuration. Fig. 3 illustrates an example of a transmission device applying a phase changing scheme. Fig. 4 illustrates another example of a transmission device applying a phase changing scheme. 25 Fig. 5 illustrates another sample frame configuration. Fig. 6 illustrates a sample phase changing scheme. Fig. 7 illustrates a sample configuration of a reception device. Fig. 8 illustrates a sample configuration of a signal processor in the reception device. Fig. 9 illustrates another sample configuration of a signal processor in the reception device. 5 Fig. 10 illustrates an iterative decoding scheme. Fig. 1 1 illustrates sample reception conditions. Fig. 12 illustrates a further example of a transmission device applying a phase changing scheme. Fig. 13 illustrates yet a further example of a transmission device applying a 10 phase changing scheme. Fig. 14 illustrates a further sample frame configuration. Fig. 15 illustrates yet another sample fiame configuration. Fig. 16 illustrates still another sample frame configuration. Fig. 17 illustrates still yet another sample £kame configuration. Fig. 18 illustrates yet a further sample fiame configuration. Figs. 19A and 19B illustrate examples of a mapping scheme. Figs. 20A and 20B illustrate further examples of a mapping scheme. Fig. 2 1 illustrates a sample configuration of a weighting unit. Fig. 22 illustrates a sample symbol rearrangement scheme. 20 Fig. 23 illustrates another example of a transmission and reception device in a spatial multiplexing MZMO system. Figs. 24A and 24B illustrate sample BER characteristics. Fig. 25 illustrates another sample phase changing scheme. Fig. 26 illustrates yet another sample phase changing scheme. Fig. 27 illustrates a further sample phase changing scheme. Fig. 28 illustrates still a further sample phase changing scheme. Fig. 29 illustrates still yet a further sample phase changing scheme. Fig. 30 illustrates a sample symbol arrangement for a modulated signal providing high received signal quality. Fig. 31 illustrates a sample fi-ame configuration for a modulated signal providing high received signal quality. 5 Fig. 32 illustrates another sample symbol arrangement for a modulated signal providing high received signal quality. Fig. 33 illustrates yet another sample symbol arrangement for a modulated signal providing high received signal quality. Fig. 34 illustrates variation in numbers of symbols and slots needed per 10 coded block when block codes are used. Fig. 35 illustrates variation in numbers of symbols and slots needed per pair of coded blocks when block codes are used. Fig. 36 illustrates an overall configuration of a digital broadcasting system. Fig. 37 is a block diagram illustrating a sample receiver. Fig. 38 illustrates multiplexed data configuration. Fig. 39 is a schematic diagram illustrating multiplexing of encoded data into streams. Fig. 40 is a detailed diagram illustrating a video stream as contained in a PES packet sequence. 20 Fig. 41 is a structural diagram of TS packets and source packets in the multiplexed data. Fig. 42 illustrates PMT data configuration. Fig. 43 illustrates information as configured in the multiplexed data. Fig. 44 illustrates the configuration of stream attribute information. Fig. 45 illustrates the configuration of a video display and audio output device. Fig. 46 illustrates a sample configuration of a communications system. Figs. 47A and 47B illustrate a variant sample symbol arrangement for a modulated signal providing high received signal quality. Figs. 48A and 48B illustrate another variant sample symbol arrangement for a modulated signal providing high received signal quality. Figs. 49A and 49B illustrate yet another variant sample symbol arrangement for a modulated signal providing high received signal quality. Figs. 50A and 50B illustrate a further variant sample symbol arrangement for a modulated signal providing high received signal quality. Fig. 5 1 illustrates a sample configuration of a transmission device. Fig. 52 illustrates another sample configuration of a transmission device. Fig. 53 illustrates a further sample configuration of a transmission device. Fig. 54 illustrates yet a fixther sample configuration of a transmission device. Fig. 55 illustrates a baseband signal switcher. (Description of Embodiments] [OO 171 Embodiments of the present invention are described below with reference to the accompanying drawings. [Embodiment 11 The following describes, in detail, a transmission scheme, a transmission device, a reception scheme, and a reception device pertaining to the present Embodiment. [OO 1 81 Before beginning the description proper, an outline of transmission schemes and decoding schemes in a conventional spatial multiplexing MlMO system is provided. Fig. 1 illustrates the structure of an NpN, spatial multiplexing MIMO system. An information vector z is encoded and interleaved. The encoded bit vector u = (ul, . . . ~~3 is obtained as the interleave output. Here, uj = (uil, . . . ulu> 11 (where M is the number of transmitted bits per symbol). For a transmit vector s = (sly ... SNJ, a received signal si = map(ui) is found for transmit antenna #i. Normalizing the transmit energy, this is expressible as ~ { l s ~=f E) a t( where E, is the total energy per channel). The receive vector y = (y,, . . . yNJT is expressed in 5 Math. 1 (formula l), below. [00 191 [Math. 11 (formula 1) 10 [0020] Here, HNtNirs the channel matrix, n = (nl, .. . nW)i s the noise vector, and the average value of ni is zero for independent and identically distributed (i.i.d) complex Gaussian noise of variance 02. Based on the relationship between transmitted symbols introduced into a receiver and the received symbols, the probability 15 distribution of the received vectors can be expressed as Math. 2 (formula 2), below, for a multi-dimensional Gaussian distribution. [002 11 [Math. 21 (formula 2) [0022] Here, a receiver performing iterative decoding is considered. Such a receiver is illustrated in Fig. 1 as being made up of an outer soft-inlsoft-out decoder and a MIMO detector. The log-likelihood ratio vector (L-value) for Fig. 1 is given by Math. 3 (formula 3) through Math. 5 (formula 5), as follows. [0023] [Math. 31 5 (formula 3) [0024] [Math. 41 (formula 4) [0025] [Math. 51 (formula 5) 15 [0026] (Iterative Detection Scheme) The following describes the MIMO signal iterative detection performed by the NpN, spatial multiplexing MIMO system. The log-likelihood ratio of u, is defined by Math. 6 (formula 6). 20 100271 wath. 61 (formula 6) Through application of Bayes' theorem, Math. 6 (formula 6) can be expressed as Math. 7 (formula 7). (formula 7) [0030] 10 Note that U, ., = { u b = *I). Through the approximation lnCaj - max In aj, Math. 7 (formula 7) can be approximated as Math. 8 (formula 8). The symbol - is herein used to signify approximation. [003 11 [Math. 81 15 (formula 8) In Math. 8 (formula 8), P(ulu,) and In P(ulu,) can be expressed as follows. [Math. 91 (formula 9) [0034] [Math. lo] (formula 10) 10 [0035] [Math. 111 (formula 11) Note that the log-probability of the equation given in Math. 2 (formula 2) can be expressed as Math. 12 (formula 12). 100371 [Math. 121 5 (formula 12) [003 81 Accordingly, given Math. 7 (formula 7) and Math. 13 (formula 13), the posterior L-value for the MAP or APP (a posteriori probability) can be can be 10 expressed as follows. . [0039] [Math. 131 (formula 13) 15 [0040] This is hereinafter termed iterative APP decoding. Also, given Math. 8 (formula 8) and Math. 12 (formula 12), the posterior L-value for the Max-log APP can be can be expressed as follows. . [0041] 20 [Math. 141 (formula 14) [0042] [Math. 151 (formula 15) 5 [0043] This is hereinafter referred to as iterative Max-log APP decoding. As such, the external information required by the iterative decoding system is obtainable by subtracting prior input from Math. 13 (formula 13) or from Math. 14 (formula 14). (System Model) 10 Fig. 23 illustrates the basic configuration of a system related to the following explanations. The illustrated system is a 2x2 spatial multiplexing MIMO system having an outer decoder for each of two streams A and B. The two outer decoders perform identical LDPC encoding (Although the present example considers a configuration in which the outer encoders use LDPC codes, the outer 15 encoders are not restricted to the use of LDPC as the error-correcting codes. The example may also be reaIized using other error-correcting codes, such as turbo codes, convolutional codes, or LDPC convolutional codes. Further, while the outer encoders are presently described as individually configured for each transmit antenna, no limitation is intended in this regard. A single outer encoder may be 20 used for a plurality of transmit antennas, or the number of outer encoders may be greater than the number of transmit antennas. The system also has interleavers (za, zb) for each of the streams A and B. Here, the modulation scheme is 2 h -(i.e~., ~ ~ h bits transmitted per symbol). [0044] 25 The receiver performs iterative detection (iterative APP (or Max-log APP) decoding) of MTMO signals, as described above. The LDPC codes are decoded using, for example, sum-product decoding. 17 [0045] Fig. 2 illustrates the fiarne configuration and describes the symbol order after interleaving. Here, (iaj,) and (ibjb)c an be expressed as follows. [OM61 5 [Math. 161 (formula 16) [0047] [Math. 171 10 (formula 17) (i,, j ,) = zb( a;, jb) [0048] Here, i, and ib represent the symbol order after interleaving, ja and jb represent the bit position in the modulation scheme (where j,jb = 1, . . . h), 7ca and nb 15 represent the interleavers of streams A and B, and Claw and nbajbrepresent the data order of streams A and B before interleaving. Note that Fig. 2 illustrates a situation where i, = ib. (Iterative Decoding) The following describes, in detail, the sum-product decoding used in 20 decoding the LDPC codes and the MIMO signal iterative detection algorithm, both used by the receiver. [0049] Sum-Product Decoding A two-dimensional MxN matrix H = (Hm) is used as the check matrix for 25 LDPC codes subject to decoding. For the set [1,N] = (1, 2 ... N), the partial sets A(m) and B(n) are defined as follows. 18 [0050] [Math. 181 (formula 18) 5 [0051] [Math. 191 (formula 19) [0052] 10 Here, A(m) signifies the set of column indices equal to 1 for row m of check matrix H, while B(n) signifies the set of row indices equal to 1 for row n of check matrix H. The sum-product decoding algorithm is as follows. Step A- 1 (Initialization): For all pairs (qn) satisfying H,, = 1, set the prior log ratio P, = 1. Set the loop variable (number of iterations) 1,- = 1, and set the 15 maximum number of loops Is-. Step A-2 (Processing): For all pairs (an) satisfjmg H,, = 1 in the order m = 1, 2, . . . M , update the extrinsic value log ratio a- using the following update formula. [0053] [Math. 201 20 (formula 20) [0054] [Math. 2 11 (formula 2 1) [Math. 221 (formula 22) exp(x) + 1 f (x) = In 5 exp(x) - 1 [0056] where f is the Gallager function. h, can then be computed as follows. Step A-3 (Column Operations): For all pairs (m,n) satisfying H, = 1 in the order n = 1, 2, ... N , update the extrinsic value log ratio p, using the following update 10 formula. [0057] wath. 231 (formula 23) Step A-4 (Log-likelihood Ratio Calculation): For n€[l,Nl, the log-likelihood ratio L, is computed as follows. 20 (formula 24) Step A-5 (Iteration Count): If I,, < I-, then 1- is incremented and the process returns to step A-2. Sum-product decoding ends when I, = 1- [006 1 1 The above describes one iteration of sum-product decoding operations. 5 Afterward, MUlIO signal iterative detection is performed. The variables m, n, h, $,, A,,, and L, used in the above explanation of sum-product decoding operations are expressed as q, n, aam, pam, L, and for stream A and as mb, nb, abmbb, pbmhb, Lb, and for stream B. (MIMO Signal Iterative Detection) 10 The following describes the calculation of h, for MIMO signal iterative detection. [0062] The following formula is derivable fiom Math. 1 (formula 1). COO631 15 [Math. 251 (formula 25) [0064] Given the fiame configuration illustrated in Fig. 2, the following functions 20 are derivable fiom Math. 16 (formula 16) and Math. 17 (formula 17). [0065] Math. 261 (formula 26) [0066] Math. 271 (formula 27) 5 [0067] where na,nb E [I ,Nl. For iteration k of MIMO signal iterative detection, the variables &, L,,,, ?L,,~, and L,b are expressed as h-, L-, L b , and Lbb. [0068] Step B-1 (Initial Detection; k = 0) 10 For initial wave detection, and bpbar e calculated as follows. For iterative APP decoding: [0069] [Math. 281 (formula 28) [0070] For iterative Max-log APP decoding: [007 11 [Math. 291 (formula 29) [Math. 301 (formula 30) [0073] 5 where X = a,b. Next, the iteration count for the MIMO signal iterative detection is set to lmim=O 0 , with the maximum iteration count being I&,. Step B-2 (Iterative Detection; Iteration k): When the iteration count is k, Math. 1 1 (formula 1 I), Math. 13 (formula 13) through Math. 15 (formula 1 S), Math. 16 (formula 16), and Math. 17 (formula 17) can be expressed as Math. 31 (formula 10 3 1) through Math. 34 (formula 34), below. Note that (X,Y) = (a,b)(b,a). For iterative APP decoding: [0074] math. 3 11 (formula 3 1) [0075] [Math. 321 (formula 32) [0076] For iterative Max-log APP decoding: [0077] [Math. 331 5 (formula 33) [0078] [Math. 341 (formula 34) [0079] Step B-3 (Iteration Count and Codeword Estimation) If lmim 1) or at a predetermined interval). The details of the phase changing pattern are explained below, in Embodiment 4. 25 [OlOl] Wireless unit 310B takes post-phase change signal 309B as input and performs processing such as quadrature modulation, band limitation, frequency 29 conversion, amplification, and so on, then outputs transmit signal 31 1B. Transmit signal 3 11B is then output as radio waves by an antenna 3 12B. [O 1 021 Fig. 4 illustrates a sample configuration of a transmission device 400 that 5 differs fiom that of Fig. 3. The points of difference of Fig. 4 from Fig. 3 are described next. An encoder 402 takes information (data) 401 and the frame configuration signal 3 13 as input, and, in accordance with the fiame configuration signal 3 13, performs error-correction coding and outputs encoded data 402. 10 [0103] A distributor 404 takes the encoded data 403 as input, performs distribution thereof, and outputs data 405A and data 405B. Although Fig. 4 illustrates only one encoder, the number of encoders is not limited as such. The present invention may also be realized using m encoders (m being an integer, rn > 1) such that the 15 distributor divides the encoded data created by each encoder into two groups for distribution. [0 1041 Fig. 5 illustrates an example of a frame configuration in the time domain for a transmission device according to the present Embodiment. Symbol 500-1 is for 20 notifying the reception device of the transmission scheme. For example, symbol 500-1 conveys information such as the error-correction scheme used for transmitting data symbols, the coding rate thereof, and the modulation scheme used for transmitting data symbols. [0 1051 Symbol 501-1 is for estimating channel fluctuations for modulated signal zl(t) (where t is time) transmitted by the transmission device. Symbol 502-1 is a data symbol transmitted by modulated signal zl(t) as symbol number u (in the time domain). Symbol 503-1 is a data symbol transmitted by modulated signal zl(t) as symbol number u+ 1. [0 1 061 Symbol 501-2 is for estimating channel fluctuations for modulated signal 5 220) (where t is time) transmitted by the transmission device. Symbol 502-2 is a data symbol transmitted by modulated signal d(t) as symbol number u (in the time domain). Symbol 5032 is a data symbol transmitted by modulated signal zl(t) as symbol number u+ 1. [0107] 10 Here, the symbols of zl(t) and of z2(t) having the same time (identical timing) are transmitted from the transmit antenna using the same (sharedcommon) frequency . [OlOS] The following describes the relationships between the modulated signals 15 zl(t) and d(t) transmitted by the transmission device and the received signals rl(t) and r2(t) received by the reception device. In Fig. 5, 504#1 and 504#2 indicate transmit antennas of the transmission device, while 505#1 and 505#2 indicate receive antmas of the reception device. The transmission device transmits modulated signal zl(t) from transmit antenna 20 504#1 and transmits modulated signal B(t) from transmit antenna 504#2. Here, the modulated signals zl(t) and z2(t) are assumed to occupy the same (sharedcommon) frequency (bandwidth). The channel fluctuations in the transmit antennas of the transmission device and the antennas of the reception device are hll(t), h12(t), h2,(t), and hu(t), respectively. Assuming that receive antenna 505#1 25 of the reception device receives received signal rl(t) and that receive antenna 505#2 of the reception device receives received signal r2(t), the following relationship holds. [0 1091 math. 401 (formula 40) [Ol lo] 5 Fig. 6 pertains to the weighting scheme (preceding scheme) and the phase changing scheme of the present Embodiment. A weighting unit 600 is a combined version of the weighting units 308A and 308B fiom Fig. 3. As shown, stream sl(t) and stream s2(t) correspond to the baseband signals 307A and 307B of Fig. 3. That is, the streams sl(t) and s2(t) are baseband signals made up of an in-phase 10 component I and a quadrature component Q conforming to mapping by a modulation scheme such as QPSK, 16-QAM, and 64-QAM. As indicated by the frame configuration of Fig. 6, stream sl (t) is represented as sl(u) at symbol number u, as sl(u+l) at symbol number u+l, and so forth. Similarly, stream s2(t) is represented as s2(u) at symbol number u, as s2(u+1) at symbol number u+l, and so 15 forth. The weighting unit 600 takes the baseband signals 307A (sl(t)) and 307B (s2(t)) as well as the signal processing scheme information 3 15 fiom Fig. 3 as input, performs weighting in accordance with the signal processing scheme information 315, and outputs the weighted signals 309A (zl(t)) and 316B(df(t)) fiom Fig. 3. The phase changer 3 17B changes the phase of weighted signal 3 16B(d'(t)) and 20 outputs post-phase change signal 309B(d(t)). [Olll] Here, given vector W1 = (wll,w12) fiom the first row of the fixed precoding matrix F, zl(t) is expressible as Math. 41 (formula 41), below. [0112] 25 wath.411 (formula 4 1) [0113] Similarly, given vector W2 = (w21,w22) om the second row of the fixed preceding matrix F, and letting the phase changing formula applied by the phase 5 changer by y(t), then z2(t) is expressible as Math. 42 (formula 42), below. [0114] [Math. 421 (formula 42) 10 [0115] Here, y(t) is a phase changing formula following a predetermined scheme. For example, given a period (cycle) of four and time u, the phase changing formula is expressible as Math. 43 (formula 43), below. [0116] 15 [Math. 431 (formula 43) [0117] Similarly, the phase changing formula for time u+l may be, for example, as 20 given by Math. 44 (formula 44). [0118] [Math. 441 (formula 44) That is, the phase changing formula for time u+k is expressible as Math. 45 (formula 45). math. 451 (formula 45) [0121] 10 Note that Math. 43 (formula 43) through Math. 45 (formula 45) are given only as an example of regular phase changing. The regular change of phase is not restricted to a period (cycle) of four. Improved reception capabilities (the error-correction capabilities, to be exact) may potentially be promoted in the reception device by increasing the period (cycle) I5 number (this does not mean that a greater period (cycle) is better, though avoiding small numbers such as two is likely ideal). [0122] Furthermore, although Math. 43 (formula 43) through Math. 45 (formula 43, above, represent a configuration in which a change in phase is carried out 20 through rotation by consecutive predetermined phases (in the above formula, every n/2), the change in phase need not be rotation by a constant amount, but may also be random. For example, in accordance with the predetermined period (cycle) of y(t), the phase may be changed through sequential multiplication as shown in Math. 46 (formula 46) and Math. 47 (formula 47). The key point of regular phase changing 25 is that the phase of the modulated signal is regularly changed. The degree of phase 34 change is preferably as even as possible, such as from -n: radians to n: radians. However, given that this describes a distribution, random changes are also possible. 101231 [Math. 461 5 (formula 46) [0 1241 math. 471 (formula 47) [0 1251 As such, the weighting unit 600 of Fig. 6 performs precoding using fixed, predetermined precoding weights, and the phase changer 3 17B changes the phase of the signal input thereto while regularly varying the phase changing degree. 15 [0126] When a specialized precoding matrix is used in a LOS environment, the reception quality is likely to improve tremendously. However, depending on the direct wave conditions, the phase and amplitude components of the direct wave may greatly differ fiom the specialized precoding matrix, upon reception. The LOS 20 environment has certain rules. Thus, data reception quality is tremendously improved through a regular change applied to a transmit signal that obeys those rules. The present invention offers a signal processing scheme for improvements in the LOS environment. [0 1 271 Fig. 7 illustrates a sample configuration of a reception device 700 pertaining 5 to the present embodiment. Wireless unit 703-X receives, as input, received signal 702-X received by antenna 701-X, performs processing such as frequency conversion, quadrature demodulation, and the like, and outputs baseband signal 704-X. [0 1281 10 Channel fluctuation estimator 705-1 for modulated signal zl transmitted by the transmission device takes baseband signal 704-X as input, extracts reference symbol 501-1 for channel estimation from Fig. 5, estimates the value of hll from Math. 40 (formula 40), and outputs channel estimation signal 706-1. [0 1 291 15 Channel fluctuation estimator 7052 for modulated signal 22 transmitted by the transmission device takes baseband signal 704-X as input, extracts reference symbol 5012 for channel estimation fiom Fig. 5, estimates the value of h12 from Math. 40 (formula 40), and outputs channel estimation signal 706-2. [0130] 20 Wireless unit 703-Y receives, as input, received signal 702-Y received by antenna 701 - X, performs processing such as frequency conversion, quadrature demodulation, and the like, and outputs baseband signal 704-Y. Channel fluctuation estimator 707-1 for modulated signal zl transmitted by the transmission device takes baseband signal 704-Y as input, extracts reference 25 symbol 501-1 for channel estimation from Fig. 5, estimates the value of hZ1 from Math. 40 (formula 40), and outputs channel estimation signal 708-1. [0131] Channel fluctuation estimator 707-2 for modulated signal 22 transmitted by the transmission device takes baseband signal 70- as input, extracts reference symbol 5012 for channel estimation ffom Fig. 5, estimates the value of hU fiom Math. 40 (formula 40), and outputs channel estimation signal 7082. 5 [0132] A control information decoder 709 receives baseband signal 70- and baseband signal 704-Y as input, detects symbol 500-1 that indicates the transmission scheme ffom Fig. 5, and outputs a transmission scheme information signal 7 10 for the transmission device. 10 [0133] A signal processor 711 takes the baseband signals 70- and 704-Y, the channel estimation signals 706 -1, 706-2, 708-1, and 708-2, and the transmission scheme information signal 710 as input, performs detection and decoding, and then outputs received data 71 2-1 and 7 12-2. 15 [0134] Next, the operations of the signal processor 71 1 from Fig. 7 are described in detail. Fig. 8 illustrates a sample configuration of the signal processor 711 pertaining to the present embodiment. As shown, the signal processor 71 1 is primarily made up of an inner MIMO detector, soft-inlsoft-out decoders, and a 20 coefficient generator. Non-Patent Literature 2 and Non-Patent Literature 3 describe a scheme of iterative decoding using this structure. The MIMO system described in Non-Patent Literature 2 and Non-Patent Literature 3 is a spatial multiplexing MIMO system, while the present Embodiment differs fiom Non-Patent Literature 2 and Non-Patent Literature 3 in describing a MIMO system that regularly 25 changes the phase over time while using the same precoding matrix. Taking the (channel) matrix H(t) of Math. 36 (formula 36), then by letting the precoding weight matrix fiom Fig. 6 be F (here, a fixed precoding matrix remaining unchanged for a given received signal) and letting the phase changing formula used by the phase 37 changer from Fig. 6 be Y(t) (here, Y(t) changes over time t), then the receive vector R(t) = (rl(t),r2(t)lT and the stream vector S(t) = (~l(t),s2(t))th~e following fimction is derived: [0135] 5 wath. 481 (formula 48) where Here, the reception device may use the decoding schemes of Non-Patent Literature 2 and 3 on R(t) by computing H(t)xY(t)xF. Accordingly, the coefficient generator 8 19 from Fig. 8 takes a transmission scheme information signal 8 18 (corresponding to 71 0 from Fig. 7) indicated by the 15 transmission device (information for specifjmg the fixed preceding matrix in use and the phase changing pattern used when the phase is changed) and outputs a signal processing scheme information signal 820. [0137] The inner MIMO detector 803 takes the signal processing scheme 20 information signal as input and performs iterative detection and decoding using the signal and the relationship thereof to Math. 48 (formula 48). The operations thereof are described below. [0138] The processing unit illustrated in Fig. 8 uses a processing scheme, as 25 illustrated by Fig. 10, to perform iterative decoding (iterative detection). First, detection of one codeword (or one frame) of modulated signal (stream) sl and of 3 8 one codeword (or one frame) of modulated signal (stream) s2 is performed. As a result, the soft-inlsoft-out decoder obtains the log-likelihood ratio of each bit of the codeword (or frame) of modulated signal (stream) sl and of the codeword (or frame) of modulated signal (stream) s2. Next, the log-likelihood ratio is used to perform a 5 second round of detection and decoding. These operations are performed multiple times (these operations are hereinafter referred to as iterative decoding (iterative detection)). The following explanations center on the creation scheme of the log-likelihood ratio of a symbol at a specific time within one fiame. [0 1391 10 In Fig. 8, a memory 815 takes baseband signal 801X (corresponding to baseband signal 704-X from Fig. 7), channel estimation signal group 802X (corresponding to channel estimation signals 706-1 and 706-2 from Fig. 7), baseband signal 801Y (corresponding to baseband signal 704-Y from Fig. 7), and channel estimation signal group 802Y (corresponding to channel estimation signals 15 708-1 and 708-2 fkom Fig. 7) as input, executes (computes) H(t)xY(t)xF from Math. 48 (formula 48) in order to perform iterative decoding (iterative detection) and stores the resulting matrix as a transformed channel signal group. The memory 8 15 then outputs the above-described signals as needed, specifically as baseband signal 8 16X, transformed channel estimation signal group 8 17X, baseband signal 8 16Y, 20 and transformed channel estimation signal group 8 17Y. [0 1401 Subsequent operations are described separately for initial detection and for iterative decoding (iterative detection). (Initial Detection) The inner MlMO detector 803 takes baseband signal 801X, channel estimation signal group 802X, baseband signal 801Y, and channel estimation signal group 802Y as input. Here, the modulation scheme for modulated signal (stream) s 1 and modulated signal (stream) s2 is taken to be 1 6-QAM. 3 9 [0141] The inner MIMO detector 803 first computes H(t)xY(t)xF fiom the channel estimation signal groups 802X and 802Y, thus calculating a candidate signal point corresponding to baseband signal 801X. Fig. 11 represents such a calculation. In 5 Fig. 11, each black dot is a candidate signal point in the IQ plane. Given that the modulation scheme is 16-QAM, 256 candidate signal points exist. (However, Fig. 11 is only a representation and does not indicate all 256 candidate signal points.) Letting the four bits transmitted in modulated signal sl be bO, bl, b2, and b3 and the four bits transmitted in modulated signal s2 be b4, b5, b6, and b7, candidate signal 10 points corresponding to (bO, bl, b2, b3, b4, b5, b6, b7) are found in Fig. 11. The Euclidean squared distance between each candidate signal point and each received signal point 1101 (corresponding to baseband signal 801X) is then computed. The Euclidian squared distance between each point is divided by the noise variance 2. Accordingly, Ex@O, bl, b2, b3, b4, b5, b6, b7) is calculated. That is, Ex is the 15 Euclidian squared distance between a candidate signal point corresponding to (bO, bl, b2, b3, b4, b5, b6, b7) and a received signal point, divided by the noise variance. Here, each of the baseband signals and the modulated signals sl and s2 is a complex signal. [0 142) 20 Similarly, the inner MlMO detector 803 computes H(t)xY(t)xF fiom the channel estimation signal groups 802X and 802Y, calculates candidate signal points corresponding to baseband signal 801Y, computes the Euclidean squared distance between each of the candidate signal points and the received signal points (corresponding to baseband signal 801Y), and divides the Euclidean squared 25 distance by the noise variance $. Accordingly, EaO, bl, b2, b3, b4, b5, b6, b7) is calculated. That is, Ey is the Euclidian squared distance between a candidate signal point corresponding to (bO, bl, b2, b3, b4, b5, b6, b7) and a received signal point, divided by the noise variance. 40 [0143] Next, EXPO, bl, b2, b3, b4, b5, b6, b7) + Ey(bO, bl, b2, b3, b4, b5, b6, b7) = E(b0, bl, b2, b3, b4, b5, b6, b7) is computed. 5 The inner MIMO detector 803 outputs E(b0, bl, b2, b3, b4, b5, b6, b7) as a signal 804. Log-likelihood calculator 805A takes the signal 804 as input, calculates the log-likelihood of bits bO, bl, b2, and b3, and outputs log-likelihood signal 806A. Note that this log-likelihood calculation produces the log-likelihood of a bit being 1 10 and the log-likelihood of a bit being 0. The calculation scheme is as shown in Math. 28 (formula 28), Math. 29 (formula 29), and Math. 30 (formula 30), and the details are given by Non-Patent Literature 2 and 3. [0 1451 Similarly, log-likelihood calculator 805A takes the signal 804 as input, 15 calculates the log-likelihood of bits bO, bl, b2, and b3, and outputs log-likelihood signal 806B. A deinterleaver (807A) takes log-likelihood signal 806A as input, performs deinterleaving corresponding to that of the interleaver (the interleaver (304A) from Fig. 3), and outputs deinterleaved log-likelihood signal 808A. [0 1461 20 Similarly, a deinterleaver (807B) takes log-likelihood signal 806B as input, performs deinterleaving corresponding to that of the interleaver (the interleaver (304B) from Fig. 3), and outputs deinterleaved log-likelihood signal 808B. .[O 1471 Log-likelihood ratio calculator 809A takes deinterleaved log-likelihood 25 signal 808A as input, calculates the log-likelihood ratio of the bits encoded by encoder 302A from Fig. 3, and outputs log-likelihood ratio signal 8 10A. 101481 Similarly, log-likelihood ratio calculator 809B takes deinterleaved log-likelihood signal 808B as input, calculates the log-likelihood ratio of the bits encoded by encoder 302B fiom Fig. 3, and outputs log-likelihood ratio signal 8 10B. [0 1491 5 Soft-idsoft-out decoder 8 1 1A takes log-likelihood ratio signal 8 10A as input, performs decoding, and outputs decoded log-likelihood ratio 8 12A. Similarly, soft-inlsoft-out decoder 81 1B takes log-likelihood ratio signal 810B as input, performs decoding, and outputs decoded log-likelihood ratio 812B. [0150] 10 (Iterative Decoding (Iterative Detection), k Iterations) The interleaver (813A) takes the k-lth decoded log-likelihood ratio 8 12A decoded by the soft-idsoft-out decoder as input, performs interleaving, and outputs interleaved log-likelihood ratio 8 14A. Here, the interleaving pattern used by the interleaver (813A) is identical to that of the interleaver (304A) from Fig. 3. 15 [0151] Another interleaver (813B) takes the k-lth decoded log-likelihood ratio 812B decoded by the soft-in/soft-out decoder as input, performs interleaving, and outputs interleaved log-likelihood ratio 814B. Here, the interleaving pattern used by the other interleaver (813B) is identical to that of another interleaver (304B) from 20 Fig. 3. [0 1 521 The inner MIMO detector 803 takes baseband signal 816X, transformed channel estimation signal group 8 17X, baseband signal 8 16Y, transformed channel estimation signal group 817Y, interleaved log-likelihood ratio 814A, and interleaved 25 log-likelihood ratio 814B as input. Here, baseband signal 816X, transformed channel estimation signal group 8 17X, baseband signal 8 16Y, and transformed channel estimation signal group 817Y are used instead of baseband signal 801X, channel estimation signal group 802X, baseband signal 801Y, and channel 42 estimation signal group 802Y because the latter cause delays due to the iterative decoding. LO1531 The iterative decoding operations of the inner MIMO detector 803 differ 5 fkom the initial detection operations thereof in that the interleaved log-likelihood ratios 814A and 814B are used in signal processing for the former. The inner MIMO detector 803 first calculates E(b0, bl, b2, b3, b4, b5, b6, b7) in the same manner as for initial detection. In addition, the coefficients corresponding to Math. 11 (formula 11) and Math. 32 (formula 32) are computed fkom the interleaved 10 log-likelihood ratios 814A and 814B. The value of E(b0, bl, b2, b3, b4, b5, b6, b7) is corrected using the coefficients so calculated to obtain E1(bO, bl, b2, b3, b4, b5, b6, b7), which is output as the signal 804. [0 1541 Log-likelihood calculator 805A takes the signal 804 as input, calculates the 15 log-likelihood of bits bO, bl, b2, and b3, and outputs the log-likelihood signal 806A. Note that this log-likelihood calculation produces the log-likelihood of a bit being 1 and the log-likelihood of a bit being 0. The calculation scheme is as shown in Math. 3 1 (formula 31) through Math. 35 (formula 39, and the details are given by Non-Patent Literature 2 and 3. 20 [0155] Similarly, log-likelihood calculator 805B takes the signal 804 as input, calculates the log-likelihood of bits b4, b5, b6, and b7, and outputs the log-likelihood signal 806A. Operations performed by the deinterleaver onwards are similar to those performed for initial detection. 25 [0156] While Fig. 8 illustrates the configuration of the signal processor when performing iterative detection, this structure is not absolutely necessary as good reception improvements are obtainable by iterative detection alone. As long as the 43 components needed for iterative detection are present, the configuration need not include the interleavers 813A and 813B. In such a case, the inner MIMO detector 803 does not perform iterative detection. [0157] 5 The key point for the present Embodiment is the calculation of H(t)xY(t)xF. As shown in Non-Patent Literature 5 and the like, QR decomposition may also be used to perform initial detection and iterative detection. Also, as indicated by Non-Patent Literature 11, MMSE m i m u m Mean-Square Error) and ZF (Zero-Forcing) linear operations may be performed 10 based on H(t)xY(t)xF when performing initial detection. [0158] Fig. 9 illustrates the configuration of a signal processor, unlike that of Fig. 8, that serves as the signal processor for modulated signals transmitted by the transmission device fi-om Fig. 4. The point of difference from Fig. 8 is the number 15 of soft-inlsoft-out decoders. A soft-idsoft-out decoder 90 1 takes the log-likelihood ratio signals 810A and 810B as input, performs decoding, and outputs a decoded log-likelihood ratio 902. A distributor 903 takes the decoded log-likelihood ratio 902 as input for distribution. Otherwise, the operations are identical to those explained for Fig. 8. 20 [0159] As described above, when a transmission device according to the present Embodiment using a MIMO system transmits a plurality of modulated signals from a plurality of antennas, changing the phase over time while multiplying by the precoding matrix so as to regularly change the phase results in improvements to data 25 reception quality for a reception device in a LOS environment where direct waves are dominant, in contrast to a conventional spatial multiplexing MIMO system. [0 1601 In the present Embodiment, and particularly in the configuration of the reception device, the number of antennas is limited and explanations are given accordingly. However, the Embodiment may also be applied to a greater number of antennas. In other words, the number of antennas in the reception device does 5 not affect the operations or advantageous effects of the present Embodiment. [0161] Also, although LDPC codes are described as a particular example, the present Embodiment is not limited in this manner. Furthermore, the decoding scheme is not limited to the sum-product decoding example given for the 10 soft-idsoft-out decoder. Other soft-idsoft-out decoding schemes, such as the BCJR algorithm, SOVA, and the Max-Log-Map algorithm may also be used. Details are provided in Non-Patent Literature 6. [0 1621 In addition, although the present Embodiment is described using a 15 single-carrier scheme, no limitation is intended in this regard. The present Embodiment is also applicable to multi-carrier transmission. Accordingly, the present Embodiment may also be realized using, for example, spread-spectrum communications, OFDM (Orthogonal Frequency-Division Multiplexing), SC-FDMA (Single Carrier Frequency-Division Multiple Access), SC-OFDM 20 (Single Carrier Orthogonal Frequency-Division Multiplexing), wavelet OFDM as described in Non-Patent Literature 7, and so on. Furthermore, in the present Embodiment, symbols other than data symbols, such as pilot symbols (preamble, unique word, etc) or symbols transmitting control information, may be arranged within the frame in any manner. 25 [0163] The following describes an example in which OFDM is used as a multi-carrier scheme. Fig. 12 illustrates the configuration of a transmission device using OFDM. In Fig. 12, components operating in the manner described for Fig. 3 use identical reference numbers. [0 1641 5 OFDM-related processor 1201A takes weighted signal 309A as input, performs OFDM-related processing thereon, and outputs transmit signal 1202A. Similarly, OFDM-related processor 1201B takes post-phase change 309B as input, performs OFDM-related processing thereon, and outputs transmit signal 1202A [0 1651 10 Fig. 13 illustrates a sample configuration of the OFDM-related processors 1201A and 120 1B and onward fiom Fig. 12. Components 1301A through 13 10A belong between 1201A and 3 12A fiom Fig. 12, while components 1301B through 1310B belong between 1201B and 3 12B. 101661 15 Serial-to-parallel converter 1302A performs serial-to-parallel conversion on weighted signal 1301A (corresponding to weighted signal 309A fiom Fig. 12) and outputs parallel signal 1303A. [0 1671 Reorderer 1304A takes parallel signal 1303A as input, performs reordering 20 thereof, and outputs reordered signal 1305k Reordering is described in detail later. IFFT (Inverse Fast Fourier Transform) unit 1306A takes reordered signal 1305A as input, applies an IFFT thereto, and outputs post-IFFT signal 1307A. [0 1681 25 Wireless unit 1308A takes post-IFFT signal 1307A as input, performs processing such as fiequency conversion and amplification, thereon, and outputs modulated signal 1309A. Modulated signal 1309A is then output as radio waves by antenna 13 10A. 46 [O 1691 Serial-to-parallel converter 1302B performs serial-to-parallel conversion on weighted signal 130 1B (corresponding to post-phase change 309B from Fig. 12) and outputs parallel signal 1303B. 5 [0170] Reorderer 1304B takes parallel signal 1303B as input, performs reordering thereof, and outputs reordered signal 1305B. Reordering is described in detail later. IFFT unit 1306B takes reordered signal 1305B as input, applies an IFFT 10 thereto, and outputs post-IFFT signal 1307B. [0171] Wireless unit 1308B takes post-IFFT signal 1307B as input, performs processing such as frequency conversion and amplification thereon, and outputs modulated signal 1309B. Modulated signal 1309B is then output as radio waves by 15 antenna 13 10A. [0 1 721 The transmission device fiom Fig. 3 does not use a multi-carrier transmission scheme. Thus, as shown in Fig. 6, the change of phase is performed to achieve a period (cycle) of four and the post-phase change symbols are arranged 20 with respect to the time domain. As shown in Fig. 12, when multi-carrier transmission, such as OFDM, is used, then, naturally, precoded post-phase change symbols may be arranged with respect to the time domain as in Fig. 3, and this applies to each (sub-)carrier. However, for multi-carrier transmission, the arrangement may also be in the frequency domain, or in both the frequency domain 25 and the time domain. The following describes these arrangements. [0 1731 Figs. 14A and 14B indicate frequency on the horizontal axes and time on the vertical axes thereof, and illustrate an example of a symbol reordering scheme 47 used by the reorderers 1301A and 1301B from Fig. 13. The frequency axes are made up of (sub-)carriers 0 through 9. The modulated signals zl and 22 share common times (timing) and use a common frequency band. Fig. 14A illustrates a reordering scheme for the symbols of modulated signal zl, whle Fig. 14B illustrates 5 a reordering scheme for the symbols of modulated signal 22. With respect to the symbols of weighted signal 1301A input to serial-to-parallel converter 1302A, the assigned ordering is #0, #I, #2, #3, and so on. Here, given that the example deals with a period (cycle) of four, #0, #1, #2, and #3 are equivalent to one period (cycle). Similarly, #4n, #4n+l, #4n+2, and #4n+3 (n being a non-zero positive integer) are 10 also equivalent to one period (cycle). [0 1 741 As shown in Fig. 14A, symbols #O, #1, #2, #3, and so on are arranged in order, beginning at carrier 0. Symbols #O through #9 are given time $1, followed by symbols #lo through #19 which are given time #2, and so on in a regular 15 arrangement. Note that the modulated signals zl and 22 are complex signals. [0 1 751 Similarly, with respect to the symbols of weighted signal 1301B input to serial-to-parallel converter 1302B, the assigned ordering is #O, #1, #2, #3, and so on. Here, given that the example deals with a period (cycle) of four, a different change 20 of phase is applied to each of #O, #I, #2, and #3, which are equivalent to one period (cycle). Similarly, a different change of phase is applied to each of Mn, #4n+l, #4n+2, and #4n+3 (n being a non-zero positive integer), which are also equivalent to one period (cycle) [0 1 761 25 As shown in Fig. 14B, symbols #0, #I, #2, #3, and so on are arranged in order, beginning at carrier 0. Symbols #O through #9 are given time $1, followed by symbols #10 through #19 which are given time #2, and so on in a regular arrangement. 48 [0 1771 The symbol group 1402 shown in Fig. 14B corresponds to one period (cycle) of symbols when the phase changing scheme of Fig. 6 is used. Symbol #O is the symbol obtained by using the phase at time u in Fig. 6, symbol #1 is the 5 symbol obtained by using the phase at time u+l in Fig. 6, symbol #2 is the symbol obtained by using the phase at time u+2 in Fig. 6, and symbol #3 is the symbol obtained by using the phase at time u+3 in Fig. 6. Accordingly, for any symbol #x, symbol #x is the symbol obtained by using the phase at time u in Fig. 6 when x mod 4 equals 0 (i.e., when the remainder of x divided by 4 is 0, mod being the modulo 10 operator), symbol #x is the symbol obtained by using the phase at time u+l in Fig. 6 when x mod 4 equals 1, symbol #x is the symbol obtained by using the phase at time u+2 in Fig. 6 when x mod 4 equals 2, and symbol #x is the symbol obtained by using the phase at time u+3 in Fig. 6 when x mod 4 equals 3. [0178] 15 In the present Embodiment, modulated signal zl shown in Fig. 14A has not undergone a change of phase. As such, when using a multi-carrier transmission scheme such as OFDM, and unlike single carrier transmission, symbols may be arranged with respect to the fiequency domain. Of course, the symbol arrangement scheme is not limited to 20 those illustrated by Figs. 14A and 14B. Further examples are shown in Figs. 15A, 15B, 16A, and 16B. [0 1791 Figs. 15A and 15B indicate fiequency on the horizontal axes and time on the vertical axes thereof, and illustrate an example of a symbol reordering scheme 25 used by the reorderers 1301A and 1301B from Fig. 13 that differs from that of Figs. 14A and 14B. Fig. 15A illustrates a reordering scheme for the symbols of modulated signal zl, while Fig. 15B illustrates a reordering scheme for the symbols of modulated signal 22. Figs. 15A and 15B differ from Figs. 14A and 14B in that 49 different reordering schemes are applied to the symbols of modulated signal zl and to the symbols of modulated signal 22. In Fig. 15B, symbols #O through #5 are arranged at carriers 4 through 9, symbols #6 though #9 are arranged at carriers 0 through 3, and this arrangement is repeated for symbols #10 through #19. Here, as 5 in Fig. 14B, symbol group 1502 shown in Fig. 15B corresponds to one period (cycle) of symbols when the phase changing scheme of Fig. 6 is used. [0180] Figs. 16A and 16B indicate frequency on the horizontal axes and time on the vertical axes thereof, and illustrate an example of a symbol reordering scheme 1 0 used by the reorderers 130 1 A and 1 30 1 B from Fig. 1 3 that differs from that of Figs. 14A and 14B. Fig. 16A illustrates a reordering scheme for the symbols of modulated signal zl, while Fig. 16B illustrates a reordering scheme for the symbols of modulated signal 22. Figs. 16A and 16B differ fiom Figs. 14A and 14B in that, while Figs. 14A and 14B showed symbols arranged at sequential carriers, Figs. 16A 15 and 16B do not arrange the symbols at sequential carriers. Obviously, for Figs. 16A and 16B, different reordering schemes may be applied to the symbols of modulated signal zl and to the symbols of modulated signal 22 as in Figs. 15A and 15B. [0181] 20 Figs. 17A and 17B indicate frequency on the horizontal axes and time on the vertical axes thereof, and illustrate an example of a symbol reordering scheme used by the reorderers 1301A and 130 1B from Fig. 13 that differs from those of Figs. 14A through 16B. Fig. 17A illustrates a reordering scheme for the symbols of modulated signal zl and Fig. 17B illustrates a reordering scheme for the symbols of 25 modulated signal 22. While Figs. 14A through 16B show symbols arranged with respect to the frequency axis, Figs. 17A and 17B use the frequency and time axes together in a single arrangement. While Fig. 6 describes an example where a change of phase is performed in a four slot period (cycle), the following example describes an eight slot period (cycle). In Figs. 17A and 17B, the symbol group 1702 is equivalent to one period (cycle) of symbols when the phase changing scheme is used (i.e., to eight symbols) 5 such that symbol #O is the symbol obtained by using the phase at time u, symbol #1 is the symbol obtained by using the phase at time u+l, symbol #2 is the symbol obtained by using the phase at time u+2, symbol #3 is the symbol obtained by using the phase at time u+3, symbol #4 is the symbol obtained by using the phase at time u+4, symbol #5 is the symbol obtained by using the phase at time u+5, symbol #6 is 10 the symbol obtained by using the phase at time u+6, and symbol #7 is the symbol obtained by using the phase at time u+7. Accordingly, for any symbol #x, symbol #x is the symbol obtained by using the phase at time u when x mod 8 equals 0, symbol #x is the symbol obtained by using the phase at time u+l when x mod 8 equals 1, symbol #x is the symbol obtained by using the phase at time u+2 when x 15 mod 8 equals 2, symbol #x is the symbol obtained by using the phase at time u+3 when x mod 8 equals 3, symbol #x is the symbol obtained by using the phase at time u+4 when x mod 8 equals 4, symbol #x is the symbol obtained by using the phase at time u+5 when x mod 8 equals 5, symbol #x is the symbol obtained by using the phase at time u+6 when x mod 8 equals 6, and symbol #x is the symbol obtained by 20 using the phase at time u+7 when x mod 8 equals 7. In Figs. 17A and 17B four slots along the time axis and two slots along the frequency axis are used for a total of 4x2 = 8 slots, in which one period (cycle) of symbols is arranged. Here, given mxn symbols per period (cycle) (i-e., mxn different phases are available for multiplication), then n slots (carriers) in the frequency domain and m slots in the 25 time domain should be used to arrange the symbols of each period (cycle), such that m > n. This is because the phase of direct waves fluctuates slowly in the time domain relative to the frequency domain. Accordingly, the present Embodiment performs a regular change of phase that reduces the influence of steady direct waves. 5 1 Thus, the phase changing period (cycle) should preferably reduce direct wave fluctuations. Accordingly, m should be greater than n. Taking the above into consideration, using the time and frequency domains together for reordering, as shown in Figs. 17A and 17B, is preferable to using either of the frequency domain 5 or the time domain alone due to the strong probability of the direct waves becoming regular. As a result, the effects of the present invention are more easily obtained. However, reordering in the frequency domain may lead to diversity gain due the fact that frequency-domain fluctuations are abrupt. As such, using the frequency and time domains together for reordering is not always ideal. 10 [0183] Figs. 18A and 18B indicate frequency on the horizontal axes and time on the vertical axes thereof, and illustrate an example of a symbol reordering scheme used by the reorderers 130 1A and 1301B from Fig. 13 that differs from that of Figs. 17A and 14B. Fig. 18A illustrates a reordering scheme for the symbols of 15 modulated signal zl, while Fig. 18B illustrates a reordering scheme for the symbols of modulated signal 22. Much like Figs. 17A and 17B, Figs. 18A and 18B illustrate the use of the time and frequency domains, together. However, in contrast to Figs. 17A and 17B, where the frequency domain is prioritized and the time domain is used for secondary symbol arrangement, Figs. 18A and 18B 20 prioritize the time domain and use the frequency domain for secondary symbol arrangement. In Fig. 18B, symbol group 1802 corresponds to one period (cycle) of symbols when the phase changing scheme is used. [0 1 841 In Figs. 17A, 17B, 18A, and 18B, the reordering scheme applied to the 25 symbols of modulated signal zl and the symbols of modulated signal 22 may be identical or may differ as in Figs. 15A and 15B. Both approaches allow good reception quality to be obtained. Also, in Figs. 17A, 17B, 18A, and 18B, the symbols may be arranged non-sequentially as in Figs. 16A and 16B. Both approaches allows good reception quality to be obtained. [0185] Fig. 22 indicates frequency on the horizontal axis and time on the vertical 5 axis thereof, and illustrates an example of a symbol reordering scheme used by the reorderers 1301A and 1301B fiom Fig. 13 that differs from the above. Fig. 22 illustrates a regular phase changing scheme using four slots, similar to times u through u+3 fiom Fig. 6. The characteristic feature of Fig. 22 is that, although the symbols are reordered with respect the frequency domain, when read along the time 10 axis, a periodic shift of n (n = 1 in the example of Fig. 22) symbols is apparent. The frequency-domain symbol group 2210 in Fig. 22 indicates four symbols to which the change of phase is applied at times u through u+3 fiom Fig. 6. [0186] Here, symbol #O is obtained through a change of phase at time u, symbol #1 15 is obtained through a change of phase at time u+l, symbol #2 is obtained through a change of phase at time u+2, and symbol #3 is obtained through a change of phase at time u+3. [0187] Similarly, for frequency-domain symbol group 2220, symbol #4 is obtained 20 through a change of phase at time u, symbol #5 is obtained through a change of phase at time u+l, symbol #6 is obtained through a change of phase at time u+2, and symbol #7 is obtained through a change of phase at time u+3. [0188] The above-described change of phase is applied to the symbol at time $1. 25 However, in order to apply periodic shifting in the time domain, the following phase changes are applied to symbol groups 2201,2202,2203, and 2204. [O 1 891 For time-domain symbol group 2201, symbol #O is obtained through a change of phase at time u, symbol #9 is obtained through a change of phase at time u+l, symbol #18 is obtained through a change of phase at time u+2, and symbol #27 is obtained through a change of phase at time u+3. [0 1 901 For time-domain symbol group 2202, symbol #28 is obtained through a change of phase at time u, symbol #1 is obtained through a change of phase at time u+l, symbol #10 is obtained through a change of phase at time u+2, and symbol #19 is obtained through a change of phase at time u+3. [0191] For time-domain symbol group 2203, symbol #20 is obtained through a change of phase at time u, symbol #29 is obtained through a change of phase at time u+l, symbol #2 is obtained through a change of phase at time u+2, and symbol #11 is obtained through a change of phase at time u+3. [0 192) For time-domain symbol group 2204, symbol #12 is obtained through a change of phase at time u, symbol #21 is obtained through a change of phase at time u+l, symbol #30 is obtained through a change of phase at time u+2, and symbol #3 is obtained through a change of phase at time u+3. [0 1931 The characteristic feature of Fig. 22 is seen in that, taking symbol #11 as an example, the two neighbouring symbols thereof having the same time in the frequency domain (#lo and #12) are both symbols changed using a different phase than symbol #11, and the two neighbouring symbols thereof having the same carrier in the time domain (#2 and #20) are both symbols changed using a different phase than symbol #11. This holds not only for symbol #11, but also for any symbol having two neighboring symbols in the frequency domain and the time domain. Accordingly, phase changing is effectively carried out. This is highly likely to 54 improve date reception quality as influence from regularizing direct waves is less prone to reception. [0 1 943 Although Fig. 22 illustrates an example in which n = 1, the invention is not 5 limited in this manner. The same may be applied to a case in which n = 3. Furthermore, although Fig. 22 illustrates the realization of the above-described effects by arranging the symbols in the frequency domain and advancing in the time domain so as to achieve the characteristic effect of imparting a periodic shift to the symbol arrangement order, the symbols may also be randomly (or regularly) 10 arranged to the same effect. [O 1 951 [Embodiment 21 In Embodiment 1, described above, phase changing is applied to a weighted @recoded with a fixed precoding matrix) signal z(t). The following Embodiments 15 describe various phase changing schemes by which the effects of Embodiment 1 may be obtained. [0196] In the above-described Embodiment, as shown in Figs. 3 and 6, phase changer 3 17B is configured to perform a change of phase on only one of the signals 20 output by the weighting unit 600. However, phase changing may also be applied before precoding is performed by the weighting unit 600. In addition to the components illustrated in Fig. 6, the transmission device may also feature the weighting unit 600 before the phase changer 3 17B, as shown in Fig. 25. 25 [0197] In such circumstances, the following configuration is possible. The phase changer 317B performs a regular change of phase with respect to baseband signal s2(t), on which mapping has been performed according to a selected modulation 55 scheme, and outputs s2'(t) = s2(t)y(t) (where y(t) varies over time t). The weighting unit 600 executes precoding on s2't, outputs z2(t) = W2s2'(t) (see Math. 42 (formula 42)) and the result is then transmitted. [0 1981 5 Alternatively, phase changing may be performed on both modulated signals sl(t) and s2(t). As such, the transmission device is configured so as to include a phase changer taking both signals output by the weighting unit 600, as shown in Fig. 26. [0 1991 10 Like phase changer 3 17B, phase changer 3 17A performs regular a regular change of phase on the signal input thereto, and as such changes the phase of signal zl '(t) precoded by the weighting unit. Post-phase change signal zl(t) is then output to a transmitter. [0200] 15 However, the phase changing rate applied by the phase changers 317A and 3 17B varies simultaneously in order to perform the phase changing shown in Fig. 26. (The following describes a non-limiting example of the phase changing scheme.) For time u, phase changer 3 17A fiom Fig. 26 performs the change of phase such that zl(t) = yI(t)zl '(t), while phase changer 31 7B performs the change of phase such that 20 z2(t) = y2(t)z2'(t). For example, as shown in Fig. 26, for time u, y,(u) = @ and y2(u) = e-*, for time u+l, yl(u+l) = dd4 and y2(u+l) = e73d4, and for time u+k, y,(u+k) = e'b"I4 and y2(u+k) = d(R3d4-d). Here, the regular phase changing period (cycle) may be the same for both phase changers 3 17A and 3 17B, or may vary for each. 25 [0201] Also, as described above, a change of phase may be performed before precoding is performed by the weighting unit. In such a case, the transmission device should be configured as illustrated in Fig. 27. 56 ~02023 When a change of phase is carried out on both modulated signals, each of the transmit signals is, for example, control infomation that includes information about the phase changing pattern. By obtaining the control information, the 5 reception device knows the phase changing scheme by which the transmission device regularly varies the change, i.e., the phase changing pattern, and is thus able to demodulate (decode) the signals correctly. [0203] Next, variants of the sample configurations shown in Figs. 6 and 25 are 10 described with reference to Figs. 28 and 29. Fig. 28 differs fiom Fig. 6 in the inclusion of phase change ON/OFF infomation 2800 and in that the change of phase is performed on only one of zll(t) and z2'(t) (i.e., performed on one of zll(t) and z2'(t), which have identical times or a common fkequency). Accordingly, in order to perform the change of phase on one of zl '(t) and z2'(t), the phase changers 15 317A and 317B shown in Fig. 28 may each be ON, and performing the change of phase, or OFF, and not performing the change of phase. The phase change ON/OFF information 2800 is control information therefor. The phase change ONIOFF information 2800 is output by the signal processing scheme information generator 3 14 shouin in Fig. 3. 20 [0204] Phase changer 317A of Fig. 28 changes the phase to produce zl(t) = yl(t)zlr(t), while phase changer 317B changes the phase to produce z2(t) = y2(t)df(t). [0205] 25 Here, a change of phase having a period (cycle) of four is, for example, applied to zll(t). (Meanwhile, the phase of z2'(t) is not changed.) Accordingly, for time u, y,(u) = do and y2(u) = 1, for time u+l, yl(u+l) = eiXR and y2(u+l) = 1, for time u+2, yl(u+2) = @ and y2(u+2) = 1, and for time u+3, yl(u+3) = e'3& and y2(u+3) = 1. [0206] Next, a change of phase having a period (cycle) of four is, for example, 5 applied to dr(t). (Meanwhile, the phase of zlr(t) is not changed.) Accordingly, for time u+4, yl(u+4) = 1 and y2(u+4) = dQ, for time u+5, yl(u+5) = 1 and y2(u+5) = dfl, for time u+6, yl(u+6) = 1 and y2(u+6) = ei", and for time u+7, yl(u+7) = 1 and y2(u+7) = Cfl. [0207] 10 Accordingly, given the above examples. for any time 8k, y1(8k) = $ and yz(8k) = 1, for any time 8k+l, y1(8k+l) = dfl and y2(8k+l) = 1, for any time 8k+2, y1(8k+2) = e'A and y2(8k+2) = 1, for any time 8k+3, y1(8k+3) = d3& and yz(8k+3) = 1, 15 for any time 8k+4, y1(8k+4) = 1 and y2(8k+4) = do, for any time 8k+5, yl(8k+3) = 1 and yz(8k+5) = e'&, for any time 8k+6, y1(8k+6) = 1 and y2(8k+6) = e'", and for any time 8k+7, yl(8k+7) = 1 and yz(8k+7) = e'3fl. [0208] 20 As described above, there are two intervals, one where the change of phase is performed on zlr(t) only, and one where the change of phase is performed on z2'(t) only. Furthermore, the two intervals form a phase changing period (cycle). While the above explanation describes the interval where the change of phase is performed on zlr(t) only and the interval where the change of phase is performed on 25 d'(t) only as being equal, no limitation is intended in this manner. The two intervals may also differ. In addition, while the above explanation describes performing a change of phase having a period (cycle) of four on zlr(t) only and then performing a change of phase having a period (cycle) of four on z2'(t) only, no 5 8 limitation is intended in this manner. The changes of phase may be performed on zl'(t) and on z2'(t) in any order (e.g., the change of phase may alternate between being performed on zl '(t) and on z2'(t), or may be performed in random order). Phase changer 317A of Fig. 29 changes the phase to produce slf(t) = 5 yl(t)sl (t), while phase changer 3 17B changes the phase to produce s2'(t) = y2(t)s2(t). [0209] Here, a change of phase having a period (cycle) of four is, for example, applied to sl(t). (Meanwhile, s2(t) remains unchanged). Accordingly, for time u, yl(u) = and y2(u) = 1, for time u+l, yl(u+l) = em and y2(u+l) = 1, for time u+2, 10 yl(u+2) = e'n and y2(u+2) = 1, and for time u+3, yl(u+3) = e'3"/2 and y2(u+3) = 1. [02 1 01 Next, a change of phase having a period (cycle) of four is, for example, applied to s2(t). (Meanwhile, sl (t) remains unchanged). Accordingly, for time u+4, yl(u+4) = 1 and y2(u+4) = $, for time u+5, yl(u+5) = 1 and y2(u+5) = dfl, for time 15 u+6, yl(u+6) = 1 and y2(u+6) = e'n, and for time u+7, yl(u+7) = 1 and y2(u+7) = e'3fl 102 1 11 Accordingly, given the above examples, for any time 8k, y1(8k) = e" and y2(8k) = 1, 20 for any time 8k+l, y1(8k+l) = GXR and y2(8k+l) = 1, for any time 8kt-2, y1(8k+2) = e'X and y2(8k+2) = 1, for any time 8k+3, y1(8k+3) = d3XRan d y2(8k+3) = 1, for any time 8k+4, y1(8k+4) = 1 and y2(8k+4) = 2, for any time 8k+5, y1(8k+5) = 1 and y2(8k+5) = dd, 25 for any time 8k+6, y1(8k+6) = 1 and y2(8k+6) = e'n, and for any time 8ki-7, y1(8k+7) = 1 and y2(8k+7) = d3&. 102 121 As described above, there are two intervals, one where the change of phase is performed on sl(t) only, and one where the change of phase is performed on s2(t) only. Furthermore, the two intervals form a phase changing period (cycle). Although the above explanation describes the interval where the change of phase is 5 performed on sl(t) only and the interval where the change of phase is performed on s2(t) only as being equal, no limitation is intended in this manner. The two intervals may also differ. In addition, while the above explanation describes performing the change of phase having a period (cycle) of four on sl(t) only and then performing the change of phase having a period (cycle) of four on s2(t) only, no 10 limitation is intended in this manner. The changes of phase may be performed on sl(t) and on s2(t) in any order (e.g., may alternate between being performed on sl(t) and on s2(t), or may be performed in random order). Accordingly, the reception conditions under which the reception device receives each transmit signal zl(t) and z2(t) are equalized. By periodically 15 switching the phase of the symbols in the received signals zl(t) and z2(t), the ability of the error corrected codes to correct errors may be improved, thus ameliorating received signal quality in the LOS environment. [02 131 Accordingly, Embodiment 2 as described above is able to produce the same 20 results as the previously described Embodiment 1. Although the present Embodiment used a single-carrier scheme, i.e., time domain phase changing, as an example, no limitation is intended in this regard. The same effects are also achievable using multi-carrier transmission. Accordingly, the present Embodiment may also be realized using, for example, spread-spectrum 25 communications, OFDM, SC-FDMA (Single Carrier Frequency-Division Multiple Access), SC-OFDM, wavelet OFDM as described in Non-Patent Literature 7, and so on. As previously described, while the present Embodiment explains the change of phase as changing the phase with respect to the time domain t, the phase may 60 alternatively be changed with respect to the frequency domain as described in Embodiment 1. That is, considering the phase changing scheme in the time domain t described in the present Embodim'ent and replacing t with f (f being the ((sub-) carrier) frequency) leads to a change of phase applicable to the fiequency 5 domain. Also, as explained above for Embodiment 1, the phase changing scheme of the present Embodiment is also applicable to changing the phase with respect both the time domain and the fiequency domain. [02 141 Accordingly, although Figs. 6, 25, 26, and 27 illustrate changes of phase in 10 the time domain, replacing time t with carrier f in each of Figs. 6, 25, 26, and 27 corresponds to a change of phase in the fiequency domain. In other words, replacing (t) with (t, f ) where t is time and f is frequency corresponds to performing the change of phase on time-frequency blocks. [02 151 15 Furthermore, in the present Embodiment, symbols other than data symbols, such as pilot symbols (preamble, unique word, etc) or symbols transmitting control information, may be arranged within the frame in any manner. [02 161 [Embodiment 31 20 Embodiments 1 and 2, described above, discuss regular changes of phase. Embodiment 3 describes a scheme of allowing the reception device to obtain good received signal quality for data, regardless of the reception device arrangement, by considering the location of the reception device with respect to the transmission device. 25 [0217] Embodiment 3 concerns the symbol arrangement within signals obtained through a change of phase. Fig. 31 illustrates an example of frame configuration for a portion of the symbols within a signal in the time-frequency domain, given a transmission scheme where a regular change of phase is performed for a multi-carrier scheme such as OFDM. 5 [0218] First, an example is explained in which the change of phase is performed one of two baseband signals, precoded as explained in Embodiment 1 (see Fig. 6). (Although Fig. 6 illustrates a change of phase in the time domain, switching time t with carrier f in Fig. 6 corresponds to a change of phase in the frequency 10 domain. In other words, replacing (t) with (t, 0 where t is time and f is frequency corresponds to performing phase changes on time-frequency blocks.) Fig. 31 illustrates the frame configuration of modulated signal d', which is input to phase changer 317B from Fig. 12. Each square represents one symbol (although both signals sl and s2 are included for precoding purposes, depending on 15 the precoding matrix, only one of signals sl and s2 may be used). [02 191 Consider symbol 3 100 at carrier 2 and time $2 of Fig. 3 1. The carrier here described may alternatively be termed a sub-carrier. Within carrier 2, there is a very strong correlation between the channel 20 conditions for symbol 3100 at carrier 2, time $2 and the channel conditions for the time domain nearest-neighbour symbols to time $2, i.e., symbol 3013 at time $1 and symbol 3 10 1 at time $3 within carrier 2. [0220] Similarly, for time $2, there is a very strong correlation between the channel 25 conditions for symbol 3 100 at carrier 2, time $2 and the channel conditions for the frequency-domain nearest-neighbour symbols to carrier 2, i-e., symbol 3104 at carrier 1, time $2 and symbol 3 104 at time $2, carrier 3. [022 1 ] 62 As described above, there is a very strong correlation between the channel conditions for symbol 3 100 and the channel conditions for symbols 3 101, 3102, 3103, and 3104. The present description considers N different phases (N being an integer, N 5 > 2) for multiplication in a transmission scheme where the phase is regularly changed. The symbols illustrated in Fig. 31 are indicated as do, for example. This signifies that this symbol is signal 22' fiom Fig. 6 phase-changed through multiplication by d'. That is, the values indicated in Fig. 31 for each of the symbols are the values of y(t) fiom Math. 42 (formula 42), which are also the values 10 of z2(t) = y2(t)df(t) described in Embodiment 2. [0222] The present Embodiment takes advantage of the high correlation in channel conditions existing between neigbouring symbols in the frequency domain andlor neighbouring symbols in the time domain in a symbol arrangement enabling high 15 data reception quality to be obtained by the reception device receiving the phase-changed symbols. [0223] In order to achieve this high data reception quality, conditions #1 and #2 are necessary. 20 (Condition #I) As shown in Fig. 6, for a transmission scheme involving a regular change of phase performed on precoded baseband signal 22' using multi-carrier transmission such as OFDM, time X, carrier Y is a symbol for transmitting data (hereinafter, data symbol), neighbouring symbols in the time domain, i-e., at time X-1, carrier Y and 25 at time X+1, carrier Y are also data symbols, and a different change of phase should be performed on precoded baseband signal 22' corresponding to each of these three data symbols, i.e., on precoded baseband signal 22' at time X, carrier Y, at time X-1, carrier Y and at time X+ 1, carrier Y. 63 [0224] (Condition #2) As shown in Fig. 6, for a transmission scheme involving a regular change of phase performed on precoded baseband signal 22' using multi-carrier transmission 5 such as OFDM, time X, carrier Y is a data symbol, neighbouring symbols in the frequency domain, i.e., at time X, carrier Y-1 and at time X, carrier Y+1 are also data symbols, and a different change of phase should be performed on precoded baseband signal 22' corresponding to each of these three data symbols, i.e., on precoded baseband signal 22' at time X, carrier Y, at time X, carrier Y-1 and at time 10 X,carrierY+l. [0225] Ideally, data symbols satisfylng Condition #I should be present. Similarly, data symbols satisfylng Condition #2 should be present. The reasons supporting Conditions #1 and #2 are as follows. 15 [0226] A very strong correlation exists between the channel conditions of given symbol of a transmit signal (hereinafter, symbol A) and the channel conditions of the symbols neighbouring symbol A in the time domain, as described above. [0227] 20 Accordingly, when three neighbouring symbols in the time domain each have different phases, then despite reception quality degradation in the LOS environment (poor signal quality caused by degradation in conditions due to direct wave phase relationships despite high signal quality in terms of SNR) for symbol A, the two remaining symbols neighbouring symbol A are highly likely to provide good 25 reception quality. As a result, good received signal quality is achievable after error correction and decoding. [0228] Similarly, a very strong correlation exists between the channel conditions of given symbol of a transmit signal (hereinafter, symbol A) and the channel conditions of the symbols neighbouring symbol A in the fiequency domain, as described above. [0229] 5 Accordingly, when three neighbouring symbols in the frequency domain each have different phases, then despite reception quality degradation in the LOS environment (poor signal quality caused by degradation in conditions due to direct wave phase relationships despite high signal quality in terms of SNR) for symbol A, the two remaining symbols neighbouring symbol A are highly likely to provide good 10 reception quality. As a result, good received signal quality is achievable after error correction and decoding. [0230] Combining Conditions #1 and #2, ever greater data reception quality is likely achievable for the reception device. Accordingly, the following Condition 15 #3 can be derived. [023 11 (Condition #3) As shown in Fig. 6, for a transmission scheme involving a regular change of phase performed on precoded baseband signal 22' using multi-carrier transmission 20 such as OFDM, time X, carrier Y is a data symbol, neighbouring symbols in the time domain, i.e., at time X-1, carrier Y and at time X+I, carrier Y are also data symbols, and neighbouring symbols in the fi-equency domain, i.e., at time X, canier Y-1 and at time X, carrier Y+l are also data symbols, and a different change in phase should be performed on precoded baseband signal 22' corresponding to each 25 of these five data symbols, i.e., on precoded baseband signal 22' at time X, carrier Y, at time X, carrier Y-1, at time X, carrier Y+1, at a time X-1, carrier Y, and at time X+1, carrier Y. [0232] Here, the different changes in phase are as follows. Changes in phase are defined from 0 radians to 2n radians. For example, for time X, carrier Y, a phase change of dBxYis applied to precoded baseband signal 22' fkom Fig. 6, for time X-1, carrier Y, a phase change of e' 'OX-1,Y is applied to precoded baseband signal 22' from 5 Fig. 6, for time X+1, carrier Y, a phase change of gX+isl yapYplie d to precoded baseband signal 22' from Fig. 6, such that 0 i €IXy < 2n, 0 i < 27t, and 0 I < 2n, all units being in radians. Accordingly, for Condition #1, it follows that # OX-l,Y, # OX+l,Y, and that OX-l,Y # OX+l,Y. Similarly, for Condition #2, it follows that exY # OYY-l, # OX,Y+l, and that OxY-l # OX,Y+l. And, for 10 Condition #3, it follows that # exY# OXYf OXY-l, OXY f OX,Y-l, OX-1,~# ~x+I,Y,O X-I,Y # ~x,Y-1, ex-I,Y# ~x+I,Y, ~ X + I , Y# &-I,Y, ~X+I,Y# ~x,Y+aIn,d that 0X,Y-l # Qx,Y+l. LO2331 Ideally, a data symbol should satisfy Condition #3. 15 Fig. 3 1 illustrates an example of Condition #3 where symbol A corresponds to symbol 3100. The symbols are arranged such that the phase by which precoded baseband signal 22' fkom Fig. 6 is multiplied differs for symbol 3100, for both neighbouring symbols thereof in the time domain 3 101 and 3 102, and for both neighbouring symbols thereof in the frequency domain 3102 and 3104. 20 Accordingly, despite received signal quality degradation of symbol 3100 for the receiver, good signal quality is highly likely for the neighbouring signals, thus guaranteeing good signal quality after error correction. [0234] Fig. 32 illustrates a symbol arrangement obtained through phase changes 25 under these conditions. As evident fiom Fig. 32, with respect to any data symbol, a different change in phase is applied to each neighbouring symbol in the time domain and in the frequency domain. As such, the ability of the reception device to correct errors may be improved. [0235] In other words, in Fig. 32, when all neighbouring symbols in the time 5 domain are data symbols, Condition #1 is satisfied for all Xs and all Ys. Similarly, in Fig. 32, when all neighbouring symbols in the frequency domain are data symbols, Condition #2 is satisfied for all Xs and all Ys. [023 61 Similarly, in Fig. 32, when all neighbouring symbols in the frequency 10 domain are data symbols and all neighbouring symbols in the time domain are data symbols, Condition #3 is satisfied for all Xs and all Ys. LO2371 The following describes an example in which a change of phase is performed on two precoded baseband signals, as explained in Embodiment 2 (see 15 Fig. 26). When a change of phase is performed on precoded baseband signal zl' and precoded baseband signal 22' as shown in Fig. 26, several phase changing schemes are possible. The details thereof are explained below. [023 81 20 Scheme 1 involves a change in phase performed on precoded baseband signal 22' as described above, to achieve the change in phase illustrated by Fig. 32. In Fig. 32, a change of phase having a period (cycle) of 10 is applied to precoded baseband signal 22'. However, as described above, in order to satisfy Conditions #1, #2, and #3, the change in phase applied to precoded baseband signal 22' at each 25 (sub-)carrier varies over time. (Although such changes are applied in Fig. 32 with a period (cycle) of ten, other phase changing schemes are also possible.) Then, as shown in Fig. 33, the change in phase performed on precoded baseband signal 21' produces a constant value that is one-tenth of that of the change in phase performed 67 on precoded baseband signal 22'. In Fig. 33, for a period (cycle) (of change in phase performed on precoded baseband signal 22') including time $1, the value of the change in phase performed on precoded baseband signal zl' is p. Then, for the next period (cycle) (of change in phase performed on precoded baseband signal 22') 5 including time $2, the value of the change in phase performed on precoded baseband signal z 1 ' is dfl, and so on. 102391 The symbols illustrated in Fig. 33 are indicated as dU, for example. This s i d l e s that this symbol is signal zl' from Fig. 26 on which a change in phase as 10 been applied through multiplication by 8. That is, the values indicated in Fig. 33 for each of the symbols are the values of zll(t) = y2(t)zlf(t) described in Embodiment 2 for yl(t). [0240] As shown in Fig. 33, the change in phase performed on precoded baseband 15 signal zl' produces a constant value that is one-tenth that of the change in phase performed on precoded baseband signal 22' such that the post-phase change value varies with the number of each period (cycle). (As described above, in Fig. 33, the value is @for the first period (cycle), gd9 for the second period (cycle), and so on.) As described above, the change in phase performed on precoded baseband 20 signal 22' has a period (cycle) of ten, but the period (cycle) can be effectively made greater than ten by taking the change in phase applied to precoded baseband signal zl' and to precoded baseband signal 22' into consideration. Accordingly, data reception quality may be improved for the reception device. [024 11 25 Scheme 2 involves a change in phase of precoded baseband signal 22' as described above, to achieve the change in phase illustrated by Fig. 32. In Fig. 32, a change of phase having a period (cycle) of ten is applied to precoded baseband signal 22'. However, as described above, in order to satisfy Conditions #1, #2, and 6 8 #3, the change in phase applied to precoded baseband signal 22' at each (sub-)carrier varies over time. (Although such changes are applied in Fig. 32 with a period (cycle) of ten, other phase changing schemes are also possible.) Then, as shown in Fig. 30, the change in phase performed on precoded baseband signal zl' differs from that 5 performed on precoded baseband signal 22' in having a period (cycle) of three rather than ten. [0242] The symbols illustrated in Fig. 30 are indicated as 8, for example. This signifies that this symbol is signal zl' from Fig. 26 to which a change in phase has 10 been applied through multiplication by d'. That is, the values indicated in Fig. 30 for each of the symbols are the values of zl(t) = yl(t)zl '(t) described in Embodiment 2 for yl(t). LO2431 As described above, the change in phase performed on precoded baseband 15 signal 22' has a period (cycle) of ten, but by taking the changes in phase applied to precoded baseband signal zlf and precoded baseband signal 22' into consideration, the period (cycle) can be effectively made equivalent to 30 for both precoded baseband signals zl' and 22'. Accordingly, data reception quality may be improved for the reception device. An effective way of applying scheme 2 is to 20 perform a change in phase on precoded baseband signal zl' with a period (cycle) of N and perform a change in phase on precoded baseband signal 22' with a period (cycle) of M such that N and M are coprime. As such, by taking both precoded baseband signals zl' and 22' into consideration, a period (cycle) of NxM is easily achievable, effectively making the period (cycle) greater when N and M are 25 coprime. [0244] The above describes an example of the phase changing scheme pertaining to Embodiment 3. The present invention is not limited in this manner. As explained 69 for Embodiments 1 and 2, a change in phase may be performed with respect the frequency domain or the time domain, or on time-frequency blocks. Similar improvement to the data reception quality can be obtained for the reception device in all cases. 5 LO2451 The same also applies to frames having a configuration other than that described above, where pilot symbols (SP (Scattered Pilot) and symbols transmitting control information are inserted among the data symbols. The details of change in phase in such circumstances are as follows. 10 [0246] Figs. 47A and 47B illustrate the frame configuration of modulated signals (precoded baseband signals) zl or zl' and 22' in the time-frequency domain. Fig. 47A illustrates the frame configuration of modulated signal @recoded baseband signals) zl or zl' while Fig. 47B illustrates the frame configuration of modulated 15 signal @recoded baseband signals) 22'. In Figs. 47A and 47B, 4701 marks pilot symbols while 4702 marks data symbols. The data symbols 4702 are symbols on which precoding or precoding and a change in phase have been performed. [0247] Figs. 47A and 47B, like Fig. 6, indicate the arrangement of symbols when a 20 change in phase is applied to precoded baseband signal 22' (while no change of phase is performed on precoded baseband signal 21). (Although Fig. 6 illustrates a change in phase with respect to the time domain, switching time t with carrier f in Fig. 6 corresponds to a change in phase with respect to the frequency domain. In other words, replacing (t) with (t, f ) where t is time and f is frequency corresponds to 25 performing a change of phase on time-frequency blocks.) Accordingly, the numerical values indicated in Figs. 47A and 47B for each of the symbols are the values of precoded baseband signal 22' after the change in phase. No values are given for the symbols of precoded baseband signal zl' (21) as no change in phase is performed thereon. [0248] The key point of Figs. 47A and 47B is that the change in phase is performed 5 on the data symbols of precoded baseband signal z2', i.e., on precoded symbols. (The symbols under discussion, being precoded, actually include both symbols sl and s2.) Accordingly, no change of phase is performed on the pilot symbols inserted into 22'. [0249] 10 Figs. 48A and 48B illustrate the fiarne configuration of modulated signals (precoded baseband signals) zl or zl' and 22' in the time-fi-equency domain. Fig. 48A illustrates the fi-ame configuration of modulated signal (precoded baseband signals) zl or zl' while Fig. 47B illustrates the fi-ame configuration of modulated signal (precoded baseband signals) 22'. In Figs. 48A and 48B, 4701 marks pilot 15 symbols while 4702 marks data symbols. The data symbols 4702 are symbols on which precoding, or precoding and a change in phase, have been performed. LO25 01 Figs. 48A and 48B, like Fig. 26, indicate the arrangement of symbols when a change in phase is applied to precoded baseband signal zl' and to precoded 20 baseband signal 22'. (Although Fig. 26 illustrates a change in phase with respect to the time domain, switching time t with carrier f in Fig. 26 corresponds to a change in phase with respect to the frequency domain. In other words, replacing (t) with (t, f) where t is time and f is fi-equency corresponds to performing a change of phase on time-frequency blocks.) Accordingly, the numerical values indicated in Figs. 48A 25 and 48B for each of the symbols are the values of precoded baseband signal zl' and 22' after the change in phase. [025 11 The key point of Fig. 47 is that a change of phase is performed on the data symbols of precoded baseband signal zl', that is, on the precoded symbols thereof, and on the data symbols of precoded baseband signal z2', that is, on the precoded symbols thereof. (The symbols under discussion, being precoded, actually include 5 both symbols sl and s2.) Accordingly, no change of phase is performed on the pilot symbols inserted in zl', nor on the pilot symbols inserted in 22'. [0252] Figs. 49A and 49B illustrate the frame configuration of modulated signals (precoded baseband signals) zl or zl' and 22' in the time-frequency domain. Fig. 10 49A illustrates the frame configuration of modulated signal (precoded baseband signals) zl or zl' while Fig. 49B illustrates the frame configuration of modulated signal (precoded baseband signal) 22'. In Figs. 49A and 49B, 4701 marks pilot symbols, 4702 marks data symbols, and 4901 marks null symbols for which the in-phase component of the baseband signal I = 0 and the quadrature component Q = 15 0. As such, data symbols 4702 are symbols on which precoding or precoding and the change in phase have been performed. Figs. 49A and 49B differ from Figs. 47A and 47B in the configuration scheme for symbols other than data symbols. The times and carriers at which pilot symbols are inserted into modulated signal zl' are null symbols in modulated signal 22'. Conversely, the times and carriers at which pilot symbols are inserted into modulated signal 22' are null symbols in modulated signal zl '. [0253] Figs. 49A and 49B, like Fig. 6, indicate the arrangement of symbols when a change in phase is applied to precoded baseband signal 22' (while no change of phase is performed on precoded baseband signal zl). (Although Fig. 6 illustrates a change of phase with respect to the time domain, switching time t with carrier f in Fig. 6 corresponds to a change of phase with respect to the frequency domain. In other words, replacing (t) with (t, f) where t is time and f is frequency corresponds to performing a change of phase on time-fiequency blocks.) Accordingly, the numerical values indicated in Figs. 49A and 49B for each of the symbols are the values of precoded baseband signal 22' after a change of phase is performed. No values are given for the symbols of precoded baseband signal zl' (zl) as no change 5 of phase is performed thereon. [0254] The key point of Figs. 49A and 49B is that a change of phase is performed on the data symbols of precoded baseband signal 22', i.e., on precoded symbols. (The symbols under discussion, being precoded, actually include both symbols sl 10 and s2.) Accordingly, no change of phase is performed on the pilot symbols inserted into 22'. 1025 51 Figs. 50A and 50B illustrate the fiame configuration of modulated signals (precoded baseband signals) zl or zl' and 22' in the time-frequency domain. Fig. 15 50A illustrates the fi-me configuration of modulated signal (precoded baseband signal) zl or zl' while Fig. 50B illustrates the frame configuration of modulated signal (precoded baseband signal) 22'. In Figs. 50A and SOB, 4701 marks pilot symbols, 4702 marks data symbols, and 4901 marks null symbols for which the in-phase component of the baseband signal I = 0 and the quadrature component Q = 20 0. As such, data symbols 4702 are symbols on which precoding, or precoding and a change of phase, have been performed. Figs. 50A and 50B differ fiom Figs. 48A and 48B in the configuration scheme for symbols other than data symbols. The times and carriers at which pilot symbols are inserted into modulated signal zl' are null symbols in modulated signal 22'. Conversely, the times and carriers at which 25 pilot symbols are inserted into modulated signal 22' are null symbols in modulated signal 21'. [0256] Figs. 50A and 50B, like Fig. 26, indicate the arrangement of symbols when a change of phase is applied to precoded baseband signal zl' and to precoded baseband signal 22'. (Although Fig. 26 illustrates a change of phase with respect to the time domain, switching time t with carrier f in Fig. 26 corresponds to a change of 5 phase with respect to the frequency domain. In other words, replacing (t) with (t, f ) where t is time and f is frequency corresponds to performing a change of phase on time-frequency blocks.) Accordingly, the numerical values indicated in Figs. 50A and 50B for each of the symbols are the values of precoded baseband signal zl' and 22' after a change of phase. 10 [0257] The key point of Figs. 50A and 50B is that a change of phase is performed on the data symbols of precoded baseband signal zl', that is, on the precoded symbols thereof, and on the data symbols of precoded baseband signal 22', that is, on the precoded symbols thereof. (The symbols under discussion, being precoded, 15 actually include both symbols sl and s2.) Accordingly, no change of phase is performed on the pilot symbols inserted in zl', nor on the pilot symbols inserted in 22'. [0258] Fig. 51 ilIustrates a sample configuration of a transmission device 20 generating and transmitting modulated signal having the frame configuration of Figs. 47A, 47B, 49A, and 49B. Components thereof performing the same operations as those of Fig. 4 use the same reference symbols thereas. In Fig. 5 1, the weighting units 308A and 308B and phase changer 3 17B only operate at times indicated by the frame configuration signal 313 as 25 corresponding to data symbols. [0259] In Fig. 5 1, a pilot symbol generator 5 10 1 (that also generates null symbols) outputs baseband signals 5 102A and 5 102B for a pilot symbol whenever the frame configuration signal 3 13 indicates a pilot symbol (or a null symbol). [0260] 5 Although not indicated in the fi-ame configurations from Figs. 47A through 50B, when precoding (or phase rotation) is not performed, such as when transmitting a modulated signal using only one antenna (such that the other antenna transmits no signal) or when using a space-time coding transmission scheme (particularly, space-time block coding) to transmit control information symbols, then the fi-ame 10 configuration signal 313 takes control information symbols 5104 and control information 5 103 as input. When the frame configuration signal 3 13 indicates a control information symbol, baseband signals 5 102A and 5 102B thereof are output. [0261] Wireless units 310A and 310B of Fig. 5 1 take a plurality of baseband 15 signals as input and select a desired baseband signal according to the frame configuration signal 313. Wireless units 310A and 310B then apply OFDM signal processing and output modulated signals 31 1A and 31 1B conforming to the frame configuration. [0262] 20 Fig. 52 illustrates a sample configuration of a transmission device generating and transmitting modulated signal having the fiame configuration of Figs. 48A, 48B, 50A, and 50B. Components thereof performing the same operations as those of Figs. 4 and 51 use the same reference symbols thereas. Fig. 5 1 features an additional phase changer 317A that only operates when the frame configuration 25 signal 313 indicates a data symbol. At all other times, the operations are identical to those explained for Fig. 5 1. [0263] Fig. 53 illustrates a sample configuration of a transmission device that differs fiom that of Fig. 51. The following describes the points of difference. As shown in Fig. 53, phase changer 3 17B takes a plurality of baseband signals as input. Then, when the fi-ame configuration signal 313 indicates a data symbol, phase 5 changer 317B performs a change of phase on precoded baseband signal 316B. When frame configuration signal 313 indicates a pilot symbol (or null symbol) or a control information symbol, phase changer 317B pauses phase changing operations, such that the symbols of the baseband signal are output as-is. (This may be interpreted as performing forced rotation corresponding to do.) 10 A selector 5301 takes the plurality of baseband signals as input and selects a baseband signal having a symbol indicated by the frame configuration signal 3 13 for output. [0264] Fig. 54 illustrates a sample configuration of a transmission device that 15 differs from that of Fig. 52. The following describes the points of difference. As shown in Fig. 54, phase changer 317B takes a plurality of baseband signals as input. Then, when the fiame configuration signal 313 indicates a data symbol, phase changer 317B performs a change of phase on precoded baseband signal 316B. When frame configuration signal 313 indicates a pilot symbol (or null symbol) or a 20 control information symbol, phase changer 3 17B pauses phase changing operations such that the symbols of the baseband signal are output as-is. W s may be interpreted as performing forced rotation corresponding to Go.) Similarly, as shown in Fig. 54, phase changer 5201 takes a plurality of baseband signals as input. Then, when the frame configuration signal 3 13 indicates 25 a data symbol, phase changer 5201 performs a change of phase on precoded baseband signal 309A. When frame configuration signal 313 indicates a pilot symbol (or null symbol) or a control information symbol, phase changer 5201 pauses phase changing operations such that the symbols of the baseband signal are 76 output as-is. (This may be interpreted as performing forced rotation corresponding to SO.) The above explanations are given using pilot symbols, control symbols, and data symbols as examples. However, the present invention is not limited in this 5 manner. When symbols are transmitted using schemes other than preceding, such as single-antenna transmission or transmission using space-time block coding, not performing a change of phase is important. Conversely, performing a change of phase on symbols that have been precoded is the key point of the present invention. [0265] 10 Accordingly, a characteristic feature of the present invention is that the change of phase is not performed on all symbols within the fiame configuration in the time-frequency domain, but only performed on signals that have been precoded. [0266] [Embodiment 41 15 Embodiments 1 and 2, described above, discuss a regular change of phase. Embodiment 3, however, discloses performing a different change of phase on neighbouring symbols. [0267] The present Embodiment describes a phase changing scheme that varies 20 according to the modulation scheme and the coding rate of the error-correcting codes used by the transmission device. Table 1, below, is a list of phase changing scheme settings corresponding to the settings and parameters of the transmission device. LO2681 25 [Table 11 Phase Changing Pattern No. of Modulated Transmission Signals Modulation Scheme Coding Rate [0269] In Table 1, #1 denotes modulated signal sl fi-om Embodiment 1 described above (baseband signal sl modulated with the modulation scheme set by the 2 2 2 2 2 2 2 2 2 2 2 #I :QPSK, #2: QPSK #l:QPSK, #2: QPSK #l:QPSK, #2: QPSK #l:QPSK,#2:QPSK #I :QPSK, #2: QPSK #I: QPSK, #2: 16-QAM #1: QPSK, #2: 16-QAM #1: QPSK, #2: 16-QAM #I: QPSK, #2: 16-QAM #1: QPSK, #2: 16-QAM #1: 16-QAM, #2: 16-QAM #1: 112, #2 213 #1: 112, #2: 314 #1: 213, #2: 315 #1:2/3,#2: 213 #I: 313, #2: 213 #1: 112, #2: 213 #1: 112, #2: 314 #1: 112, #2: 315 #1: 213, #2: 314 #1: 213, #2: 516 #1: 1/2, #2: 213 #1: -, #2:A #1: A, #2: B #1: A, #2: C #I: C, #2: - #l: D, #2: E #1: B, #2: A #1: A, #2: C #1: -, #2:E #1: D, #2: - #1: D, #2: B #1: -, #2:E transmission device) and #2 denotes modulated signal s2 (baseband signal s2 modulated with the modulation scheme set by the transmission device). The coding rate column of Table 1 indicates the coding rate of the error-correcting codes for modulation schemes #1 and #2. The phase changing pattern column of Table 1 5 indicates the phase changing scheme applied to precoded baseband signals zl (zl') and 22 (227, as explained in Embodiments 1 through 3. Although the phase changing patterns are labeled A, B, C, D, E, and so on, this refers to the phase change degree applied, for example, in a phase changing pattern given by Math. 46 (formula 46) and Math. 47 (formula 47), above. In the phase changing pattern 10 column of Table 1, the dash signifies that no change of phase is applied. [0270] The combinations of modulation scheme and coding rate listed in Table 1 are examples. Other modulation schemes (such as 128-QAM and 256-QAM) and coding rates (such as 7/23) not listed in Table 1 may also be included. Also, as 15 described in Embodiment 1, the error-correcting codes used for sl and s2 may differ (Table 1 is given for cases where a single type of error-correcting codes is used, as in Fig. 4). Furthermore, the same modulation scheme and coding rate may be used with different phase changing patterns. The transmission device transmits information indicating the phase changing patterns to the reception device. The 20 reception device specifies the phase changing pattern by cross-referencing the information and Table 1, then performs demodulation and decoding. When the modulation scheme and error-correction scheme determine a unique phase changing pattern, then as long as the transmission device transmits the modulation scheme and information regarding the error-correction scheme, the reception device knows the 25 phase changing pattern by obtaining that information. As such, information pertaining to the phase changing pattern is not strictly necessary. [0271] In Embodiments 1 through 3, the change of phase is applied to precoded baseband signals. However, the amplitude may also be modified along with the phase in order to apply periodical, regular changes. Accordingly, an amplification modification pattern regularly modifying the amplitude of the modulated signals 5 may also be made to conform to Table 1. In such circumstances, the transmission device should include an amplification modifier that modifies the amplification after weighting unit 308A or weighting unit 308B fiom Fig. 3 or 4. In addition, amplification modification may be performed on only one of or on both of the precoded baseband signals zl(t) and z2(t) (in the former case, the amplification 10 modifier is only needed after one of weighting unit 308A and 308B). [0272] Furthermore, although not indicated in Table 1 above, the mapping scheme may also be regularly modified by the mapper, without a regular change of phase. That is, when the mapping scheme for modulated signal sl(t) is 16-QAM 15 and the mapping scheme for modulated signal s2(t) is also 16-QAM, the mapping scheme applied to modulated signal s2(t) may be regularly changed as follows: fiom 16-QAM to 16-APSK, to 16-QAM in the IQ plane, to a first mapping scheme producing a signal point layout unlike 16-APSK, to 16-QAM in the IQ plane, to a second mapping scheme producing a signal point layout unlike 16-APSK, and so on. 20 As such, the data reception quality can be improved for the reception device, much like the results obtained by a regular change of phase described above. [0273] In addition, the present invention may use any combination of schemes for a regular change of phase, mapping scheme, and amplitude, and the transmit signal 25 may transmit with all of these taken into consideration. LO2741 The present Embodiment may be realized using single-camer schemes as well as multi-carrier schemes. Accordingly, the present Embodiment may also be 80 realized using, for example, spread-spectrum communications, OFDM, SC-FDM, SC-OFDM, wavelet OFDM as described in Non-Patent Literature 7, and so on. As described above, the present Embodiment describes changing the phase, amplitude, and mapping schemes by performing phase, amplitude, and mapping scheme 5 modifications with respect to the time domain t. However, much like Embodiment 1, the same changes may be carried out with respect to the frequency domain. That is, considering the phase, amplitude, and mapping scheme modification in the time domain t described in the present Embodiment and replacing t with f (fbeing the ((sub-) carrier) fiequency) leads to phase, amplitude, and mapping scheme 10 modification applicable to the fiequency domain. Also, the phase, amplitude, and mapping scheme modification of the present Embodiment is also applicable to phase, amplitude, and mapping scheme modification in both the time domain and the frequency domain. LO2751 15 Furthermore, in the present Embodiment, symbols other than data symbols, such as pilot symbols (preamble, unique word, etc) or symbols transmitting control information, may be arranged within the fiame in any manner. [0276] Embodiment All 20 The present Embodiment describes a scheme for regularly changing the phase when encoding is performed using block codes as described in Non-Patent Literature 12 through 15, such as QC (Quasi-Cyclic) LDPC Codes (not only QC-LDPC but also LDPC codes may be used), concatenated LDPC and BCH (Bose-Chaudhuri-Hocquenghem) codes, Turbo codes or Duo-Binary Turbo Codes 25 using tail-biting, and so on. The following example considers a case where two streams sl and s2 are transmitted. However, when encoding has been performed using block codes and control information and the like is not required, the number of bits making up each coded block matches the number of bits making up each block 8 1 code (control information and so on described below may yet be included). When encoding has been performed using block codes or the like and control information or the like (e.g., CRC (cyclic redundancy check) transmission parameters) is required, then the number of bits making up each coded block is the sum of the 5 number of bits making up the block codes and the number of bits making up the information. [0277] Fig. 34 illustrates p e varying numbers of symbols and slots needed in each coded block when block codes are used. Fig. 34 illustrates the varying numbers of 10 symbols and slots needed in each coded block when block codes are used when, for example, two streams sl and s2 are transmitted as indicated by the transmission device from Fig. 4, and the transmission device has only one encoder. (Here, the transmission scheme may be any single-carrier scheme or multi-carrier scheme such as OFDM.) 15 As shown in Fig. 34, when block codes are used, there are 6000 bits making up a single coded block. In order to transmit these 6000 bits, the number of required symbols depends on the modulation scheme, being 3000 symbols for QPSK, 1500 symbols for 16-QAM, and 1000 symbols for 64-QAM. 102781 20 Then, given that the transmission device from Fig. 4 transmits two streams simultaneously, 1500 of the aforementioned 3000 symbols needed when the modulation scheme is QPSK are assigned to sl and the other 1500 symbols are assigned to s2. As such, 1500 slots for transmitting the 1500 symbols (hereinafter, slots) are required for each of sl and s2. 25 [0279] By the same reasoning, when the modulation scheme is 16-QAM, 750 slots are needed to transmit all of the bits making up a single coded block, and when the modulation scheme is 64-QAM, 500 slots are needed to transmit all of the bits making up a single coded block. [0280] The following describes the relationship between the above-defined slots 5 and the phase of multiplication, as pertains to schemes for a regular change of phase. Here, five different phase changing values (or phase changing sets) are assumed as having been prepared for use in the scheme for a regular change of phase. That is, five different phase changing values (or phase changing sets) have been prepared for the phase changer of the transmission device from Fig. 4 (equivalent to 10 the period (cycle) fkom Embodiments 1 through 4) (As in Fig. 6, five phase changing values are needed in order to perform a change of phase with a period (cycle) of five on precoded baseband signal 22' only. Also, as in Fig. 26, two phase changing values are needed for each slot in order to perform the change of phase on both precoded baseband signals zl' and 22'. These two phase changing values are 15 termed a phase changing set. Accordingly, five phase changing sets should ideally be prepared in order to perform the change of phase with a period (cycle) of five in such circumstances). These five phase changing values (or phase changing sets) are expressed as PHASE[O], PHASE[I], PHASE[2], PHASE[3], and PHASE[4]. [028 11 20 For the above-described 1500 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is QPSK, PHASE[O] is used on 300 slots, PHASE[l] is used on 300 slots, PHASE[2] is used on 300 slots, PHASE[3] is used on 300 slots, and PHASE[4] is used on 300 slots. This is due to the fact that any bias in phase usage causes great influence to be exerted by the more 25 frequently used phase, and that the reception device is dependent on such influence for data reception quality. LO2821 Similarly, for the above-described 700 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is 16-QAM, PHASE[O] is used on 150 slots, PHASE[l] is used on 150 slots, PHASE[2] is used on 150 slots, PHASE[3] is used on 150 slots, and PHASE[4] is used on 150 slots. 5 [0283] Furthermore, for the above-described 500 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is 64-QAM, PHASE[O] is used on 100 slots, PHASE[l] is used on 100 slots, PHASE[2] is used on 100 slots, PHASE[3] is used on 100 slots, and PHASE[4] is used on 100 slots. 10 [0284] As described above, a scheme for a regular change of phase requires the preparation of N phase changing values (or phase changing sets ) (where the N different phases are expressed as PHASE[O], PHASE[l], PHASE[2] ... PHASEP-21, PHASE[N-11). As such, in order to transmit all of the bits making 15 up a single coded block, PHASE[O] is used on & slots, PHASE[l] is used on K1 slots, PHASE[i] is used on Ki slots (where i = 0, 1,2 ... N-1 (i denotes an integer that satisfies 033-1)), and PHASE[N-I] is used on KN-I slots, such that Condition #A01 is met. 20 (Condition #A01) K,,=K1...=Ki=...KN-I. Thatis,K,=&(Vaand%wherea,b,=O, 1,2 ... N-1 (a denotes an integer that satisfies 0-a-1, b denotes an integer that satisfies 05bSN-1), a f b). [0285] 25 Then, when a communication system that supports multiple modulation schemes selects one such supported modulation scheme for use, Condition #A01 is preferably satisfied for the supported modulation scheme. [0286] 84 However, when multiple modulation schemes are supported, each such modulation scheme typically uses symbols transmitting a different number of bits per symbols (though some may happen to use the same number), Condition #A01 may not be satisfied for some modulation schemes. In such a case, the following 5 condition applies instead of Condition #AOl. (Condition #A02) The difference between K, and & satisfies 0 or 1. That is, [I(, - satisfies 0 or 1 (Va, Vb, where a, b = 0, 1, 2 ... N-1 (a denotes an integer that satisfies 0 3 9 - 1 , b denotes an integer that satisfies 0 3 9 - 1 ) , a # b) 10 Fig. 35 illustrates the varying numbers of symbols and slots needed in two coded blocks when block codes are used. Fig. 35 illustrates the varying numbers of symbols and slots needed in each coded block when block codes are used when, for example, two streams sl and s2 are transmitted as indicated by the transmission device from Fig. 3 and Fig. 12, and the transmission device has two encoders. (Here, 15 the transmission scheme may be any single-carrier scheme or multi-carrier scheme such as OFDM.) As shown in Fig. 35, when block codes are used, there are 6000 bits making up a single coded block. In order to transmit these 6000 bits, the number of required symbols depends on the modulation scheme, being 3000 symbols for QPSK, 20 1500 symbols for 16-QAM, and 1000 symbols for 64-QAM. [0287] The transmission device fiom Fig. 3 and the transmission device fiom Fig. 12 each transmit two streams at once, and have two encoders. As such, the two streams each transmit different code blocks. Accordingly, when the modulation 25 scheme is QPSK, two coded blocks drawn from sl and s2 are transmitted within the same interval, e.g., a first coded block drawn from sl is transmitted, then a second coded block drawn from s2 is transmitted. As such, 3000 slots are needed in order to transmit the first and second coded blocks. 85 1028 81 By the same reasoning, when the modulation scheme is 16-QAM, 1500 slots are needed to transmit all of the bits making up the two coded blocks, and when the modulation scheme is 64-QAM, 1000 slots are needed to transmit all of 5 the bits making up the two coded blocks. [0289] The following describes the relationship between the above-defined slots and the phase of multiplication, as pertains to schemes for a regular change of phase. Here, five different phase changing values (or phase changing sets) are 10 assumed as having been prepared for use in the scheme for a regular change of phase. That is, five different phase changing values (or phase changing sets) have been prepared for the phase changers of the transmission devices fiom Figs. 3 and 12 (equivalent to the period (cycle) fiom Embodiments 1 through 4) (As in Fig. 6, five phase changing values are needed in order to perform a change of phase having a 15 period (cycle) of five on precoded baseband signal 22' only. Also, as in Fig. 26, two phase changing values are needed for each slot in order to perform the change of phase on both precoded baseband signals zl' and 22'. These two phase changing values are termed a phase changing set. Accordingly, five phase changing sets should ideally be prepared in order to perform the change of phase with a period 20 (cycle) of five in such circumstances). These five phase changing values (or phase changing sets) are expressed as PHASE[O], PHASE[l], PHASE[2], PHASE[3], and PHASE[4]. [0290] For the above-described 3000 slots needed to transmit the 6000x2 bits 25 making up a single coded block when the modulation scheme is QPSK, PHASE[O] is used on 600 slots, PHASE[l] is used on 600 slots, PHASE[2] is used on 600 slots, PHASE[3] is used on 600 slots, and PHASE[4] is used on 600 slots. This is due to the fact that any bias in phase usage causes great influence to be exerted by the more frequently used phase, and that the reception device is dependent on such influence for data reception quality. [0291] Furthermore, in order to transmit the first coded block, PHASE[O] is used 5 on slots 600 times, PHASE[l] is used on slots 600 times, PHASE[2] is used on slots 600 times, PHASE[3] is used on slots 600 times, and PHASE[4] is used on slots 600 times. Furthermore, in order to transmit the second coded block, PHASE[O] is used on slots 600 times, PHASE[l] is used on slots 600 times, PHASE[2] is used on slots 600 times, PHASE[3] is used on slots 600 times, and PHASE[4] is used on 10 slots 600 times. [0292] Similarly, for the above-described 1500 slots needed to transmit the 6000x2 bits making up the two coded blocks when the modulation scheme is 16-QAM, PHASE[O] is used on 300 slots, PHASE[l] is used on 300 slots, PHASE[2] is used 15 on 300 slots, PHASE[3] is used on 300 slots, and PHASE[4] is used on 300 slots. LO2931 Furthermore, in order to transmit the first coded block, PHASE[O] is used on slots 300 times, PHASE[l] is used on slots 300 times, PHASE[2] is used on slots 300 times, PHASE[3] is used on slots 300 times, and PHASE[4] is used on slots 300 20 times. Furthermore, in order to transmit the second coded block, PHASE[O] is used on slots 300 times, PHASE[l] is used on slots 300 times, PHASE[2] is used on slots 300 times, PHASE[3] is used on slots 300 times, and PHASE[4] is used on slots 300 times. 25 Similarly, for the above-described 1000 slots needed to transmit the 6000x2 bits making up the two coded blocks when the modulation scheme is 64-QAM, PHASE[O] is used on 200 slots, PHASE[l] is used on 200 slots, PHASE[2] is used on 200 slots, PHASE[3] is used on 200 slots, and PHASE[4] is used on 200 slots. 87 [0295] Furthermore, in order to transmit the first coded block, PHASE[O] is used on slots 200 times, PHASE[l] is used on slots 200 times, PHASE[2] is used on slots 200 times, PHASE[3] is used on slots 200 times, and PHASE[4] is used on slots 200 5 times. Furthermore, in order to transmit the second coded block, PHASE[O] is used on slots 200 times, PHASE[l] is used on slots 200 times, PHASE[2] is used on slots 200 times, PHASE[3] is used on slots 200 times, and PHASE[4] is used on slots 200 times. [0296] 10 As described above, a scheme for regularly changing the phase requires the preparation of phase changing values (or phase changing sets ) expressed as PHASE[O], PHASE[l 1, PHASE[2] . . . PHASEP-21, PHASEP-I]. As such, in order to transmit all of the bits making up two coded blocks, PHASE[O] is used on & slots, PHASE[l] is used on K1 slots, PHASE[i] is used on Ki slots (where i = 0, 1, 15 2...N-1 (i denotes an integer that satisfies OSilN-1), and PHASE[N-11 is used on slots, such that Condition #A03 is met. (Condition #A03) &=K ,... =Ki=...KN-l. Thatis,K,=&,(VaandVbwherea,b,=0,1,2 ... N-1 20 (a denotes an integer that satisfies OSaIN-1, b denotes an integer that satisfies OeSN-1), a # b). Further, in order to transmit all of the bits making up the fist coded block, PHASE[O] is used &,l times, PHASE[l] is used times, PHASE[i] is used 25 times (where i = 0, 1, 2...N-l(i denotes an integer that satisfies 033-1), and PHASE[N-11 is used times, such that Condition #A04 is met. (Condition #A04) 88 &,l=K1,l = ... Ki,l = ... KN-,,i. Thati~,K,~=&(,V~a andVbwherea,b,=O, 1, 2 ... N-1 (a denotes an integer that satisfies 0 9 3 - 1 , b denotes an integer that satisfies O35N-1), a # b). 5 Furthermore, in order to transmit all of the bits making up the second coded block, PHASE[O] is used b2 times, PHASE[l] is used KI2 times, PHASE[i] is used Ki2 times (where i = 0, 1, 2 ... N-1 (i denotes an integer that satisfies 099-1), and PHASEIN-11 is used KN-,,2 times, such that Condition #A05 is met. (Condition #A05) 10 GJ = K12 = ... Ki,2 = ... That is, = (Va and Vb where a, b, = 0, 1, 2 ... N-1 (a denotes an integer that satisfies O%-