[Document name] SPECIFICATION
[Title of the Invention] Switching Power Supply Circuit
[What is Claimed is:]
[Claim 1]
A switching power supply circuit, comprising:
rectifier smoothing means for receiving a commercial ac power supply as an input and producing and outputting a rectified smoothed DC voltage;
switching means having a switching element and intended for receiving and switchably outputting the DC voltage;
a self-excited oscillation drive circuit forming said switching means and intended for switchably driving said switching element in a self-excited manner;
an insulating converter transformer for transmitting the output of said switching means to a secondary winding side of said switching power supply circuit;
a primary winding side resonance circuit formed at least from a leakage inductance component including a primary winding of said insulating converter transformer and a capacitance of a resonance capacitor, said primary winding side resonance circuit operating said switching means in a resonance type operation;
a secondary winding side resonance circuit formed on the secondary winding side of said switching power supply circuit, including a leakage inductance component of a secondary winding of said insulating converter transformer anci a capacitance of a secondary side resonance capacitor;
03/01/2001
ORIGINAL
IN/PCT/2001/00014/MUM

DC output voltage production means, including said secondary side resonance circuit, for receiving an alternating voltage obtained at said secondary winding of said insulating converter transformer as an input thereto and performing a full-wave rectification operation for the input alternating voltage to produce a secondary winding side DC output voltage;
switching frequency variation means for varying an inductance of said self-excited oscillation drive circuit in response to a control current level supplied thereto to vary a switching frequency of said switching element;
constant voltage control means for varying the control current in response to the level of the secondary winding side DC output voltage and supplying the varied control current to said switching frequency variation means to variably control the switching frequency thereby to effect constant voltage control for the secondary side DC output voltage; and
switching frequency control means for receiving a start-up power supply obtained upon, or immediately after start up of said switching power supply circuit as an input thereto and for supplying a control current of a predetermined level o\ er a predetermined period of time after start-up in place of said constant voltage control means to control the switching frequency so that the switching frequency remains within a predetermined range. [Claim 2]
The switching power supply circuit according to claim 1, further comprising:
2

a tertiary winding wound at a position on said primary winding of said insulating converter transformer spaced by more than a predetermined physical distance from said secondary winding; and
a rectifier smoothing circuit connected to said tertiary winding,
the start-up power supply which is supplied to said switching frequency control means being a DC voltage obtained by said rectifier smoothing circuit. [Claim 3]
The switching power supply circuit as claimed in claim 1, further comprising:
standby power supply means for receiving the commercial ac power supply as an input thereto and producing a DC voltage as a standby power supply voltage from the input commercial ac power supply, and switching start-up means for making use of the standby power supply voltage to produce a switching start-up signal which can be used to start said switching means; and
the switching start-up signal being used as the start-up power supply to be supplied to said switching frequency control means. [Claim 4]
The switching power supply circuit according to claim 1, further comprising operation period setting means for variably setting an operation period of said switching frequency control means. [Detailed Description of the Invention] [0001]
3

[Technical Field to which the Invention pertains]
This invention relates to a switching power supply circuit which can be incorporated as a power supply in various electronic apparatus. [0002] [Prior Art]
A switching power supply circuit which adopts a switching converter in the form of, for example, a flyback converter or a forward converter is widely known. Because switching converters of these types employ a rectangular waveform signal to control a switching operation, these switching converters are limited in the amount of switching noise they can suppress. It is also known that these switching converters also are limited in power conversion efficiency because of their operation characteristics.
Thus, various switching power supply circuits that employ resonance type converters have been proposed by the present applicant. A resonance type converter is advantageous in that a high power conversion efficiency can be readily obtained while maintaining low noise characteristics because the waveform controlling the switching operation is a sine waveform. This resonance type converter is also advantageous in that it can be simply formed from a comparatively small number of parts. [0003]
FIG. 7 shows as a prior art an example of a switching power supply circuit
4

constructed based on the invention which has been previously proposed by the present applicant. This power supply circuit includes a voltage resonance type converter including a single switching element Ql that performs a switching operation in a self-excited manner in accordance with a single end system. [0004]
The switching power supply circuit illustrated in this drawing includes a rectifier smoothing circuit for receiving a commercial ac power supply (ac input voltage VAC) and producing a DC input voltage. The rectifier smoothing circuit is formed as a full-wave rectifier circuit, comprising a bridge rectifier circuit Di and a smoothing capacitor Ci. The rectifier smoothing circuit produces a rectified smoothed voltage Ei of a level equal to the ac input voltage VAC. Further, an inrush current limitation resistor Ri is interposed in a rectifier current path of the rectifier smoothing circuit in order to suppress any initial inrush current spike from flowing into smoothing capacitor, for example, when an initial power supply is provided to the circuit.
Further, in the drawing an AC switch SW is interposed in the commercial ac power supply line. AC switch SW is switched on/off to start/stop flow of power to power the illustrated supply circuit. [0005]
A voltage resonance type switching converter which is provided in the power supply circuit has a self-exciting construction including switching element Ql. In this
5

instance, the switching element Ql may be formed of a bipolar transistor (BJT:
junction transistor) having a high voltage withstanding property.
[0006]
The base of switching element Ql is connected to the positive electrode side of smoothing capacitor Ci (rectified smoothed voltage Ei) through a starting resistor RS so that the base current upon start-up of the circuit may be obtained from the rectifier smoothing circuit. Further, a resonance circuit adapted to be driven in a self-excited oscillation state is connected between the base of switching element Q I and a primary side ground. The resonance circuit is formed from the series circuit connection of an inductor LB, a detection driving winding NB, a resonance capacitor CB, and a base current limiting resistor RB.
A damper diode DD is interposed between the base of the switching element Ql and the negative electrode (set at a primary side ground) of smoothing capacitor Ci and forms a path for damper current which flows when the switching element QI is switched off. The collector of switching element Ql is connected to an end oi a primary winding Nl of an insulating converter transformer PIT. The emitter of switching element Ql is grounded. [0007]
A parallel resonance capacitor Cr is connected in parallel between the collecto r and the emitter of switching element Ql. Parallel resonance capacitor Cr forms. based on a capacitance thereof and a leakage inductance LI of primary winding N 1
6

of insulating converter transformer PIT, a primary side parallel resonance voltage resonance type converter circuit. Although detailed description is omitted here, when the switching clement Ql is off, a voltage resonance type operation is obtained by an action of the parallel resonance circuit which causes the voltage Vcp across resonance capacitor Cr to actually exhibit a pulse wave of a sine waveform. [0008]
An orthogonal control transformer PRT shown in this drawing is a saturatabie reactor on which a detection winding ND, a drive winding NB and a control winding NC are wound. Orthogonal control transformer PRT is provided for driving switching element Ql and controlling an output voltage to be constant.
Though not shown, orthogonal control transformer PRT is formed with a structure wherein a three dimensional core is formed such that two double channel-shaped cores each having four magnetic legs are joined to each other at the ends of the magnetic legs thereof. Detection winding ND and Drive winding NB are wound in the same winding direction around two predetermined ones of the magneiic legs of the three dimensional core, and the control winding NC is wound around two predetermined ones of the magnetic legs of the three dimensional core such that the winding direction thereof is orthogonal to detection winding ND and the drive winding NB. [0009]
Detection winding ND of orthogonal control transformer PRT (frequency
7

variation means) is interposed in series between the positive electrode of smoothing capacitor Ci and primary winding Nl of insulating converter transformer PIT so that a switching output of the switching element Ql is transmitted to detection winding ND through primary winding Nl. In orthogonal control transformer PRT, the switching output obtained in detection winding ND is excited in driving winding NB via transformer coupling, and consequently, an alternating drive voltage is generated in driving winding NB. The drive voltage is output as drive current from the series resonance circuit (NB and CB), which forms the self-excited oscillation drive circuit. to the base of switching element Ql through base current limiting resistor RB. Consequently, switching element Ql performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit (NB and CB). [0010]
Insulating converter transformer PIT transmits a switching output of the switching element Ql to the secondary side thereof.
As shown in FIG. 8, the insulating converter transformer PIT includes an EE-shaped core which includes a pair of E-shaped cores CR1 and CR2 made of, for example, a ferrite material and coupled with each other such that magnetic legs thereof are opposed to each other. A primary winding Nl and a secondary winding N2 are wound separately from each other on the central magnetic legs of the EE-shaped core using a split bobbin B. As seen from FIG. 8, a gap G is formed
8

between the central magnetic legs of the EE-shaped core. Consequently, a loose coupling having a required coupling coefficient can be obtained.
The gap G can be formed by providing the central magnetic legs of the E-shaped cores CR1 and CR2 shorter than the other two outer magnetic legs. The coupling coefficient k in this instance is, for example, k = 0 .85 which is a coupling coefficient of a loose coupling. Consequently, a saturation condition is less likely to be obtained as much. [0011]
One end of primary winding Nl of insulating converter transformer PIT is connected to the collector of the switching element Ql. The other end of primary winding Nl is connected to the positive electrode of smoothing capacitor Ci (rectified smoothed voltage Ei) through a series connection of detection winding ND as shown in the drawing. [0012]
On the secondary side of insulating converter transformer PIT, an alternating voltage induced by primary winding Nl appears in secondary winding N2. A secondary side parallel resonance capacitor C2 is connected in parallel to secondary winding N2. A parallel resonance circuit is therefore formed from a leakage inductance L2 of secondary winding N2 and a capacitance of secondary side parallel resonance capacitor C2. The alternating voltage induced in secondary winding N2 is converted into a resonance voltage by the parallel resonance circuit. In short, a
9

voltage resonance operation is obtained on the secondary side. [0013]
Thus, in the power supply circuit, a parallel resonance circuit for generating a voltage resonance type switching operation is provided on the primary side of insulated converter transformer PIT, and a parallel resonance circuit for obtaining a full-wave rectification operation (voltage resonance operation) is provided on the secondary side of insulated converter transformer PIT. It is to be noted that, in the present specification, a switching converter of a construction which includes resonance circuits for both of the primary side and the secondary side in this manner is referred to as "composite resonance type switching converter". [0014]
In the parallel resonance circuit on the secondary side of insulated converter transformer PIT formed as described above, center taps are provided for secondary winding N2, and rectifier diodes DOl, D02, D03 and D04 and smoothing capacitois C01 and C02 are connected in such a manner as shown in the drawing to provide two full-wave rectifier circuits; a first full-wave rectifier circuit [including rectifier diodes DOl and D02 and smoothing capacitor C01], and a second full-wave rectifier circuit [including rectifier diodes D03 and D04 and smoothing capacitor C02],
The first full-wave rectifier circuit composed of [rectifier diodes DOl and D02 and smoothing capacitor C01] receives a resonance voltage supplied from the secondary side parallel resonance circuit and produces a DC output voltage EOl. The
10

second full-wave rectifier circuit composed of [rectifier diodes D03 and D04 and smoothing capacitor C02] similarly receives the resonance voltage supplied from the secondary side parallel resonance circuit and produces a DC output voltage E02.
The DC output voltage E01 and the DC output voltage E02 are also separately input to a control circuit 1. Control circuit 1 utilizes the DC output voltage E01 as a detection voltage and utilizes the DC output voltage E02 as an operation power supply therefor. [0015]
In insulating converter transformer PIT, the mutual inductance between the inductance LI of primary winding Nl and the inductance L2 of secondary winding N2 may have a value +M or a value -M depending upon the relationship between the polarities (winding directions) of primary winding Nl and secondary winding N 2 and the connection of rectifier diodes DO (D01 and D02/ D03 and D04).
For example, if the components are connected in a configuration as shown in FIG. 9(a), then the mutual inductance is +M. If the components are connected in a configuration as shown in FIG. 9(b), then the mutual inductance is -M.
If this is examined in connection with operation of the secondary side of the circuit, for example, the operation that rectified current flows through the rectifier diode D01 (D03) when the alternating voltage obtained at the secondary winding N2 has the positive polarity can be regarded as an operation mode of +M (forward mode). On the contrary, the operation that rectified current flows through the rectifier diode
11

D02 (D04) when the alternating voltage obtained at the secondary winding N2 has the negative polarity can be regarded as an operation mode of -M (flyback mode). In other words, the power supply circuit operates in the +M/-M mode of the mutual inductance each time the alternating voltage obtained at the secondary winding becomes positive/negative. [0016]
In the power supply circuit having the construction described above, power output from the secondary winding side parallel resonance circuit may be increased based upon the supply of power thereto. In this instance, for example, if a full-wave rectifier circuit is connected to the secondary winding side parallel resonance circuit as in the circuit shown in the drawing, rectified current flows alternately in both of the +M/-M operation modes of the mutual inductance as described above. In other words, a rectified output from the circuit is obtained within both of the periods within which the alternating voltage is positive and negative. By this described operation. supplied power increases, and also the rate of reaching the maximum load power increases. [0017]
The construction for obtaining the full-wave rectification operation illustrated in FIG. 9 is realized by forming gap G in insulating converter transformer PIT as previously described with reference to FIG. 8 to obtain a loose coupling of a predetermined coupling coefficient to establish a condition wherein the insulating
12

converter transformer is less liable to reach a saturation condition. For example. where gap G is not provided in the insulating converter transformer PIT, there is .1 high degree of possibility that, upon performance of flyback operation, insulating converter transformer PIT may be put into a saturation condition and operate abnormally. The full-wave rectification operation described above is therefore less likely to be properly performed in such a saturation condition. [0018]
Control circuit 1 varies the control current (DC current) level supplied to control winding NC in response to the level of the DC output voltage (E01) of the secondary winding side. This supplied current level is adjusted to variably control the inductance LB of drive winding NB wound on the orthogonal control transformer PRT in a feedback control loop-type setup. Consequently, the resonance condition of the series resonance circuit formed of inductance LB of the drive winding NB varies. This variation in turn varies the switching frequency of the switching element Ql as hereinafter described with reference to FIG. 8, and the secondary winding side DC output voltage is stabilized by the variation of the switching frequency of switching element Ql. [0019]
In the power supply circuit shown, in order to vary the switching frequencv, the period within which switching element Ql is off is fixed whereas the period within which switching element Ql is on is variably controlled. While employing a
13

constant voltage control, the control operation for the power supply circuit operates to variably control the switching frequency of switching element Ql to perform resonance impedance control for the switching output, and simultaneously performs continuity angle control (PWM control) of the switching element in a switching period. This composite control operation is realized with a single control circuit system.
Here, the switching frequency control is performed such that, when the secondary side output voltage rises as a result of, for example, decreasing of the load thereon, the switching frequency is raised to suppress the secondary winding side output power. [0020] [Problems that the Invention is to solve]
During use, AC switch SW in power supply circuit shown in FIG. 7 is switched from an off-state to an on-state. Charging current then flows into smoothing capacitor Ci through inrush current limiting resistor Ri and bridge rectifier circuit I)i so that charging of smoothing capacitor Ci is performed. This charging continues until the rectified smoothed voltage Ei which is a voltage across smoothing capacitor Ci rises to a level corresponding to the ac input voltage level. Then, when starting current is supplied from the rectified smoothed voltage Ei to the base of switching element Ql through starting resistor RS, switching element Ql is turned on to begin oscillation thereof. Thereafter, the switching element Ql performs a switching
14

operation according to the oscillation thereof. [0021]
When this process begins, and Ql is in the on state, the secondary winding side of insulated converter transformer PIT has a low impedance. An excessively high charging current flows into smoothing capacitors CO (COl and C02) in which no charge has yet been accumulated. In other words, the secondary winding side DC output voltage is in a transient state until a steady level is obtained, and in this state, control of the switching frequency by the control circuit 1 is not performed. At this time, the switching element Ql performs a switching operation with the lowest switching frequency which depends upon the time constant of the self-excited oscillation drive circuit (CB and NB). [0022]
Therefore, upon startup, the period of switching element Ql becomes long in accordance with operation of the PWM control described above. The resonance voltage pulse Vcp generated by switching element Ql within a period within which the switching element Ql is off therefore becomes excessively high. Consequent]}, because of excessive inrush current flowing to smoothing capacitors CO (COl. and C02) on the secondary winding side of insulated converter transformer PIT, excessive collector current Icp flows to switching element Ql in the primary side from the smoothing capacitor Ci through detection winding ND and primary winding Nl. [0023]
15

Because a resonance voltage pulse Vcp and collector current Icp of excessively high levels are generated in such a manner upon startup of the circuit, switching element Ql must be selected to have a high voltage withstanding property and a high current flow capacity to withstand them. A switching element having such characteristics is expensive and large in size, and therefore becomes an obstacle to reducing the cost and miniaturizing the size of the circuit. [0024]
Therefore, it is also a common practice to provide an over current limiting circuit in order to allow use of a switching element having a comparatively low voltage withstanding property and current capacity. A construction of a power supply circuit of this type is shown in FIG. 10. It is to be noted that in FIG. 10, like reference characters to those of FIG. 7 denote like elements, and description thereof is omitted herein to avoid redundancy. [0025]
The circuit shown in FIG. 10 includes an over current limiting circuit 10 in addition to the circuit described hereinabove with reference to FIG. 7.
Over current limiting circuit 10 includes a current detection resistor RE and a series connection circuit of voltage dividing resistors Rll and R12 interposed in parallel to each other between the emitter of switching element Ql and the primary winding side ground. The base of a transistor Q2, used for conduction control, is connected to a junction between voltage dividing resistors Rll and R12.
16

The collector of transistor Q2 is connected to the base of switching element Q1 through a diode D10. The anode of the diode D10 is connected to the base of switching element Ql. The cathode of diode D10 is connected to the collector of transistor Q2. The emitter of transistor Q2 is grounded to the primary winding side ground.
A capacitor C10, for example, for noise absorption, is connected in parallel between the collector and the emitter of transistor Q2. [0026]
In the power supply circuit having the construction shown, collector current Icp which is generated upon start-up flows from the collector through the emitter of switching element Ql and is detected by current detection resistor RE and the series connection of voltage dividing resistors Rll and R12. A current level corresponding to a voltage dividing ratio of voltage dividing resistors Rll and R12 flows to the base of transistor Q2. When the level of the collector current Icp of transistor switching element Ql becomes higher than a certain predetermined level and the base current amount of transistor Q2 also becomes higher than a certain predetermined level, transistor Q2 is rendered conductive thereby to render diode D10 conductive.
When diode D10 and transistor Q2 are conductive, forward current flowing to the base of transistor Q2 flows to the primary winding side ground through diode D10 and transistor Q2 (collector - emitter). In short, the collector current Icp is limited by limiting excessive base current to transistor Q2 upon start-up.
17

[0027]
While excessive current can be limited in such a manner, the power supply circuit still has the following problems.
First, because current detection resistor RE is connected in series to the emitter of switching element Ql, when a load placed thereon is heavy, high power loss occurs and the power conversion efficiency deteriorates.
Further, when the power supply circuit shown in FIG. 10 is to be actually designed, a margin must be provided against a malfunction when the ac input voltage VAC rises high or against a malfunction which arises from a variation of some component of the over current limiting circuit, even upon steady operation, so that the malfunction does not occur. [0028]
FIG. 11 shows waveform diagrams illustrating operation of the power supply circuit of FIG. 10 which is designed to include a margin against a malfunction.
Particularly, FIGS. 11(a) and 11(b) show waveforms of the resonance voltage Vcp and the collector current Icp, respectively, when the maximum load power Pomax is 150 W and the ac input voltage VAC is 120 V. FIGS. 11(c) and 11(d) show waveforms of the resonance voltage Vcp and the collector current Icp, respectively. in a steady state condition. [0029]
The switching frequency of switching element Ql is controlled so as to be
18

lower in a maximum load power operation condition than in a steady state operation. Switching element Ql is thus controlled such that, while the period TOFF within which the switching element Ql is off is fixed, the period TON within which the switching element Ql is on increases. Actually, in the maximum load power operation condition illustrated in FIGS. 11(a) and 11(b), the period TOFF is 2 microseconds, and the period TON is 9 microseconds. [0030]
As will be understood from the drawing, the resonance voltage Vcp is 700 Vp (FIG. 11(c)) during steady state operation, but rises up to 900 Vp (FIG. 11(a)) in a maximum load power operation condition. Meanwhile, the collector current Icp is 4.5 Ap (FIG. 11(d)) in steady state operation, but rises up to 6.5 Ap (FIG. ll(c )) in a maximum load power operation condition.
The resonance voltage Vcp and the collector current Icp in a maximum load power operation condition exhibit a rise of approximately 20% to 30% from the levels exhibited in steady state operation. Thus, switching element Ql must be able to withstand the voltage of 900 Vp. Actually, however, a device having a voltage withstanding property against 1,200 V is selected for the switching element Ql. A device having a voltage withstanding property against 1,200 V is comparatively expensive and large in size.
In short, even if an over current limiting circuit, such as that shown in FIG. 10 is provided to suppress a possible application of too much current upon the
19

application of a heavy load, a device having a corresponding high voltage withstanding property must be selected for the switching element. Consequently, the circuit cannot be reduced in cost or size. [0031]
Further, because the circuit construction shown in FIG. 10 does not involve particular control for a rise of the secondary side DC voltage upon start-up of the circuit, the rise time of the secondary side DC output voltage level is correspondingly short. Therefore, the power supply circuit has a problem also in that a margin against a malfunction of the circuit upon startup cannot be provided. [0032] [Means for solving the Problems]
In order to solve the problems mentioned above the present invention provides an improved switching power supply circuit having a construction as follows.
Generally speaking, an improved switching power supply circuit according to the invention comprises rectifier smoothing means for receiving a commercial ac power supply as an input thereto, producing a rectified smoothed voltage from the inputted commercial ac power supply and outputting the rectified smoothed voltage as a DC input voltage, switching means including a switching element for receiving and switchably outputting the DC input voltage, a self-excited oscillation drive circui t constructing the switching means and intended for switchably driving the switching element in a self-excited manner, an insulating converter transformer for transmitting
20

the output of the switching element to a secondary winding side of the circuit, a primary winding side resonance circuit formed from at least a leakage inductance component including a primary winding of the insulating converter transformer and a capacitance of a resonance capacitor for providing resonance type operation of the switching element.
Further on the secondary winding side of the circuit there are provided a secondary winding side resonance circuit formed from the leakage inductance component of the secondary winding of the insulating converter transformer and the capacitance of a secondary winding side resonance capacitor, and DC output voltage producer means including the secondary winding side resonance circuit, for receiving an alternating voltage obtained at the secondary winding of the insulating converter transformer as an input thereto, and for performing a full-wave rectification operation for the input alternating voltage to produce a secondary winding side DC output voltage.
A switching frequency varying element is provided for varying an inductance of the self-excited oscillation drive circuit in response to a control current level supplied thereto to vary a switching frequency of the switching element. A constant voltage controller for varying the control current in response to the level of the secondary winding side DC output voltage and supplying the varied control current to the switching frequency variation means to variably control the switching frequency thereby to effect constant voltage control for the secondary side DC output
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voltage in a feedback control type setup is also provided.
Furthermore, a switching frequency controller is provided for receiving a start-up power supply upon or immediately after start-up of the switching power supply circuit as an input thereto, and for supplying the control current of a predetermined level over a predetermined period of time after start-up in place of a control current supplied by the constant voltage controller to control the switching frequency so that the switching frequency remains within a predetermined desired range. [0033]
For example, in a self-excited composite resonance switching converter, within a transient period until the secondary winding side DC output voltage is stabilized at its steady-state level after start-up of the power supply, it is not possible to control the switching frequency appropriately to stabilize the same using the standard switching control used at steady-state.
Therefore, the switching frequency controller operates upon start-up or immediately after start-up of the power supply circuit so that the switching frequency thereof remains within a predetermined required range. In this manner, the switching frequency is controlled so that the secondary winding side DC output voltage level may be suppressed as in the switching power supply circuit of the present invention. Thus, a sudden rise of the secondary side DC output voltage upon start-up ss suppressed, and a soft start-up is realized.
22

[0034] [Embodiments]
FIG. 1 illustrates a power supply circuit constructed in accordance with one embodiment of the invention. It is to be noted that in FIG. 1, like reference characters to those of FIG. 7 denote like elements, and description thereof is omitted herein to avoid redundancy. [0035]
Now, in FIG. 1 there is shown a concrete circuit arrangement of a control circuit 1. Control circuit 1 includes a pair of resistors R3 and R4 connected in series between a DC output voltage E01 and a secondary winding side ground. A shunt regulator Q3 is provided with a control terminal connected to a junction (voltage dividing point) between resistors R3 and R4. The anode of shunt regulator Q3 is grounded, and the cathode of shunt regulator Q3 is connected to a line of a DC voltage E3 through a control winding NC of an orthogonal control transformer PRT, described below.
The cathode of shunt regulator Q3 is connected to the junction between resistors R3 and R4 through a series connection of a capacitor Cll and a resistor R2. Further, a series connection circuit of a capacitor C3 and a resistor R5 is connected in parallel to resistor R4. [0036]
Control circuit 1 having the connection configuration mentioned above
23

functions as an error amplifier which receives DC output voltage E01 as a detection input thereto. In particular, a voltage obtained by voltage division of the DC output voltage E01 by means of resistors R3 and R4 is input to the control terminal of shunt regulator Q3. Accordingly, shunt regulator Q3 supplies current of a level corresponding to the DC output voltage E01 to control winding NC. The level of the control current to flow through control winding NC is therefore variably controlled. Accordingly, the switching frequency of switching element Ql is varied to execute constant voltage control as described above with reference to FIG. 7. [0037]
Power supply circuit illustrated further includes a soft start-up circuit 2 and a rectifier circuit for producing a DC voltage E3 for driving the soft start-up circuit 2.
A construction of the rectifier circuit for producing the DC voltage E3 will be first described.
The rectifier circuit includes a tertiary winding N3 wound on an insulating converter transformer PIT, a diode D6, and a smoothing capacitor C6 for half-wave rectifying an alternating voltage excited in tertiary winding N3. Accordingly, DC voltage E3 is obtained across smoothing capacitor C6. A resistor R7 is connected in parallel to smoothing capacitor C6. DC voltage E3 across smoothing capacitor C6 is supplied as a detection voltage (operating power supply) to soft start-up circuit 2 through a Zener diode D7. Zener diode D7 is set, for example, in combination with resistor R7 connected in parallel to smoothing capacitor C6, so as to become
24

conducting with a reverse voltage corresponding to 18 V. [0038]
FIG. 2 illustrates a structure of an insulating converter transformer PIT on which tertiary winding N3 is wound. In FIG. 2 like reference characters to those of FIG. 8 denote like elements, and description thereof is omitted herein to avoid redundancy.
Portions of the insulating converter transformer PIT of the embodiment shown in FIG. 2 are constructed similarly to those in insulating converter transformer PIT described above in reference to FIG. 8. Thus, a gap G is formed between the middle magnetic legs thereof so that a loose coupling having a coupling coefficient of, for example, approximately 0.85 may be obtained. Also, the mounting structures of primary winding Nl and secondary winding N2 are similar to those in FIG. 8.
However, in the insulating converter transformer PIT shown in FIG. 2 tertiary winding N3 is additionally wound, for example, as shown in FIG. 2, at a position on the outer side (upper side) of a split bobbin B adjacent the primary winding Nl and farthest from the secondary winding N2.
Alternatively, tertiary winding N3 may be wound at a position on the center side (lower side) of the split bobbin B farthest from the secondary winding N2 as indicated as (N3) in FIG. 2. For tertiary winding N3, for example, a triple insulating line is adopted.
Tertiary winding N3 is wound at a position farthest from secondary winding
25

N2 so that the coupling degree of tertiary winding N3 to secondary winding N2 is minimized. If too much coupling is present, the level of DC voltage E3 upon start-up of the power supply may exceed a steady voltage as will be described below making reference to FIG. 4. Only if a coupling degree below a particular level can be obtained can the tertiary winding N3 be wound at a particular position, for example. nearer to secondary winding N2 than the position shown in FIG. 2. [0039]
Soft start-up circuit 2 shown in FIG. 1 has the following construction.
A pair of resistors R6 and R5 are connected between the anode side of Zencr diode D7 to which the DC voltage E3 is applied and the secondary winding side ground. The base of a transistor Q5 (NPN) is connected to the junction between resistors R6 and R5.
The collector of transistor Q5 is connected to the base of a transistor Q4 through a resistor R7. The emitter of transistor Q5 is grounded. Further, a capacitor C4 for adjusting the base potential applied to transistor Q4 is interposed in parallel between the base of transistor Q4 (PNP) and the secondary winding side ground. The emitter of transistor Q4 is connected to a junction between control winding NC of orthogonal control transformer PRT and the cathode of shunt regulator Q3. The collector of transistor Q4 is connected to the secondary winding side ground. [0040]
Operation of soft start-up circuit 2 will now be described making reference to
26

FIGS. 3 and 4.
FIGS. 3(a) and 3(b) illustrate a resonance voltage Vcp and a parallel collector current Icp corresponding, for example, to a switching operation of switching elemen t Ql approximately 20 ms after an AC switch SW is switched on.
FIG. 4 illustrates level variations of the secondary side DC output voltage HO (E01) and the DC voltage E3 upon start-up. In FIG. 4, the x-axis indicates the time-after AC switch SW is switched on and the y-axis indicates the voltage level. [0041]
After AC switch SW is switched on, DC voltage E3 rises as shown in FIG. 4 and approximately after the lapse of 20 ms, a level (18 V or more) higher than 15 V which is to be the resulting steady state voltage level, is obtained. This operation arises from the tertiary winding N3 wound in such a manner as described above in FIG. 2. [0042]
When DC voltage E3 becomes 18 V or more as described above, the Zenei diode D7 is rendered conducting. Consequently, DC voltage E3 is divided by voltage dividing resistors R6 and R5 in soft start-up circuit 2, and current of a level corresponding to the divided voltage level flows to the base of transistor Q5. [0043]
Consequently, transistor Q5 is rendered conducting, and collector current of transistor Q5 flows as base current to the base of transistor Q4 through the resistor
27

R7. Consequently, transistor Q4 is rendered conductive between the emitter and the
collector thereof.
[0044]
At this time, secondary winding side DC output voltage E0 is in a transition period within which it rises toward 135 V which is the steady state level thereof as shown in FIG. 4. Control circuit 1 is not operating during this time. [0045]
Accordingly, when approximately 20 ms elapses after AC switch SW is switched on, emitter current of transistor Q4 of soft start-up circuit 2 flows to control winding NC of orthogonal control transformer PRT.
Control circuit 1 of power supply circuit of the illustrated embodiment controls so that the switching frequency rises as the level of the control current flowing through control winding NC increases. Accordingly, as the emitter current of transistor Q4 of soft start-up circuit 2 flows to control winding NC, control current of a sufficiently high level flows to control winding NC. [0046]
The switching frequency of switching element Ql is controlled so that approximately 20 ms after AC switch SW is switched on, the switching frequency is adjusted, for example, up to approximately 250 KHz which is the highest frequency within a predetermined allowable range of switching frequency variation. [0047]
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Operation of switching element Ql 20 ms after AC switch SW is switched on is illustrated in the waveform diagrams of FIG. 3.
In particular, as can be seen from the waveforms of the resonance voltage Vcp of FIG. 3(a) and the collector current Icp of FIG. 3(b), switching element Ql performs a switching operation in a cycle of the period TOFF = period TON = 2 microseconds.
Because the switching operation is performed with a high switching frequencv (250 KHz) in this manner, the resonance voltage Vcp obtained when the switching element Ql is off (within the period TOFF) is controlled to be approximately to 500 Vp as seen from FIG. 3(a), and the parallel collector current Icp obtained when switching element Ql is on (within the period TON) is 2 Ap. [0048]
Thereafter, charging of capacitor C4 connected to the base of transistor Q4 is performed in accordance with a time constant provided by the capacitance thereof and a DC resistance component of control winding NC. This acts to raise the base potential of transistor Q4. As the base current level of transistor Q4 drops, the control current (emitter current) flowing through the control winding NC decreases and the switching frequency of switching element Ql gradually drops. [0049]
In short, the power supply circuit of the illustrated embodiment performs control so that approximately 20 ms after AC switch SW is switched on by operation
29

of soft start-up circuit 2, the switching frequency is raised compulsorily, and thereafter, the switching frequency is dropped gradually.
Consequently, as can be recognized from comparison of the variations of the secondary side DC output voltage EO by the power supply circuit of FIG. 1 according to the present invention indicated by a solid line in FIG. 4 and the power supply circuit of FIG. 10 indicated by a broken line in FIG. 4, the power supply circuit of FIG. 1 operates so that the time until the secondary side DC output voltage EO rises to its steady-state level (135 V) upon start-up is longer. [0050]
From FIG. 4, it can be seen that power supply circuit of FIG. 1 is set so that the rise time of the secondary winding side DC output voltage EO from power supply circuit of FIG. 1 is approximately five times the rise time of the power supply circuit of FIG. 5.
In particular, while the power supply circuit of FIG. 10 reaches its steady-state level at approximately 20 ms after the AC switch SW is switched on, power supply circuit of FIG. 10 that is the embodiment of the present invention is controlled so that its steady-state level is reached approximately 100 ms after AC switch SW is switched on. [0051]
Because control circuit 1 operates properly when secondary winding side DC output voltage E0 returns to its steady level approximately 100 ms after AC switch
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SW is switched on, the switching frequency control operation for the switching element Ql is passed from soft start-up circuit 2 to control circuit 1. Further, at this point of time, in such a manner as shown in FIG. 4, because DC voltage E3 returns to 15 V, its steady level, Zener diode D7 is rendered non-conducting. Accordingly, operation of soft start-up circuit 2 is also stopped. [0052]
In the power supply circuit according to the embodiment of the invention, as described above a soft start-up operation upon start-up of a power supply is realized. According to this soft start-up operation, the resonance voltage Vcp upon start-up is controlled to 500 Vp with the power supply circuit of FIG. 1, while in a conventional power supply circuit, the resonance voltage Vcp upon start-up is 900 Vp to 1,000 Vp with the power supply circuit of FIG. 10 as shown in the waveforms of FIG. 3. Also collector current level upon start-up is controlled to be 2 Ap according to the power supply circuit of FIG. 1 of the invention while this level may rise to 6.5 Ap with the power supply circuit of FIG. 10.
Consequently, a device having a voltage withstanding property of only 700 V and having a comparatively small current capacity can be employed for switching element Ql of the power supply circuit of FIG. 1. An inexpensive device of a comparatively small size can therefore be produced. Additionally, the capacitor Cr on the primary winding side of the circuit can employ a device able to withstand only approximately 800 V. Size and cost of the device can be also reduced.
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[0053]
In the power supply circuit of the illustrated embodiment, the time for transition after a soft start-up operation until reaching steady state operation depends upon the time constant provided by capacitor C4 in soft start-up circuit 2, and control winding NC. The time for transition after a soft start-up operation until reaching a steady-state operation can be set arbitrarily by adjusting at least one of, for example, the capacitance of capacitor C4 and the DC resistance component (turn number) of control winding NC. Accordingly, a malfunction margin upon start-up can be increased by adjustment of the time constant. Because the DC resistance component (turn number) of control winding NC is principally set depending upon a control characteristic of the orthogonal control transformer PRT, preferably the capacitance of capacitor C4 is varied. [0054]
FIG. 5 shows another power supply circuit constructed in accordance with a second embodiment of the invention.
In FIG. 5, like reference characters to those of FIG. 1 denote like elements, and description thereof is omitted herein.
The illustrated power supply circuit includes the power supply circuit described in reference to FIG. 1 as a main power supply circuit, and additionally includes a standby power supply circuit. Power supply circuit shown in FIG. 5 is therefore practically more advantageous than power supply circuit of FIG. 1.
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[0055]
Standby power supply circuit 4 of FIG. 5 includes a standby transformer SBT which also receives ac input voltage Vac on the primary side thereof.
Standby transformer SBT includes a secondary winding, a pair of recti Tier diodes Dll and D12 and a smoothing capacitor C03 connected in such a manner as shown in FIG. 5 to form a center tap type full-wave rectifier circuit. Standby transformer SBT produces a rectified smoothed voltage of 7 V using ac input voltage Vac input to the primary side thereof.
The rectified smoothed voltage of 7 V is supplied to a regulator RG. The regulator RG stabilizes the input voltage of 7 V and outputs a stabilized voltage of 5 V. The stabilized DC voltage of 5 V is obtained as a voltage across a smoothing capacitor C04. The DC voltage of 5 V is supplied as a standby power supply voltage and as an operation power supply, for example, for a microcomputer, an IC for reception of infrared rays and so forth included in the apparatus. DC voltage of 5 V is supplied also as an operation power supply for a relay drive circuit 3. [0056]
Relay drive circuit 3 is provided to control an on/off operation of AC sw itch SW with an on/off control signal for the main power supply circuit output under the control of, for example, the microcomputer. To this end, an electromagnetic relay RY of the relay drive circuit 3 operates in association with an on/off operation of AC switch SW.
33

[0057]
In power supply circuit, the on/off control signal for the main power supply circuit is a DC voltage having a level higher than a predetermined level to drive relay drive circuit 3 and soft start-up circuit 2 in such a manner as described below. Then, when the main power supply circuit is to be turned on, the DC voltage mentioned above is output, but when the main power supply circuit is to be turned off, the output of the DC voltage is stopped.
The DC voltage as the on/off control signal for the main power supply circuit is obtained, though not shown in FIG, by the microcomputer making use of the DC voltage of 5 V produced by standby power supply circuit 4. [0058]
In relay drive circuit 3, a pair of voltage dividing resistors R21 and R22 are interposed between the line for the on/off control signal for the main power supply circuit and the secondary winding side ground. The base of a transistor Q2 is connected to a junction between voltage dividing resistors R21 and R22. A capacitor CIO is connected between the base of the transistor Q2 and the secondary winding side ground.
The collector of transistor Q2 is connected to the DC voltage line of 5 V obtained from standby power supply circuit 4 through electromagnetic relay RY. The emitter of transistor Q2 is grounded to the secondary winding side ground A protecting diode D8 is connected in a direction shown in FIG. 5 to electromagnetic
34

relay RY. [0059]
Further, in the illustrated power supply circuit, the on/off control signal for the main power supply circuit is input as a detection voltage to soft start-up circuit 2 In soft start-up circuit 2 shown in the second embodiment, the DC voltage level as the on/off control signal is divided by the voltage dividing resistors R6 and R5 to perform detection thereby to effect conduction control of transistors Q5 and Q4.
Accordingly, in the second embodiment, the tertiary winding N3 used in the first embodiment is not required. Consequently, the insulating converter transformer PIT of FIG. 8 can be used as the insulating converter transformer in power supply circuit. [0060]
A diode D5 shown in FIG. forms a path from the power supply for operating control circuit 1 upon soft start-up. The anode of diode D5 is connected to the DC voltage line of 5 V of the standby power supply circuit 4. The cathode of diode D5 is connected to the cathode of shunt regulator Q3 through control winding NC.
A diode D4 forms a path from the power supply for operating control circuit 1 during steady operation after soft start-up. The anode of diode D4 is connected to the line for the secondary winding side DC output voltage E02 of the main power supply circuit while the cathode of diode D4 is connected to the cathode of shunt regulator Q3 through control winding NC.
35

[0061]
In the power supply circuit, if a DC voltage is output as the on/off control signal for turning the main power supply on, then transistor Q2 of relay drive circuit 3 is rendered conductive, thereby to drive electromagnetic relay RY. Consequently. the AC switch SW is switched on, and power supply to the main power supply side is started.
The DC voltage as the on/off control signal is input also to the series connection circuit of resistors R6 and R5 of soft start-up circuit 2. Consequently, transistors Q5 and Q4 are rendered conducting, and control current flows to the control winding NC as described above.
The switching frequency of switching element Ql is controlled so that it rises substantially up to the highest frequency level (250 KHz) within an allowable variation control range in a similar manner as in the first embodiment described above. Thus, secondary side DC output voltages EO (EOl and E02) do not reach their steady levels too quickly with a steep slope of increase. [0062]
The controlling condition in this instance is shown in FIG. 6. Level variations of the secondary winding side DC output voltages E0 (DC output voltage E01 and E02) upon start-up are shown. In FIG. 6, the x-axis indicates the time after AC switch SW is switched on and the y-axis indicates the voltage level.
As is seen from FIG. 6, in the power supply circuit of the second embodiment
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shown in FIG. 5, a time of approximately 100 ms is required after AC switch SW is switched on until the DC output voltages E01 and E02 reach their respective steady-state levels (here, DC output voltage E01 = 135 V and E02 = 15 V). It is extended to substantially five times as compared with the circuit shown in FIG. 10.
An operation similar to that described above with reference to the waveform diagrams of FIGS. 3A and 3B is obtained as the switching operation of switching element Ql immediately after startup. Accordingly, in the circuit of the second embodiment a device able to withstanding a voltage of 700 V and having a comparatively small current capacity can be employed for switching element 01. Also a device able to withstand a voltage of approximately 800 V can be selectively used for parallel resonance capacitor Cr. Further, by adjusting at least one of the capacitance of capacitor C4 and the DC resistance component (turn number) of control winding NC, a malfunction margin upon starting of power supply can be set similarly as in the power supply circuit of the first embodiment. [0063]
The switching power supply circuit of the present invention is not limited to the specific forms described hereinabove but may have various forms. For example, detailed constructions of the control circuit 1, soft start-up circuit 2, relay drive circuit 3 and so forth of the power supply apparatus described hereinabove may be modified suitably.
Further, while also a push-pull type voltage resonance converter wherein two
37

switching elements are turned on/off alternately is known as a voltage resonance converter, the present invention can be applied also to a self-excited voltage resonance converter which employs the push-pull system. [0064]
Furthermore, while also a composite resonance switching converter wherein a resonance capacitor is connected in series to the secondary winding of insulating converter transformer PIT to form a secondary side series resonance circuit has been proposed by the present applicant, the present invention can be applied also 10 a composite switching converter of the type just described. [0065]
Further, while a voltage resonance converter is used in the power supply circuit described hereinabove in the illustrated embodiments, the present invention can be applied also to a power supply circuit in which a current resonance converter wherein stabilization is performed by self-excited switching frequency control is provided on the primary side. [0066] [Advantages of the Invention]
As described above, in a switching power supply circuit according to the present invention a composite resonance type switching converter for driving a switching element by means of self-excited oscillation includes a soft starting circuit (switching frequency control means) by which upon starting of power supply, the
38

switching frequency of the switching element is raised to a frequency substantially near to an upper limit of variable width as a desired switching frequency range, whereby an abrupt raising of a secondary side DC output can be suppressed.
As a result, the levels of a resonance voltage applied to the switching element and a switching output current (collector current) fed through the switching element when it starts its switching operation are suppressed. It is therefore possible to select a switching element having a lower withstanding voltage characteristic and a comparatively small current capacity
In such an arrangement, it is not necessary that the switching element is provided with a circuit for limiting any over current and thus a power loss which may occurs in a current limiting circuit can be avoided so that a power conversion efficiency is improved. [0067]
A start-up power supply for operating the soft starting circuit is obtained by DC voltage which is produced by a rectifying an alterenating voltage generated on tertiary winding wound at a position on the primary winding of the insulating converter transformer physically farthest from the secondary winding for example. Therefore, the power supply circuit having the soft starting circuit according to the present invention can operate the soft starting circuit using a single independent arrangement. [0068]
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Furthermore, standby power supply voltage can be used as a power supply source for operating the soft starting circuit in which an on/off control signal for the main power supply can be used. In this case, the provision of the tertiary winding is not required and a conventional insulating converter transformer having a simple construction may be suitably used. [0069]
According to the present invention, the period during which the soft starting circuit is operated is determined by the time constant provided by the capacitance of capacitor connected to the base of transistor in the soft starting circuit and the resistance component of the control winding to which a control current is fed. In other words, the operation time of the soft starting circuit can be set arbitrarily by adjusting the capacitance of capacitor or the resistance component of control winding. That is, operation time setting means is provided for arbitrarily setting the operation time of the soft starting circuit.
Accordingly, if the operation time of the soft starting circuit is set arbitrarily in such a manner, it is possible to suitably set the time until the secondary output voltage rises to its steady-state level. This has an advantage that malfunction margin upon start-up can be increased. [Brief Description of the Drawings]
[FIG. 1]
A circuit diagram showing a power supply circuit constructed in accordance
40

with the first embodiment of the invention.
[FIG. 2]
A side elevational sectional view showing a structure of an insulating converter transformer of the invention provided in the power supply circuit of the first embodiment.
[FIG. 3]
Waveform diagrams illustrating a switching operation of the power supply circuit of the first embodiment.
[FIG. 4]
An explanation view showing a soft start-up operation of the power supply circuit of the first embodiment of the invention as compared with operation upon start-up of a conventional power supply circuit.
[FIG. 5]
A circuit diagram showing a construction of a power supply circuit according to the first embodiment of the invention.
[FIG. 6]
A diagram view comparing a soft start-up operation of the power supply circuit of the first embodiment of the invention and operation upon start-up of a conventional power supply circuit.
[FIG. 7]
A circuit diagram showing a construction of a standard power supply circuit.
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[FIG. 8]
A sectional view showing a structure of an insulating converter transformer of the conventional power supply circuit and the power supply circuit of the present invention.
[FIG. 9]
Circuit diagrams illustrating operations when the mutual inductance is +M and -M.
[FIG. 10]
A circuit diagram showing a construction of a standard power supply circuit which includes an over current limitation circuit.
[FIG. 11]
Waveform diagrams illustrating switching operation of the standard power supply circuit shown in FIG. 10.
[Explanation of Reference Characters]
1 : Control circuit
2 : Soft starting circuit
3 : Relay drive circuit
Ci : Smoothing capacitor
C2 : Secondary side parallel resonance capacitor Cs : Secondary side series resonance capacitor Di: Bridge rectifier circuit
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