1. A switching power supply circuit comprising: rectifying and smoothing means (Di, Ci) for obtaining a direct-current input voltage by rectifying and smoothing an alternating input voltage; switching means formed with a switching device (Ql) supplied with said direct-current input voltage and performing switching operation; switching-driving means (2) for switching-driving said switching device; an isolated converter transformer (PIT) formed by at least winding a primary winding (Nl) supplied with a switching output obtained by the switching operation of said switching means and a secondary winding (N2) in which an alternating voltage is induced by the switching output obtained in said primary wining (Nl); a primary side parallel resonant circuit formed by a leakage inductance component (LI) of the primary winding (Nl) of said isolated converter transformer (PIT) and a capacitance of a primary side parallel resonant capacitor (Cr), the primary side parallel resonant circuit converting operation of said switching means into a voltage resonant type operation; a secondary side parallel resonant circuit formed by a leakage inductance component (L2) of said secondary winding of said isolated converter transformer (PIT) and a capacitance of a secondary side parallel resonant capacitor (C2) connected in parallel with the secondary winding (N2); secondary side direct-current output voltage generating means for generating a secondary side direct-current output voltage by performing a rectifying operation on the alternating voltage induced in the secondary winding (N2) of said isolated converter transformer (PIT) and input to the secondary side direct-current output voltage generating means and smoothing a rectified output resulting from the rectifying operation by a secondary side smoothing capacitor (Co); and constant-voltage control means (1) for performing constant- voltage control on said secondary side direct-current output voltage by varying switching frequency of said switching means by controlling said switching-driving means according to level of said secondary side direct-current output voltage; characterized in that a coupling coefficient fog loose coupling between a primary side and a secondary side of said isolated converter transformer (PIT) is set such that an electromagnetic coupling type resonant circuit formed with said primary side parallel resonant circuit and said secondary side parallel resonant circuit has a unimodal characteristic as characteristic of output in response to an input of a frequency signal having said switching frequency, and a resonant frequency of said primary side parallel resonant circuit and a resonant frequency of said secondary side parallel resonant circuit are set so as to obtain power conversion efficiency at a certain level and higher under predetermined load conditions at least.

The present invention relates to a switching power supply circuit.
The present invention contains subject matter related to Japanese Patent Application JP 2005-028798 filed in the Japanese Patent Office on February 4,.2005, the entire contents of which being incorporated herein, by reference.
BACKGROUND OF THE INVENTION
The present invention relates to a switching power supply circuit including a voltage resonant converter.
As so-called soft-switching power supply of a resonant type, a current resonant type and a voltage resonant type are widely known. In a present situation, a current resonant converter having two switching devices coupled by a half-bridge coupling system is in wide use because such a current resonant converter is easily put to practical use.
However, the characteristics o.f a high withstand voltage switching device, for example, are now being improved, and therefore the problem of withstand voltage in putting a voltage resonant converter to practical use is being cleared up. In addition, a voltage resonant

converter formed by a single-ended system with one
switching device is known to be advantageous as compared
with a current resonant forward converter having one
switching device in terms of input feedback noise, the
noise component of a direct-current output voltage line,
and the like.
FIG. 12 shows an example of configuration of a
switching power supply circuit including a voltage
resonant converter of the single-ended system.
In the switching power supply circuit shown in FIG.
12, a rectifying and smoothing circuit formed by a bridge
rectifier circuit Di and a smoothing capacitor Ci
rectifies and smoothes an alternating input voltage VAC,
and thereby generates a rectified and smoothed voltage Ei
as a voltage across the smoothing capacitor Ci.
Incidentally, a noise filter formed by a set of
common mode choke coils CMC and two across capacitors CL
and removing common-mode noise is provided in the line of
a commercial alternating-current power supply AC.
The rectified and smoothed voltage Ei is input as a
direct-current input voltage to the voltage resonant
converter. As described above, the voltage resonant
converter employs the single-ended system with one
switching device Ql. The voltage resonant converter in
this case is an externally excited converter. The MOS-FET
switching device Ql is switching-driven by an oscillation
and drive circuit 2.
A MOS-FET body diode DD is connected in parallel
with the switching device Ql. A primary-side parallel
resonant capacitor Cr is connected in parallel with the
source and drain of the switching device Ql.
The primary-side parallel resonant capacitor Cr
forms a primary side parallel resonant circuit (voltage
resonant circuit) in conjunction with the leakage
inductance LI of a primary winding Nl of an isolated
converter transformer PIT. This primary side parallel
resonant circuit provides a voltage resonant operation as
the switching operation of the switching device Ql.
The oscillation and drive circuit 2 applies a gate
voltage as a drive signal to the gate of the switching
device Ql to switching-drive the switching device Ql.
Thus the switching device Ql performs switching operation
at a switching frequency corresponding to the cycle of
the drive signal.
The isolated converter transformer PIT transmits
the switching output of the switching device Ql to a
secondary side.
The isolated converter transformer PIT has for
example an EE type core formed by combining E-type cores
of ferrite material with each other. A winding part is
divided into a primary side winding part and a secondary
side winding part. The primary winding Nl and a secondary
winding N2 are wound around the central magnetic leg of
the EE type core.
In addition, a gap of about 1.0 mm is formed in the
central magnetic leg of the EE type core of the isolated
converter transformer PIT. Thereby a coupling coefficient
k = about 0.80 to 0.85 is obtained between the primary
side and the secondary side. The coupling coefficient k
at this level may be considered to represent loose
coupling, and therefore a state of saturation is not
easily obtained. The value of the coupling coefficient k
is a factor in setting the leakage inductance (LI).
One end of the primary winding Nl of the isolated
converter transformer PIT is inserted between the
switching device Ql and the positive electrode terminal
of the smoothing capacitor Ci. Thereby, the switching
output of the switching device Ql is transmitted to the
primary winding Nl. An alternating voltage induced by the
primary winding Nl occurs in the secondary winding N2 of
the isolated converter transformer PIT.
In this case, a secondary side parallel resonant
capacitor C2 is connected in parallel with the secondary
winding N2. Thus, the leakage inductance L2 of the
secondary winding N2 and the capacitance of the secondary
side parallel resonant capacitor C2 form a secondary side
parallel resonant circuit (voltage resonant circuit) .
In addition, a half-wave rectifier circuit is
formed by connecting a rectifier diode Dol and a
smoothing capacitor Co to the secondary side parallel
resonant circuit as shown in the figure. This half-wave
rectifier circuit generates a secondary side directcurrent
output voltage Eo having a level corresponding to
once an alternating voltage V2 obtained in the secondary
winding N2 (secondary side parallel resonant circuit) as
a voltage across the smoothing capacitor Co. The
secondary side direct-current output voltage Eo is
supplied to a load, and is also input to a control
circuit 1 as a detection voltage for constant-voltage
control.
The control circuit 1 inputs a detection output
obtained by detecting the level of the secondary side
direct-current output voltage Eo input as the detection
voltage to an oscillation and drive circuit 2.
The oscillation and drive circuit 2 controls the
switching operation of the switching device Ql according
to the level of the secondary side direct-current output
voltage Eo which level is indicated by the detection
output input to the oscillation and drive circuit 2 so as
to make the secondary side direct-current output voltage
Eo constant at a predetermined level. That is, the
oscillation and drive circuit 2 generates and outputs a
drive signal for controlling the switching operation.
Thereby control is performed to stabilize the secondary
side direct-current output voltage Eo.
FIGS. 13A and 13B and FIG. 14 show results of
experiments on the power supply circuit having the
configuration shown in FIG. 12. In conducting the
experiments, principal parts of the power supply circuit
of FIG. 12 are set as follows as conditions of VAC = 100
V corresponding to an AC 100 V system.
For the isolated converter transformer PIT, an EER-
35 core is selected, and the gap of the central magnetic
leg is set to a gap length of 1 mm. As for the respective
numbers of turns of the primary winding Nl and the
secondary winding N2, Nl = 43 T and N2 = 43 T. As for the
coupling coefficient k of the isolated converter
transformer PIT, k = 0.81 is set.
The primary-side parallel resonant capacitor Cr =
6800 pF and the secondary side parallel resonant
capacitor C2 = 0.01 pF are selected. Accordingly, the
resonant frequency fol = 175 kHz of the primary side
parallel resonant circuit and the resonant frequency fo2
= 164 kHz of the secondary side parallel resonant circuit
are set.
The rated level of the secondary side directcurrent
output voltage Eo is 135 V. Load power handled by
the power supply circuit is in a range of maximum load
power Pomax = 200 W to minimum load power Pomin = 0 W.
FIGS. 13A and 13B are waveform charts showing the
operations of principal parts in the power supply circuit
shown in FIG. 12 on the basis of the switching cycle of
the switching device Ql. FIG. 13A shows a switching
voltage VI, a switching current IQ1, a primary winding
current II, a secondary winding voltage V2, a secondary
winding current 12, and a secondary side rectified
current ID1 at the maximum load power Pomax = 200 W. FIG.
13B shows the switching voltage VI, the switching current
IQ1, the primary winding current II, the secondary
winding voltage V2, the secondary winding current 12, and
the secondary side rectified current IDl at the minimum
load power Pomin = 0 W.
The switching voltage VI is a voltage obtained
across the switching device Ql. The voltage VI is at a
zero level in a period TON in which the switching device
Ql is on, and forms a resonant pulse having a sinusoidal
waveform in a period TOFF in which the switching device
Ql is off. The resonant pulse waveform of the voltage VI
indicates that the operation of the primary side
switching converter is voltage resonant type operation.
The switching current IQ1 flows through the
switching device Ql (and the body diode DD). The
switching current IQ1 is at a zero level in the period
TOFF. In the period TON, the switching current IQ1 flows
through the body diode DD in a forward direction and is
thus of negative polarity at a turn-on time, and is
thereafter inverted to flow through the drain and the
source of the switching device Ql and increase until
turn-off timing. Therefore, the switching current IQ1 has
a peak level in the turn-off timing.
The primary winding current II flowing through the
primary winding Nl is obtained by combining a current
component flowing as the switching current IQ1 in the
period TON with a current flowing through the primaryside
parallel resonant capacitor Cr in the period TOFF.
The primary winding current II has a waveform as shown in
the figures.
As for the operation of a secondary side rectifier
circuit, at the maximum load power Pomax = 200 W, the
rectified current IDl flowing through the rectifier diode
Dol has a peak level at the time of turning on the
rectifier diode Dol, and thereafter decreases to a zero
level in a waveform as shown in FIG. 13A. The rectified
current IDl is at the zero level during the off period of
the rectifier diode Dol. Incidentally, at the minimum
load power Pomin = 0 W, the rectified current IDl is at
the zero level even during the on period.
The secondary winding voltage V2 in this case is
obtained in the parallel circuit of the secondary winding
N2 and the secondary side parallel resonant capacitor C2.
The secondary winding voltage V2 is clamped at the
secondary side direct-current output voltage Eo during
the on period during which the secondary side rectifier
diode Dol conducts. During the off period of the
secondary side rectifier diode Dol, the secondary winding
voltage V2 forms a sinusoidal waveform in a direction of
negative polarity. The secondary winding current 12
flowing through the secondary winding N2 is obtained by
combining the rectified current IDl with a current
flowing through the secondary side parallel resonant
circuit (N2 (L2)//C2). The secondary winding current 12
flows in waveforms as shown in FIGS. 13A and 13B, for
example.
FIG. 14 shows switching frequency fs, the on period
TON and the off period TOFF of the switching device Ql,
and AC—>DC power conversion efficiency ( T? AC—>DC) with
respect to load variation in the power supply circuit
shown in FIG. 12.
The AC-^DC power conversion efficiency ( rj AC~>DC) is
90% or more in a range of load power Po = 100 W to 200 W.
It is known that the single-ended system in which the
voltage resonant converter has one switching device Ql,
in particular, provides favorable power conversion
efficiency.
The switching frequency fs, the on period TON, and
the off period TOFF shown in FIG. 14 represent switching
operation as characteristics of constant-voltage control
against load variation in the power supply circuit of FIG.
12. In this case, the switching frequency fs is
controlled to be raised as the load becomes lighter. As
for the on period TON and the off period TOFF, the off
period TOFF is substantially constant irrespective of
load variation, whereas the on period TON is shortened as
the load becomes lighter. That is, the power supply
circuit shown in FIG. 12 variably controls the switching
frequency in such a manner as to reduce the on period TON
as the load becomes lighter, for example, while keeping
the off period TOFF constant.
By thus variably controlling the switching
frequency, inductive impedance obtained by providing the
primary side parallel resonant circuit and the secondary
side parallel resonant circuit is varied. By varying the
inductive impedance, an amount of power transmitted from
the primary side to the secondary side and an amount of
power transmitted from the secondary side parallel
resonant circuit to the load are changed, so that the
level of the secondary side direct-current output voltage
Eo is changed. The secondary side direct-current output
voltage Eo is thereby stabilized.
FIG. 15 schematically shows the constant-voltage
control characteristics of the power supply circuit shown
in FIG. 12 by relation between the switching frequency fs
(kHz) and the secondary side direct-current output
voltage Eo.
Letting fol be the resonant frequency of the
primary side parallel resonant circuit and fo2 be the
resonant frequency of the secondary side parallel
resonant circuit, in the circuit of FIG. 12, the
secondary side parallel resonance frequency fo2 is lower
than the primary side parallel resonance frequency fol,
as described above.
As for constant-voltage control characteristics
with respect to the switching frequency fs under a
condition of a constant alternating input voltage VAC, as
shown in FIG. 15, characteristic curves A and B
respectively represent constant-voltage control
characteristics at the maximum load power Pomax and the
minimum load power Pomin under the resonant impedance
corresponding to the resonant frequency fol of the
primary side parallel resonant circuit, and
characteristic curves C and D respectively represent
constant-voltage control characteristics at the maximum
load power Pomax and the minimum load power Pomin under
the resonant impedance corresponding to the resonant
frequency fo2 of the secondary side parallel resonant
circuit.
Further, when the primary side parallel resonant
circuit and the secondary side parallel resonant circuit
are provided as in the circuit of FIG. 12, there is an
intermediate resonant frequency fo between the resonant
frequencies fol and fo2. A characteristic curve E
represents a resonant impedance characteristic based on
relation between the intermediate resonant frequency fo
and the switching frequency fs at the maximum load power
Pomax. A characteristic curve F represents a resonant
impedance characteristic based on the relation between
the intermediate resonant frequency fo and the switching
frequency fs at the minimum load power Pomin.
With the voltage resonant converter provided with
the secondary side parallel resonant circuit, the level
of the secondary side direct-current output voltage Eo is
determined by the resonant impedance characteristics of
the intermediate resonant frequency fo in relation to the
switching frequency fs. The voltage resonant converter
shown in FIG. 12 employs a so-called lower side control
system in which the switching frequency fs is variably
controlled in a frequency region lower than the
intermediate resonant frequency fo.
A variable range (necessary control range) of the
switching frequency fs which range is necessary to
achieve constant voltage with the rated level (135 V in
the case of the circuit of FIG. 12) of the secondary side
direct-current output voltage Eo as a target value by
switching frequency control corresponding to lower side
control under the characteristics represented as the
characteristic curves E and F corresponding to the
intermediate resonant frequency fo in FIG. 15 is a
section indicated by Afs. In other words, by varying the
switching frequency to a required value according to load
variation in the frequency range corresponding to the
section indicated by Afs, the secondary side directcurrent
output voltage Eo is controlled to be at the
rated level tg.
For details, reference should be made to Japanese
Patent Laid-open No. 2000-152617.
SUMMARY OF THE INVENTION
With diversification of various electronic
apparatuses, there is a demand for a so-called wide range
capability that enables the power supply circuit to
operate dealing with the commercial alternating-current
power supply input of either of an AC 100 V system and an
AC 200 V system.
As described above, the power supply circuit having
the configuration shown in FIG. 12 operates so as to
stabilize the secondary side direct-current output
voltage Eo by switching frequency control, and the
variable range (necessary control range) of the switching
frequency fs which range is necessary to stabilize the
secondary side direct-current output voltage Eo is
indicated by Afs as described with reference to FIG. 15.
The power supply circuit shown in FIG. 12 deals
with load variation in a relatively wide range of 200 W
to 0 W. The actual necessary control range of the power
supply circuit of FIG. 12 meeting this load condition is
Afs = 96.7 kHz, which is a relatively wide range, with fs
= 117.6 kHz to 208.3 kHz.
The level of the secondary side direct-current
output voltage Eo is varied by change in the level of the
alternating input voltage VAC, of course. That is, as the
level of the alternating input voltage VAC is increased
or decreased, the level of the secondary side directcurrent
output voltage Eo is similarly increased or
decreased.
It can thus be said that the level of the secondary
side direct-current output voltage Eo is varied more in
dealing with the variation of alternating input voltage
in a wide range from the AC 100 V system to the AC 200 V
system than in dealing with the variation of the
alternating input voltage in a single range of only the
AC 100 V system or only the AC 200 V system, for example.
To perform constant-voltage control operation by dealing
with the thus extended variation in the level of the
secondary side direct-current output voltage Eo requires
a wider necessary control range obtained by extending the
above-mentioned range of 117.6 kHz to 208.3 kHz in a
direction of higher frequencies.
However, an upper limit of driving frequency
handled by an 1C (oscillation and drive circuit 2) for
driving the switching device in the present situation is
about 200 kHz. In addition, even if an 1C is developed
which can drive the switching device at high frequencies
as mentioned above, the high-frequency driving of the
switching device significantly decreases power conversion
efficiency, and thus makes it practically impossible to
put the power supply circuit to practical use.
It is thus understood that it is very difficult to
realize a wide range capability by the configuration of
the power supply circuit shown in FIG. 12, for example.
Because of such a situation, when a switching power
supply circuit including a resonant converter realizes a
wide range-capable operation, the switching power supply
circuit employs a configuration for switching the
configuration of the primary side switching converter to
half-bridge/full-bridge configuration according to the
commercial alternating-current power supply input of the
AC 100 V system/200 V system, for example. Alternatively,
the operation of a rectifier circuit rectifying the
alternating input voltage VAC is switched to full-wave
rectification/voltage doubler rectification according to
the commercial alternating-current power supply input of
the AC 100 V system/200 V system.
However, the following problems occur when
switching is performed between the circuit configurations
for the AC 100 V system and the AC 200 V system.
For example, for such switching according to the
level of commercial alternating-current power, a
threshold value (for example 150 V) is set for input
voltage. Circuit switching is performed to the AC 200 V
system when the input voltage exceeds the threshold value,
and to the AC 100 V system when the input voltage does
not exceed the threshold value. When only such simple
switching is performed, however, switching may be
performed to the AC 100 V system in response to a
temporary decrease in the alternating input voltage due
to an instantaneous power interruption or the like during
input of the AC 200 V system, for example. Specifically,
taking a configuration that switches rectifying operation
as an example, there is a possibility that the input of
the AC 200 V system is determined to be that of the AC
100 V system, and switching is thus performed to a
voltage doubler rectifier circuit, so that a switching
device or the like exceeds a withstand voltage thereof
and consequently breaks down.
Accordingly, in order to prevent the abovedescribed
erroneous operation, an actual circuit detects
not only the direct-current input voltage of the main
switching converter but also the direct-current input
voltage of a converter circuit on a standby power supply
side.
However, thus detecting the direct-current input
voltage of the converter circuit on the standby power
supply side means that for example a comparator 1C for
comparing the input voltage with a reference voltage is
incorporated. This increases the number of parts, and
thus contributes to increases in circuit manufacturing
cost and circuit board size.
In addition, thus detecting the direct-current
input voltage of the converter on the standby power
supply side for the purpose of preventing erroneous
operation means that the wide range-ready power supply
circuit can be actually used in only electronic devices
having a standby power supply in addition to a main power
supply. That is, electronic devices in which the power
supply circuit can be mounted are limited to types having
a standby power supply, and thus a range of applications
of the power supply circuit is correspondingly narrowed.
The configuration that switches between a half-
bridge configuration and a full-bridge configuration
requires at least four switching devices for the fullbridge
configuration. That is, while only two switching
devices are required for half-bridge configuration when
the switching is unnecessary, two other switching devices
need to be added in this case.
The configuration that switches rectifying
operation requires two smoothing capacitors Ci to obtain
voltage doubler rectifier operation. That is, one
smoothing capacitor Ci needs to be added as compared with
a configuration that performs only full-wave rectifier
operation.
In these respects, the wide range-ready
configurations involving the circuit switching as
described above increase circuit manufacturing cost and
the size of a power supply circuit board. Of parts
forming a power supply circuit, a smoothing capacitor Ci
or the like in the configuration that switches rectifying
operation, in particular, falls under the category of
large parts, thus contributing to further increase in the
size of the board.
Another problem of the wide control range of the
switching frequency as described above is a degradation
in quick response characteristics in stabilizing the
secondary side direct-current output voltage Eo.
Some recent electronic devices in particular
involve a load condition referred to as a so-called
switching load, in which load power is changed
instantaneously between a maximum load and no load
according to for example on/off operation of various
driving parts. The power supply circuit needs to perform
constant-voltage control on the secondary side directcurrent
output voltage in response to the load power thus
varied quickly over a wide range.
However, with a wide switching frequency control
range as described above, it takes a correspondingly long
time to change to a switching frequency necessary for the
constant-voltage control dealing with the load varied
between a maximum value and a minimum value. That is,
constant-voltage control response is slow.
The power supply circuit shown in FIG. 12 has the
primary side voltage resonant converter. The power supply
circuit having such a configuration is advantageous in
terms of power conversion efficiency, as described above.
In consideration of recent energy situations and
environment situations, for example, electronic devices
are desired to have higher power conversion efficiency.
Accordingly, the power supply circuit itself included in
an electronic device is desired to be further improved in
power conversion efficiency.
Accordingly, in view of the above problems, a
switching power supply circuit according to an embodiment
of the present invention is formed as follows.
The switching power supply circuit includes:
rectifying and smoothing means for obtaining a directcurrent
input voltage by rectifying and smoothing an
alternating input voltage; switching means formed with a
switching device supplied with the direct-current input
voltage and performing switching operation; and
switching-driving means for switching-driving the
switching device.
The switching power supply circuit further includes
an isolated converter transformer formed by at least
winding a primary winding supplied with a switching
output obtained by the switching operation of the
switching means and a secondary winding in which an
alternating voltage is induced by the switching output
obtained in the primary winding.
The switching power supply circuit further includes
a primary side parallel resonant circuit formed by at
least a leakage inductance component of the primary
winding of the isolated converter transformer and a
capacitance of a primary side parallel resonant capacitor,
the primary side parallel resonant circuit converting
operation of the switching means into a voltage resonant
type operation.
The switching power supply circuit further includes
a secondary side parallel resonant circuit formed by a
leakage inductance component of the secondary winding of
the isolated converter transformer and a capacitance of a
secondary side parallel resonant capacitor connected in
parallel with the secondary winding.
The switching power supply circuit further
includes: secondary side direct-current output voltage
generating means for generating a secondary side directcurrent
output voltage by performing a rectifying
operation on the alternating voltage induced in the
secondary winding of the isolated converter transformer
and input to the secondary side direct-current output
voltage generating means and smoothing a rectified output
resulting from the rectifying operation by a secondary
side smoothing capacitor; and constant-voltage control
means for performing constant-voltage control on the
secondary side direct-current output voltage by varying
switching frequency of the switching means by controlling
the switching-driving means according to level of the
secondary side direct-current output voltage.
A coupling coefficient for loose coupling between a
primary side and a secondary side of the isolated
converter transformer is set such that an electromagnetic
coupling type resonant circuit formed with the primary
side parallel resonant circuit and the secondary side
parallel resonant circuit has a unimodal characteristic
as characteristic of output in response to an input of a
frequency signal having the switching frequency. In
addition, a resonant frequency of the primary side
parallel resonant circuit and a resonant frequency of the
secondary side parallel resonant circuit are set so as to
obtain power conversion efficiency at a certain level and
higher under predetermined load conditions at least.
The thus formed power supply circuit employs the
fundamental configuration of a voltage resonant converter
provided with the secondary side parallel resonant
circuit on the secondary side. That is, the power supply
circuit has a secondary side parallel resonant circuit on
each of the primary side and the secondary side. Thus a
coupling type resonant circuit is formed by the
electromagnetic coupling of the isolated converter
transformer. Then, by setting the isolated converter
transformer in a state of loose coupling at a
predetermined coupling coefficient, a steep unimodal
characteristic can be obtained as characteristic of
output in response to the frequency signal (switching
output) having the switching frequency as an input to the
coupling type resonant circuit. As a result, a variable
control range (necessary control range) of the switching
frequency which range is necessary to stabilize the
secondary side direct-current output voltage is reduced.
In addition, the resonant frequency of the primary
side parallel resonant circuit and the resonant frequency
of the secondary side parallel resonant circuit are set
so as to obtain power conversion efficiency
characteristics at a certain level and higher under
predetermined load conditions.
Thus, the present invention reduces the variable
control range, that is, the necessary control range of
the switching frequency which range is necessary for
constant-voltage control for the voltage resonant
converter provided with the secondary side parallel
resonant circuit.
Thereby the wide range capability of the voltage
resonant switching converter can be easily realized by
only switching frequency control.
As for a fundamental configuration for realizing
such a wide range capability, it suffices to obtain a
required coupling coefficient in the isolated converter
transformer of the voltage resonant converter provided
with the secondary side parallel resonant circuit. It can
thus be said that the wide range capability is realized
without increases in cost, circuit size, weight and the
like due to increase in the number of parts, for example.
Further, according to the present invention, the
resonant frequency of the primary side parallel resonant
circuit and the resonant frequency of the secondary side
parallel resonant circuit are set so as to obtain power
conversion efficiency at a certain level and higher under
load conditions of predetermined load power. A voltage
resonant converter inherently has a characteristic of
high power conversion efficiency. The present invention
can provide a power supply circuit having a
characteristic of better power conversion efficiency as a
power supply circuit including a voltage resonant
converter.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a circuit diagram showing an example of
configuration of a power supply circuit according to a
first embodiment of the present invention;
FIG. 2 is a diagram showing an example of structure
of an isolated converter transformer provided in the
power supply circuit according to the embodiment;
FIGS. 3A and 3B are waveform charts of the
operations of principal parts in the power supply circuit
according to the first embodiment on the basis of a
switching cycle;
FIG. 4 is a diagram showing characteristics of
variation of AC—>DC power conversion efficiency, switching
frequency, and the on period of a switching device with
respect to load variation in the power supply circuit
according to the first embodiment;
FIG. 5 is a diagram showing characteristics of
variation of AC~^DC power conversion efficiency, switching
frequency, and the on period and the off period of the
switching device with respect to variation in alternating
input voltage in the power supply circuit according to
the first embodiment;
FIG. 6 is a diagram showing constant-voltage
control characteristics of the power supply circuit
according to the embodiment;
FIG. 7 is a diagram showing switching frequency
control ranges (necessary control ranges) according to
the condition of alternating input voltage and load
ariation, as constant-voltage control operation of the
power supply circuit according to the embodiment;
FIG. 8 is a circuit diagram showing an example of
configuration of a power supply circuit according to a
second embodiment of the present invention;
FIG. 9 is a circuit diagram showing an example of
configuration of a power supply circuit according to a
third embodiment of the present invention;
FIG. 10 is a circuit diagram showing an example of
configuration of a power supply circuit according to a
fourth embodiment of the present invention;
FIG. 11 is a circuit diagram showing an example of
configuration of a power supply circuit according to a
fifth embodiment of the present invention;
FIG. 12 is a circuit diagram showing an example of
configuration of a power supply circuit according to a
conventional example;
FIGS. 13A and 13B are waveform charts of the
operations of principal parts in the power supply circuit
shown in FIG. 12;
FIG. 14 is a diagram showing characteristics of
variation of AC—>DC power conversion efficiency, switching
frequency, and the on period and the off period of a
switching device with respect to load variation in the
power supply circuit shown in FIG. 12; and
FIG. 15 is a diagram conceptually showing the
constant-voltage control characteristics of the
conventional power supply circuit.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
A circuit diagram of FIG. 1 shows an example of
configuration of a power supply circuit according to a
first embodiment as the best mode (an embodiment) for
carrying out the invention. The power supply circuit
shown in this figure employs a fundamental configuration
of a voltage resonant switching converter using a singleended
system.
In the switching power supply circuit shown in this
figure, a set of common mode choke coils CMC and two
across capacitors CL are inserted in the line of a
commercial alternating-current power supply AC, as shown
in the figure. The common mode choke coils CMC and the
across capacitors CL and CL form a noise filter for
eliminating common mode noise superimposed on the line of
the commercial alternating-current power supply AC.
An alternating input voltage VAC is rectified by a
bridge rectifier circuit Di. A smoothing capacitor Ci is
charged with the rectified output of the bridge rectifier
circuit Di. Thereby a rectified and smoothed voltage Ei
is obtained as a voltage across the smoothing capacitor
Ci. The rectified and smoothed voltage Ei is a directcurrent
input voltage for the switching converter in a
subsequent stage.
In FIG. 1, the switching converter supplied with
the rectified and smoothed voltage Ei as the directcurrent
input voltage and performing switching operation
is formed as a voltage resonant converter of a singleended
system having one switching device Ql, for example.
A high withstand voltage MOS-FET is selected as the
switching device Ql in this case. A system for driving
the voltage resonant converter in this case is an
externally exciting system in which the switching device
is switching-driven by an oscillation and drive circuit 2
A switching driving signal (voltage) output from
the oscillation and drive circuit 2 is applied to the
gate of the switching device Ql.
The drain of the switching device Ql is connected
to the winding start end part of a primary winding Nl of
an isolated converter transformer PIT to be described
later. The winding termination end part of the primary
winding Nl is connected to the positive electrode
terminal of the smoothing capacitor Ci. Thus, in this
case, the direct-current input voltage (Ei) is supplied
to the switching device Ql via a series connection of the
primary winding Nl. The source of the switching device Ql
is connected to a primary side ground.
Since a MOS-FET is selected as the switching device
Ql in this case, a body diode DD is included in the
switching device Ql so as to be connected in parallel
with the drain and source of the switching device Ql, as
shown in FIG. 1. The body diode DD has an anode connected
to the source of the switching device Ql, and a cathode
connected to the drain of the switching device Ql. The
body diode DD forms a path for passing a switching
current in an opposite direction which current is
generated by on/off operation (switching operation) of
the switching device Ql.
A primary-side parallel resonant capacitor Cr is
connected in parallel with the drain and the source of
the switching device Ql.
The capacitance of the primary-side parallel
resonant capacitor Cr and a leakage inductance LI of the
primary winding Nl of the isolated converter transformer
PIT form a primary side parallel resonant circuit
(voltage resonant circuit) for a switching current
flowing through the switching device Ql. This primary
side parallel resonant circuit performs resonant
operation, whereby voltage resonant type operation is
obtained as switching operation of the switching device
Ql. Accordingly, a sinusoidal resonant pulse waveform is
obtained as a switching voltage (drain-to-source voltage)
VI across the switching device Ql during the off period
of the switching device Ql.
The oscillation and drive circuit 2 includes an
oscillating circuit to drive the switching device Ql by
an externally exciting system, for example. The
oscillation and drive circuit 2 generates a drive signal
as a gate voltage for switching-driving the MOS-FET on
the basis of an oscillating signal obtained from the
oscillating circuit, and applies the drive signal to the
gate of the switching device Ql. Thus the switching
device Ql performs continuous on/off operation according
to a switching frequency corresponding to the cycle of
the drive signal. That is, the switching device Ql
performs switching operation.
The isolated converter transformer PIT transmits
the switching output of the primary side switching
converter to a secondary side with the primary side and
the secondary side insulated from each other against
direct current.
FIG. 2 is a sectional view showing an example of
structure of the isolated converter transformer PIT
provided in the power supply circuit of FIG. 1.
As shown in FIG. 2, the isolated converter
transformer PIT has an EE type core (EE-shaped core)
formed by combining E-shaped cores CR1 and CR2 of ferrite
material with each other in such a manner that the
magnetic legs of the core CR1 are opposed to the magnetic
legs of the core CR2.
The isolated converter transformer PIT also has a
bobbin B formed by a resin, for example, and having a
primary side winding part and a secondary side winding
part divided from each other so as to be independent of
each other. The primary winding Nl is wound around one
winding part of the bobbin B. A secondary winding N2 is
wound around the other winding part of the bobbin E. The
bobbin B thus wound with the primary side winding and the
secondary side winding is attached to the above-described
EE type core (CRl and CR2) . Thereby the primary side
winding and the secondary side winding are wound around
the central magnetic leg of the EE type core in
respective different winding regions. The structure of
the isolated converter transformer PIT as a whole is thus
obtained.
In addition, a gap G having a gap length of about 2
mm or more, for example, is formed in the central
magnetic leg of the EE type core as shown in the figure.
Thereby a state of loose coupling with a coupling
coefficient k ^ about 0.7 or lower, for example, is
obtained. That is, a state of even looser coupling is
obtained than in the isolated converter transformer PIT
of the prior art power supply circuit shown in FIG. 12.
Incidentally, the gap G can be formed by making the
central magnetic leg of the E-type cores CR1 and CR2
shorter than the two outer magnetic legs of the E-type
cores CRl and CR2.
As described above, one end of the primary winding
Nl of the isolated converter transformer PIT is connected
to the drain of the switching device Ql. Thereby, the
switching output of the switching device Ql is
transmitted to the primary winding Nl, so that an
alternating voltage occurs in the primary winding Nl.
An alternating voltage induced by the primary
winding Nl occurs on the secondary side of the isolated
converter transformer PIT.
A secondary side parallel resonant capacitor C2 is
connected in series with the secondary winding N2. Thus,
the leakage inductance L2 of the secondary winding N2 and
the capacitance of the secondary side parallel resonant
capacitor C2 form a secondary side parallel resonant
circuit. This secondary side parallel resonant circuit
performs resonant operation according to rectifying
operation of a secondary side rectifier circuit to be
described later. That is, voltage resonant operation is
obtained on both the primary side and the secondary side.
The secondary side rectifier circuit in this case
is formed as a half-wave rectifier circuit by connecting
one rectifier diode Dol and one smoothing capacitor Co to
the secondary winding N2 connected in parallel with the
secondary side parallel resonant capacitor C2 as
described above. As for a mode of connection of this
half-wave rectifier circuit, the winding termination end
part side of the secondary winding N2 is connected with
the anode of the rectifier diode Dol. The cathode of the
rectifier diode Dol is connected to the positive
electrode terminal of the smoothing capacitor Co. The
winding start end part of the secondary winding N2 and
the negative electrode terminal of the smoothing
capacitor Co are connected to a secondary side ground.
Incidentally, since the rectifier diode Dol performs
on/off operation at a relatively high frequency
corresponding to the switching frequency, a fast type
(fast recovery type) diode is selected as the rectifier
diode Dol.
In the thus formed half-wave rectifier circuit, the
rectifier diode Dol conducts to pass a rectified current
and charge the smoothing capacitor Co with the rectified
current in a half-wave period in which a secondary
winding voltage V2 is of positive polarity on the winding
termination end part side of the secondary winding N2.
Thereby a secondary side direct-current output voltage Eo
having a level corresponding to once the level of the
alternating voltage induced in the secondary winding N2
is obtained as a voltage across the smoothing capacitor
Co.
The secondary side direct-current output voltage Eo
is supplied to a load. The secondary side direct-current
output voltage Eo also branches off to be output as a
detection voltage to a control circuit 1.
The control circuit 1 supplies the oscillation and
drive circuit 2 with a detection output according to
change in the level of the secondary side direct-current
output voltage Eo input to the control circuit 1. The
oscillation and drive circuit 2 drives the switching
device Ql while changing the switching frequency
according to the detection output of the control circuit
1 which output is input to the oscillation and drive
circuit 2. The changing of the switching frequency is an
operation of variably controlling the period TON of the
switching device Ql while fixing the period TOFF of the
switching device Ql. This operation is constant-voltage
control operation for the secondary side direct-current
output voltage.
By variably controlling the switching frequency of
the switching device Ql as described above, the resonant
impedances of the primary side and the secondary side in
the power supply circuit are changed, so that an amount
of power transmitted from the primary winding Nl to the
secondary winding N2 of the isolated converter
transformer PIT and an amount of power to be supplied
from the secondary side rectifier circuit to the load are
changed. The level of the secondary side direct-current
output voltage Eo is thereby controlled so as to cancel
variations in the level of the secondary side directcurrent
output voltage Eo. That is, the secondary side
direct-current output voltage Eo is stabilized.
Actual principal parts of the power supply circuit
having the circuit configuration shown in FIG. I are
formed by making settings as follows.
For the isolated converter transformer PIT, an EER-
35 core is selected, and the gap G is set to a gap length
of 2.2 mm. As for the respective numbers T of turns of
the primary winding Nl and the secondary winding N2, Nl =
63 T and N2 = 25 T are selected. A voltage induced per
turn of the secondary winding N2 is 5.4 V/T. Thereby k =
0.685 is set as the coupling coefficient k of the
isolated converter transformer PIT.
Cr = 4300 pF is selected as the capacitance of the
primary-side parallel resonant capacitor Cr. The resonant
frequency fol = 166.0 kHz of the primary side parallel
resonant circuit is set by this capacitance setting of
the primary-side parallel resonant capacitor Cr and the
leakage inductance LI of the primary winding Nl obtained
by the structure of the isolated converter transformer
PIT. C2 = 0.047 viF is selected as the capacitance of the
secondary side parallel resonant capacitor C2. The
resonant frequency fo2 = 109.0 kHz of the secondary side
parallel resonant circuit is set by this capacitance
setting of the secondary side parallel resonant capacitor
C2 and the leakage inductance L2 of the secondary winding
N2 obtained by the structure of the isolated converter
transformer PIT. It can be said that in relative terms,
there is a relation of fol ^ 1.5 X fo2.
As for load power handled by the power supply
circuit, maximum load power Pomax = 200 W, and minimum
load power Pomin = 0 W (no load) . The rated level of the
secondary side direct-current output voltage Eo is 135 V.
Waveform charts of FIGS. 3A and 3B show the
operations of principal parts in the thus formed power
supply circuit of FIG. 1 on the basis of the switching
cycle of the switching device Ql. FIG. 3A shows a
switching voltage VI, a switching current IQ1, a primary
winding current II, a secondary winding voltage V2, a
secondary winding current 12, and a secondary side
rectified current ID1 at the maximum load power Pomax =
200 W. FIG. 3B shows the respective waveforms of the
switching voltage VI, the switching current IQ1, the
primary winding current II, the secondary winding voltage
V2, the secondary winding current 12, and the secondary
side rectified current ID1 at the minimum load power
Pomin = 0 W.
The switching voltage VI is the drain-to-source
voltage of the switching device Ql. The switching current
IQ1 is a current flowing from a drain side to the
switching device Ql (and the body diode DD). The
switching voltage VI and the switching current IQ1
indicate on/off timing of the switching device Ql. One
switching cycle is divided into a period TON during which
the switching device Ql is on and a period TOFF during
which the switching device Ql is off. The switching
voltage VI is at a zero level in the period TON, and
forms a resonant pulse in the period TOFF. The resonant
pulse of the switching voltage VI is obtained as a
sinusoidal resonant waveform because the operation of the
primary side switching converter is voltage resonant type
operation.
The switching current IQ1 is at a zero level in the
period TOFF. On reaching turn-on timing when the period
TOFF is ended and the period TON is started, the
switching current IQ1 first forms a waveform of negative
polarity by flowing through the body diode DD. The
switching current IQ1 is then inverted to form a waveform
of positive polarity by flowing from the drain to the
source. Such waveforms of the switching current IQ1
indicate that ZVS is performed properly.
The primary winding current II is a current flowing
through the primary winding Nl, and is a combination of a
current component flowing through the switching device Ql
and a current flowing through the primary-side parallel
resonant capacitor Cr. The waveform of the primary
winding current II in the period TOFF corresponds to the
waveform of the current flowing through the primary-side
parallel resonant capacitor Cr.
As the alternating voltage is induced in the
secondary winding N2, the rectifier diode Dol conducts to
pass the rectified current IDl within a period of this
conduction in each half-cycle period in which the
secondary winding voltage V2 is of positive polarity and
at a certain level or higher. Correspondingly, the
secondary winding voltage V2 is clamped at the secondary
side direct-current output voltage Eo during the period
during which the voltage induced in the secondary winding
N2 makes the rectifier diode Dol conduct as a voltage
having a level equal to or higher than the secondary side
direct-current output voltage Eo. In a period when the
rectifier diode Dol does not conduct, the secondary
winding voltage V2 forms a sinusoidal envelope having a
level not higher than the secondary side direct-current
output voltage Eo. The secondary winding current 12 is
obtained by combining the rectified current IDl with a
current flowing through the secondary side parallel
resonant capacitor C2. Incidentally, the rectified
current IDl is at a zero level even within the conduction
period of the rectifier diode Dol at the minimum load
power Pomin = 0 W.
A comparison of the waveforms at the maximum load
power Pomax = 200 W as shown in FIG. 3A with waveforms at
the minimum load power Pomin = 0 W as shown in FIG. 3B
with the above-described operations of the respective
parts in mind shows that the operation of the primary
side switching converter shortens the period length of
one switching cycle (TOFF + TON), that is, increases the
switching frequency fs as a transition is made to no load.
This indicates that switching frequency control operation
that changes the switching frequency fs according to load
variation as described above is obtained as stabilizing
operation. It is also shown that in changing the
switching frequency, the period TON is changed while the
period TOFF is fixed.
FIG. 4 shows characteristics of variation of AC—>DC
power conversion efficiency (TjAC—>DC), the switching
frequency fs, and the time length of the period TON with
respect to load variation at the alternating input
voltage VAC = 100 V as results of an experiment on the
power supply circuit shown in FIG. 1. FIG. 5 shows
characteristics of variation of the AC—>DC power
conversion efficiency (7?AC—>DC), the switching frequency
fs, and the time length of the period TON/TOFF with
respect to variation in the alternating input voltage at
the maximum load power Pomax = 200 W.
According to these figures, the switching frequency
fs is changed so as to be increased as the load becomes
lighter, as described above. Also, the switching
frequency fs is changed so as to be increased as the
alternating input voltage VAC is raised. This indicates
that the constant-voltage control operation controls the
switching frequency fs so as to increase the switching
frequency fs as the secondary side direct-current output
voltage Eo is raised as the load becomes lighter and as
the alternating input voltage is raised. The time length
of the period TON is shortened as the load becomes
lighter, and is shortened as the alternating input
voltage is raised. On the other hand, the period TOFF is
fixed irrespective of variation in the alternating input
voltage. Though not shown, the period TOFF is also fixed
irrespective of load variation. This indicates the
operation of changing the switching frequency by changing
the period TON according to load variation while the
period TOFF is fixed.
As concrete values of the switching frequency fs,
fs = 112.4 kHz to 149.3 kHz, and Afs = 36.9 kHz in a
range of the maximum load power Pomax = 200 W to the
minimum load power Pomin = 0 W at the alternating input
voltage VAC = 100 V. The periods TON and TOFF
corresponding to this variation of the switching
frequency fs are TON = 6.2 us to 4.1 us, and TOFF = 2.7
us (fixed).
At the alternating input voltage VAC = 230 V, fs =
155 kHz to 190.3 kHz, and Afs =35.3 kHz in the range of
the maximum load power Pomax = 200 W to the minimum load
power Pomin = 0 W. The periods TON and TOFF corresponding
to this variation of the switching frequency fs are TON =
3.75 us to 2.55 us, and TOFF = 2.7 us (fixed).
The AC~>DC power conversion efficiency (rjAC~>DC) is
increased as the load becomes heavier, and is increased
as the alternating input voltage VAC is lowered.
Experimental results of rj AC-^DC = 93.5% at the
alternating input voltage VAC = 100 V and 77 AC~>DC = 90.5%
at the alternating input voltage VAC = 230 V are obtained
as the AC~>DC power conversion efficiency (TjAC—>DC) at
the maximum load power Pomax = 200 W.
The characteristics of the switching frequency fs
of the above-described power supply circuit of FIG. I
will first be compared with the characteristics of the
switching frequency fs of the power supply circuit of FIG.
12.
In the power supply circuit of FIG. 12, fs = 117.6
kHz to 208.3 kHz, and Afs =96.7 kHz in the variation
range of the maximum load power Pomax = 200 W to the
minimum load power Pomin = 0 W at the alternating input
voltage VAC = 100 V.
On the other hand, in the power supply circuit of
FIG. 1, fs = 112.4 kHz to 149.3 kHz, and Afs = 36.9 kHz
in the variation range of the maximum load power Pomax =
200 W to the minimum load power Pomin = 0 W at the
alternating input voltage VAC = 100 V. It is understood
that the necessary control range is greatly reduced as
compared with the characteristics of the power supply
circuit of FIG. 12. Further, in the power supply circuit
of FIG. 1, fs = 155 kHz to 190.3 kHz, and Afs = 35.3 kHz
in the variation range of the maximum load power Pomax =
200 W to the minimum load power Pomin = 0 W at the
alternating input voltage VAC = 230 V. Also in this
condition, the necessary control range is greatly reduced
as compared with the characteristics of the power supply
circuit of FIG. 12.
Such characteristics of the switching frequency fs
of the power supply circuit in FIG. 1 indicate that socalled
wide range capability is realized which enables
stabilization while dealing with the input of the
commercial alternating-current power supply in a range of
an AC 100 V system to an AC 200 V system (for example VAC
= 85 V to 264 V) under a condition of handled load power
of the maximum load power Pomax = 200 W to the minimum
load power Pomin = 0 W. This will be described in the
following.
The power supply circuit shown in FIG. 1 employs
the fundamental configuration of the voltage resonant
converter provided with the secondary side parallel
resonant circuit. That is, it can be said that the power
supply circuit shown in FIG. 1 has respective parallel
resonant circuits on the primary side and the secondary
side via electromagnetic induction of the isolated
converter transformer PIT. As viewed from a relation
between the primary side parallel resonant circuit and
the secondary side parallel resonant circuit, such a
configuration can be considered equivalent to a coupling
type resonant circuit formed by electromagnetic coupling
which circuit is supplied with a frequency signal
corresponding to the switching frequency fs.
The characteristic of constant-voltage control for
the secondary side direct-current output voltage Eo in
the power supply circuit of FIG. 1 including such an
electromagnetic coupling type resonant circuit differs
according to the degree of coupling (coupling coefficient
k) of the isolated converter transformer PIT. This will
be described with reference to FIG. 6.
FIG. 6 shows characteristics of output of the
above-described electromagnetic coupling type resonant
circuit in response to input (switching frequency signal)
That is, FIG. 6 represents characteristics of control of
the secondary side direct-current output voltage Eo by
relation of the secondary side direct-current output
voltage Eo to the switching frequency fs. In this figure,
an axis of abscissas indicates the switching frequency,
and an axis of ordinates indicates the level of the
secondary side direct-current output voltage Eo.
As described with reference to FIG. 1, in the first
embodiment, the resonant frequency fol of the primary
side parallel resonant circuit is set to about 1.5 times
the resonant frequency fo2 of the secondary side parallel
resonant circuit. Therefore the resonant frequency fol is
higher than the resonant frequency fo2. The resonant
frequencies fol and fo2 are shown in correspondence with
each other on the axis of abscissas indicating the
switching frequency in FIG. 6. Also in FIG. 6, the
resonant frequency fol is shown to be higher than the
resonant frequency fo2 in correspondence with the
relation between the resonant frequencies fol and fo2.
Suppose that a state of close coupling at a
coupling coefficient k = 1 as the degree of coupling of
the isolated converter transformer PIT is set. Then, the
leakage inductance LI of the primary winding Nl and the
leakage inductance L2 of the secondary winding N2 in this
case are each zero.
Thus, a constant-voltage control characteristic in
the state of close coupling between the primary side and
the secondary side of the isolated converter transformer
PIT is a so-called bimodal characteristic in which the
secondary side direct-current output voltage Eo peaks at
frequencies fl and f2 different from the resonant
frequency fol of the primary side parallel resonant
circuit and the resonant frequency fo2 of the secondary
side parallel resonant circuit, as represented by a
characteristic curve 1 in FIG. 6.
The frequency fl is expressed by
[Equation 1]
fl = fo/Vl + k
The frequency f2 is expressed by
[Equation 2]
f2 = fo/Vl-k
A term fo in the above (Equation 1) and (Equation
2) is an intermediate resonant frequency intermediate
between the resonant frequency fol of the primary side
parallel resonant circuit and the resonant frequency fo2
of the secondary side parallel resonant circuit. The
frequency fo is determined by a primary side impedance
and a secondary side impedance and an impedance (mutual
coupling inductance M) common to the primary side and the
secondary side. The mutual coupling inductance M is
expressed by
[Equation 3]
M = WLlxL2
When the above-described coupling coefficient k is
gradually decreased from k = 1, that is, when a degree of
loose coupling is gradually increased from the state of
close coupling, the characteristic curve 1 shown in FIG.
6 is changed such that a bimodal tendency is gradually
weakened and the characteristic curve is flattened around
the intermediate resonant frequency fo. Then, a state of
so-called critical coupling occurs when the coupling
coefficient k is decreased to a certain coupling
coefficient k. In this state of critical coupling, as
represented by a characteristic curve 2, the bimodal
characteristic tendency disappears, and the shape of the
curve is flattened around the intermediate resonant
frequency fo.
When the coupling coefficient k is further
decreased from the state of critical coupling to a state
of looser coupling, a unimodal characteristic in which
the secondary side direct-current output voltage Eo peaks
only at the intermediate frequency fo is obtained, as
represented by a characteristic curve 3 in FIG. 6. A
comparison of the characteristic curve 3 with the
characteristic curves 1 and 2 indicates that while a peak
level itself of the characteristic curve 3 is lower than
those of the characteristic curves 1 and 2, the
characteristic curve 3 has a steeper slope as a quadratic
function curve shape than the characteristic curves 1 and
2.
A state of loose coupling at the coupling
coefficient k ^ 0.7 or lower is set in the isolated
converter transformer PIT according to the first
embodiment. When the coupling coefficient k is thus set,
operation based on the unimodal characteristic
represented as the characteristic curve 3 is performed.
An actual comparison between the unimodal
characteristic shown in FIG. 6 and the constant-voltage
control characteristic shown in FIG. 15 of the
conventional power supply circuit shown in FIG. 12
indicates that the characteristic shown in FIG. 15 has a
considerably gentler slope as a quadratic function curve
as compared with the unimodal characteristic shown in FIG,
6.
Because the characteristic shown in FIG. 15 has a
gentle curve as described above, the necessary control
range of the switching frequency fs for performing
constant-voltage control on the secondary side directcurrent
output voltage Eo is for example Afs = 96.7 kHz
with fs = 117.6 kHz to 208.3 kHz even under conditions
for dealing with a single range at the alternating input
voltage VAC = 100 V. It is therefore very difficult to
realize a wide range capability by only constant-voltage
control by switching frequency control, as described
above.
On the other hand, the constant-voltage control
characteristic of the first embodiment is the unimodal
characteristic represented by the characteristic curve 3
in FIG. 6, and thus constant-voltage control operations
of the first embodiment are as shown in FIG. 7.
FIG. 7 shows four characteristic curves of the
power supply circuit according to the first embodiment
shown in FIG. 1, that is, characteristic curves A and B
at the time of maximum load power Pomax and at the time
of minimum load power Pomin, respectively, when the
alternating input voltage VAC = 100 V (AC 100 V system)
and characteristic curves C and D at the time of maximum
load power Pomax and at the time of minimum load power
Pomin, respectively, when the alternating input voltage
VAC = 230 V (AC 200 V system).
As is clear from FIG. 7, Afsl denotes the variable
control range (necessary control range) of the switching
frequency which range is necessary to hold the secondary
side direct-current output voltage Eo constant at a
required rated level tg when the alternating input
voltage VAC = 100 V corresponding to the input of the AC
100 V system. That is, the variable control range
(necessary control range) of the switching frequency is a
frequency range from the switching frequency fs at the
level tg in the characteristic curve A to the switching
frequency fs at the level tg in the characteristic curve
B.
In addition, Afs2 denotes the variable control
range (necessary control range) of the switching
frequency which range is necessary to hold the secondary
side direct-current output voltage Eo constant at the
required rated level tg when the alternating input
voltage VAC = 230 V corresponding to the input of the AC
200 V system. That is, the variable control range
(necessary control range) of the switching frequency is a
frequency range from the switching frequency fs at the
level tg in the characteristic curve C to the switching
frequency fs at the level tg in the characteristic curve
D.
As described above, the unimodal characteristic as
the characteristic of control of the secondary side
direct-current output voltage Eo in the first embodiment
has a considerably steeper slope as a quadratic function
curve than the control characteristic shown in FIG. 15.
Therefore the above-described necessary control
ranges Afsl and Afs2 when the alternating input voltage
VAC = 100 V and 230 V are considerably reduced as
compared with Afs shown in FIG. 15.
Thus, a frequency variable range (AfsA) from the
lowest switching frequency in Afsl (the switching
frequency fs at the level tg in the characteristic curve
A) to the highest switching frequency in Afs2 (the
switching frequency fs at the level tg in the
characteristic curve D) is correspondingly narrowed.
For confirmation, Afsl, Afs2, and AfsA actually
measured in the power supply circuit of FIG. 1 are
Afsl = 36.9 kHz (= 149.3 kHz - 112.4 kHz)
Afs2 = 35.3 kHz (= 190.3 kHz - 155 kHz)
AfsA =77.9 kHz (=190.3 kHz - 112.4 kHz)
The frequency variable range AfsA easily falls
within the variable range of the switching frequency
dealt with by a switching-driving 1C (oscillation and
drive circuit 2) in the present situation. That is, the
power supply circuit shown in FIG. 1 can actually control
the switching frequency variably in the frequency
variable range AfsA.
Thus, the power supply circuit shown in FIG. 1 can
stabilize the secondary side direct-current output
voltage Eo as main direct-current power properly while
dealing with the commercial alternating-current power
supply input of either of the AC 100 V system and the AC
200 V system. That is, the power supply circuit shown in
FIG. 1 achieves a wide range capability by only switching
frequency control.
Incidentally, an electromagnetic coupling type
resonant circuit is already known as a technique for
expanding an amplification bandwidth of an amplifier
circuit formed by a transistor in communication
technology as in an intermediate-frequency transformer
amplifier, for example. In such a field, however, a
bimodal characteristic in close coupling or a flat
characteristic in critical coupling is used, but a
unimodal characteristic in loose coupling is not used. It
can be said that in the techniques of such an
electromagnetic coupling type resonant circuit, the
unimodal characteristic in loose coupling that has not
been used in the field of communication technology is
aggressively used in the first embodiment in a field of
resonant switching converters. Thereby, as described
above, the variable range (necessary control range) of
the switching frequency which range is necessary to
stabilize the secondary side direct-current output
voltage Eo is reduced, and a wide range capability can be
achieved by only constant-voltage control by switching
frequency control.
Incidentally, in general, as the degree of loose
coupling between the primary side and the secondary side
of the isolated converter transformer PIT is increased, a
power loss (eddy-current loss) in the isolated converter
transformer PIT tends to increase, and power conversion
efficiency is correspondingly decreased. However, the
first embodiment provides a practically sufficient power
conversion efficiency characteristic, as will be
described later. This is because a resonant circuit (the
secondary side parallel resonant circuit) is formed also
on the secondary side.
That-is, the secondary side parallel resonant
circuit makes it possible to supply power as the
secondary side direct-current output voltage Eo including
an energy increase obtained by resonant operation of the
secondary side parallel resonant circuit, thus
compensating for the decrease in the power conversion
efficiency due to loose coupling.
In addition, the substantial reduction of the
necessary control range (Afs) of the switching frequency
fs for constant-voltage control under the condition of
the commercial alternating-current power supply input of
each of the AC 100 V system and the AC 200 V system as
described above greatly improves constant-voltage control
response and control sensitivity.
Some electronic devices perform operation that
varies the load power Po in such a manner as to change
(switch) relatively rapidly between a maximum load and no
load, such a load variation being referred to as a socalled
switching load. Devices that perform such a
switching load operation include for example a printer as
a peripheral device for a personal computer.
When a power supply circuit having a relatively
wide necessary control range Afs as shown in FIG. 15, for
example, is included in a device performing such a
switching load operation, the switching frequency fs is
controlled to be changed by a correspondingly large
amount of change following an abrupt change in load power,
as described above. It is therefore difficult to obtain
quick constant-voltage control response.
On the other hand, since the first embodiment
greatly reduces the necessary control range Afs in a
region of each single range, in particular, the first
embodiment can stabilize the secondary side directcurrent
output voltage Eo by quickly responding to an
abrupt variation in the load power Po between a maximum
load and no load. That is, the performance of constantvoltage
control response to the switching load is greatly
improved.
As for the power conversion efficiency
characteristics described above with reference to FIG. 4
and FIG. 5, the power supply circuit of FIG. 1 has
T] AC-DC = 93.5% at the alternating input voltage VAC =
100 V and the maximum load power Pomax = 200 W. This
characteristic represents an improvement of 1.2
percentage points over r] AC—>DC = 92.3% at the alternating
input voltage VAC = 100 V and the maximum load power
Pomax = 200 W as a characteristic of the conventional
power supply circuit shown in FIG. 12, for example.
Correspondingly, alternating-current input power is
reduced by 2.8 W.
While compensating for decrease in the power
conversion efficiency by providing the secondary side
resonant circuit as described above is one factor in
obtaining such favorable power conversion efficiency, the
setting of the respective resonant frequencies fol and
fo2 of the primary side parallel resonant circuit and the
secondary side parallel resonant circuit is a major
factor in obtaining such favorable power conversion
efficiency in the first embodiment. The power conversion
efficiency characteristic under the load conditions of
the first embodiment as described above is ultimately
obtained by adjusting the resonant frequencies fol and
fo2. That is, the power conversion efficiency
characteristic was obtained finally after experiments
were performed with various settings made for the
resonant frequencies fol and fo2, and fol = 166.0 kHz and
fo2 = 109.0 kHz were set as described above. A comparison
of the resonant frequencies fol and fo2 of the first
embodiment with the conventional resonant frequencies fol
and fo2 shows that while the resonant frequencies fol and
fo2 of the power supply circuit shown in FIG. 12 are fol
= 175.0 kHz and fo2 = 164.0 kHz and thus have a similar
relation of fol fo2, the values of the respective
frequencies and a frequency difference between the
frequencies are different from those of the first
embodiment. The respective frequency values of the
resonant frequencies fol and fo2 of the first embodiment
are reduced as compared with the conventional resonant
frequencies fol and fo2, and a frequency difference
between the resonant frequencies fol and fo2 of the first
embodiment is greatly increased.
One reason for the improvement of the power
conversion efficiency by setting the resonant frequencies
fol and fo2 as described above is as follows. As is
understood from a comparison of the switching current IQ1
in FIG. 3A with the switching current IQ1 in FIG. 13A,
the waveform of the switching current IQ1 in FIG. 3A
corresponding to the first embodiment has a peak level of
4 Ap before turn-off timing in which timing the period
TON of the switching device Ql is ended and a transition
is made to the period TOFF. The level is decreased to 3
Ap when the turn-off timing is reached.
The waveform of such a switching current IQ1 is
affected by the waveform of the secondary winding current
12. That is, the waveform of the switching current IQ1
has a waveform component corresponding to the current
flowing through the secondary side parallel resonant
circuit. The waveform of the secondary winding current 12
is determined by the setting of the resonant frequency
fo2 in relation to the resonant frequency fol.
Thus, the waveform of the switching current IQ1 in
the power supply circuit shown in FIG. I is obtained by
the appropriate setting of the respective resonant
frequencies fol and fo2 of the primary side parallel
resonant circuit and the secondary side parallel resonant
circuit.
The waveform of the switching current IQ1 shown in
FIGS. 3A indicates that the level of the switching
current IQ1 at the turn-off time is suppressed. When the
level of the switching current IQ1 at the turn-off time
is suppressed, a switching loss and a conduction loss at
the turn-off time are correspondingly reduced.
In the power supply circuit according to the first
embodiment, the secondary side rectified current ID1 also
has a level suppressed at the turn-off time as shown in
FIG. 3A in correspondence with the decrease from the peak
level in the waveform of the switching current IQ1 at the
turn-off time, as described above. Incidentally, the
switching current IQ1 in the conventional power supply
circuit has a peak level at the turn-off time, as shown
in FIG. 13A. Thus, a switching loss and a conduction loss
in the rectifier diode Dol are also reduced.
Such reductions of the switching loss and the
conduction loss in the switching device and the
rectifying element are a main factor in obtaining a high
power conversion efficiency characteristic in the power
supply circuit according to the first embodiment.
Variations of the secondary side rectifier circuit
are shown in FIGS. 8 to 11 as other embodiments of the
power supply circuit according to the present invention.
FIG. 8 shows a configuration of a power supply
circuit according to a second embodiment of the present
invention.
Incidentally, only an isolated converter
transformer PIT and a secondary side rectifier circuit
are shown in this figure. Parts other than the isolated
converter transformer PIT and the secondary side
rectifier circuit are the same as in FIG. 1, and are not
shown in FIG. 8. In FIG. 8, the same parts as in FIG. 1
are identified by the same reference numerals, and
description thereof will be omitted. This is true for
FIGS. 9 to 11 to be described later.
Also in the power supply circuit shown in FIG. 8, a
secondary side parallel resonant circuit is formed by
connecting a secondary side parallel resonant capacitor
C2 in parallel with a secondary winding N2. In addition,
a bridge full-wave rectifier circuit is provided as a
secondary side rectifier circuit. This bridge full-wave
rectifier circuit is formed by connecting a bridge
rectifier circuit formed by four rectifier diodes Dol,
Do2, Do3, and Do4 to a smoothing capacitor Co as shown in
the figure.
The thus formed bridge full-wave rectifier circuit
alternately performs an operation in which the rectifier
diodes Dol and Do4 conduct to charge the smoothing
capacitor Co and an operation in which the rectifier
diodes Do2 and Do3 conduct to charge the smoothing
capacitor Co in respective half cycles of a voltage
induced in the secondary winding N2. Thereby a secondary
side direct-current output voltage Eo having a level
corresponding to once the level of the voltage induced in
the secondary winding N2 is obtained as a voltage across
the smoothing capacitor Co.
Principal parts of the power supply circuit
according to the second embodiment are selected as
follows. Incidentally, as for load power handled by the
power supply circuit, maximum load power Pomax = 200 W
and minimum load power Pomin = 0 W (no load), and the
rated level of the secondary side direct-current output
voltage Eo is 135 V, as in the power supply circuit of
FIG. 1.
For the isolated converter transformer PIT, an EER-
35 core is selected, and a gap G is set to a gap length
of 2 mm. As for the number T of turns of a primary
winding Nl, Nl = 37 T. As for the number T of turns of
the secondary winding N2, N2 = 25 T. A voltage induced
per turn of the secondary winding N2 is 5.4 V/T . Thereby
k = 0.685 is set as the coupling coefficient k of the
isolated converter transformer PIT.
Cr = 3900 pF is selected as the capacitance of a
primary-side parallel resonant capacitor Cr. The resonant
frequency fol of a primary side parallel resonant circuit
is set to fol = 174.3 kHz by this capacitance setting of
the primary-side parallel resonant capacitor Cr and the
leakage inductance LI of the primary winding Nl obtained
by the structure of the isolated converter transformer
PIT. C2 = 0.047 iaF is selected as the capacitance of the
secondary side parallel resonant capacitor C2. The
resonant frequency fo2 of the secondary side parallel
resonant circuit is set to fo2 = 109.0 kHz by this
capacitance setting of the secondary side parallel
resonant capacitor C2 and the leakage inductance L2 of
the secondary winding N2 obtained by the structure of the
isolated converter transformer PIT.
Experiments conducted on the thus formed power
supply circuit according to the second embodiment show
that as operation waveforms of the principal parts,
substantially the same results as in the waveform charts
of FIGS. 3A to 3C are obtained, and that a switching
current IQ1 at heavy load has a peak level suppressed at
a turn-off time. However, the secondary side rectifier
circuit in this case performs full-wave rectifier
operation. Thus, in a half cycle of one polarity of a
secondary winding voltage V2, a rectified current ID1
flows to the smoothing capacitor Co according to
rectifying operation of the rectifier diodes Dol and Do4,
and in a half cycle of another polarity of the secondary
winding voltage V2, a rectified current ID2 flows to the
smoothing capacitor Co according to rectifying operation
of the rectifier diodes Do2 and Do3. In correspondence
with the rectified currents thus flowing in the doublewave
period, the positive and negative peak levels of the
secondary winding voltage V2 are both clamped at the
level of the secondary side direct-current output voltage
Eo.
As for characteristics of switching frequency fs
and periods TON and TOFF representing constant-voltage
control characteristics, results obtained show that fs =
100.0 kHz to 135.1 kHz, Afs = 35.1 kHz, TON = 6.6 us to
4.5 us, and TOFF = 3.4 us to 2.9 us with the commercial
alternating-current power supply input of an alternating
input voltage VAC = 100 V corresponding to an AC 100 V
system with respect to load variation from maximum load
power Pomax = 200 W to minimum load power Pomin = 0 W.
Also, results obtained show that fs = 145 kHz to 179.1
kHz, Afs = 34.1 kHz, TON = 3.6 \is to 2.1 'vis, and TOFF =
3.3 us to 2.5 us with the commercial alternating-current
power supply input of an alternating input voltage VAC =
230 V corresponding to an AC 200 V system with respect to
the load variation from the maximum load power Pomax =
200 W to the minimum load power Pomin = 0 W. AfsA is 79.1
kHz (= 179.1 kHz - 100.0 kHz).
Such characteristics indicate that the second
embodiment also achieves a wide range capability by only
switching frequency control.
As for AC—>DC power conversion efficiency ( rj AC~»DC) ,
rjAC->DC = 93.9% at the alternating input voltage VAC =
100 V and the maximum load power Pomax = 200 W. This
represents an improvement of 1.6 percentage points over
the power conversion efficiency of the conventional power
supply circuit. Alternating-current input power is
reduced by 3.7 W. At the alternating input voltage VAC =
230 V and the maximum load power Pomax = 200 W, r\ AC~»DC =
90.9%.
FIG. 9 shows a configuration of a third embodiment.
The third embodiment has a double-wave rectifier
circuit as a secondary side rectifier circuit.
In order to form the double-wave rectifier circuit,
a secondary winding N2 is provided with a center tap to
be divided into secondary winding parts N2A and N2B with
the center tap as a boundary. The center tap is connected
to a secondary side ground.
A secondary side parallel resonant capacitor C2 for
forming a secondary side parallel resonant circuit in
this case is connected in parallel with the whole of the
secondary winding N2.
In addition, two rectifier diodes Dol and Do2 and
one smoothing capacitor Co are provided as parts or
elements forming the secondary side rectifier circuit in
this case. The anode of the rectifier diode Dol is
connected to an end part on the secondary winding part
N2A side of the secondary winding N2. The anode of the
rectifier diode Do2 is connected to an end part on the
secondary winding part N2B side of the secondary winding
N2. The cathodes of the rectifier diodes Dol and Do2 are
both connected to the positive electrode terminal of the
smoothing capacitor Co. The negative electrode terminal
of the smoothing capacitor Co is connected to the
secondary side ground.
In the thus formed secondary side double-wave
rectifier circuit, in a half cycle of one polarity of a
secondary winding voltage V2 induced in the secondary
winding N2, a rectified current ID1 flows through a path
from the secondary winding part N2A to the rectifier diode
Dol to the smoothing capacitor Co to charge the smoothing
capacitor Co. In a half cycle of another polarity of the
secondary winding voltage V2 induced in the secondary
winding N2, a rectified current ID2 flows through a path
from the secondary winding part N2B to the rectifier
diode Do2 to the smoothing capacitor Co to charge the
smoothing capacitor Co. Thus, full-wave rectifier
operation is performed in which the smoothing capacitor
Co is charged with the rectified current in the period of
each of the positive and negative half cycles of the
secondary winding voltage V2. Thereby a secondary side
direct-current output voltage Eo having a level
corresponding to once the level of the voltage induced in
the secondary winding N2 is obtained as a voltage across
the smoothing capacitor Co.
The thus formed power supply circuit according to
the third embodiment is formed by selecting substantially
the same principal parts as in the second embodiment, for
example, so that the coupling coefficient k of an
isolated converter transformer PIT is k ^ 0.7 or lower.
Thereby a wide range capability is realized by only
switching frequency control. The resonant frequency fol
of a primary side parallel resonant circuit and the
resonant frequency fo2 of the secondary side parallel
resonant circuit are set so as to obtain a certain value
or higher as AC—>DC power conversion efficiency. As a
result, as in the foregoing embodiments, the waveform of
a switching current IQ1 at heavy load has a peak level
suppressed at a time of turning off a switching device Ql,
FIG. 10 shows an example of configuration of a
power supply circuit according to a fourth embodiment.
The power supply circuit shown in FIG. 10 has a
voltage doubler rectifier circuit as a secondary side
rectifier circuit. The voltage doubler rectifier circuit
is formed by connecting two rectifier diodes Dol and Do2
and two smoothing capacitors Col and Co2 to a parallel
connection circuit of a secondary winding N2 and a
secondary side parallel resonant capacitor C2 as shown in
FIG. 10.
In this case, the anode of the rectifier diode Dol
and the cathode of the rectifier diode Do2 are connected
to the winding termination end part of the secondary
winding N2. The cathode of the rectifier diode Dol is
connected to the positive electrode terminal of the
smoothing capacitor Col.
The smoothing capacitors Col and Co2 are connected
in series with each other such that the negative
electrode terminal of the smoothing capacitor Col is
connected to the positive electrode terminal of the
smoothing capacitor Co2. The winding start end part of
the secondary winding N2 is connected to a point of
connection between the smoothing capacitors Col and Co2.
The negative electrode terminal of the smoothing
capacitor Co2 and the anode of the rectifier diode Do2
are connected to a secondary side ground.
In the thus formed secondary side voltage doubler
rectifier circuit, in a half cycle of one polarity of a
secondary winding voltage V2, a rectified current flows
through a path from the secondary winding N2 to the
rectifier diode Dol to the smoothing capacitor Col to
charge the smoothing capacitor Col. In a half cycle of
another polarity of the secondary winding voltage V2, a
rectified current flows through a path from the secondary
winding N2 to the rectifier diode Do2 to the smoothing
capacitor Co2 to charge the smoothing capacitor Co2. Thus,
the charging of the smoothing capacitor Col and the
charging of the smoothing capacitor Co2 are alternately
performed in the positive and negative half cycles of the
secondary winding voltage V2. A potential corresponding
to once the level of the voltage induced in the secondary
winding N2 is obtained at each of the smoothing
capacitors Col and Co2. Thereby a secondary side directcurrent
output voltage Eo having a level corresponding to
once the level of the voltage induced in the secondary
winding N2 is obtained as a voltage across the series
connection circuit of the smoothing capacitors Col and
Co2.
Principal parts of the power supply circuit
according to the fourth embodiment are selected as
follows.
Also in this power supply circuit, as for load
power handled by the power supply circuit, maximum load
power Pomax = 200 W and minimum load power Pomin = 0 W
(no load), and the rated level of the secondary side
direct-current output voltage Eo is 135 V, as in the
power supply circuit of FIG. 1.
For an isolated converter transformer PIT, an EER-
35 core is selected, and a gap G is set to a gap length
of 2 mm. As for the number T of turns of a primary
winding Nl, Nl = 63 T. As for the number T of turns of
the secondary winding N2, N2 = 13 T. A voltage induced
per turn of the secondary winding N2 is 5.4 V/T. Thereby
k = 0.675 is set as the coupling coefficient k of the
isolated converter transformer PIT.
Cr = 3900 pF is selected as the capacitance of a
primary-side parallel resonant capacitor Cr. The resonant
frequency fol of a primary side parallel resonant circuit
is set to fol = 174.3 kHz by this capacitance setting of
the primary-side parallel resonant capacitor Cr and the
leakage inductance LI of the primary winding Nl obtained
by the structure of the isolated converter transformer
PIT. C2 = 0.18 uF is selected as the capacitance of the
secondary side parallel resonant capacitor C2. The
resonant frequency fo2 of the secondary side parallel
resonant circuit is set to fo2 = 110.0 kHz by this
capacitance setting of the secondary side parallel
resonant capacitor C2 and the leakage inductance L2 of
the secondary winding N2 obtained by the structure of the
isolated converter transformer PIT.
Results of experiments conducted on the thus formed
power supply circuit according to the fourth embodiment
show that the operation waveforms of the principal parts
of the power supply circuit according to the fourth
embodiment are substantially the same as in the waveform
charts of FIGS. 3A to 3C, and that a switching current
IQ1 at heavy load has a peak level suppressed at a turnoff
time.
As for characteristics of switching frequency fs
and periods TON and TOFF representing constant-voltage
control characteristics, results obtained show that fs =
98.5 kHz to 132.1 kHz, Afs =35.6 kHz, TON = 6.9 us to
4.8 las, and TOFF = 3.3 us to 2.8 us with the commercial
alternating-current power supply input of an alternating
input voltage VAC = 100 V corresponding to an AC 100 V
system with respect to load variation from maximum load
power Pomax = 200 W to minimum load power Pomin = 0 W.
Also, results obtained show that fs = 141 kHz to 173.5
kHz, Afs =32.5 kHz, TON = 4.7 us to 3.3 us, and TOFF =
3.0 \is to 2.5 us with the commercial alternating-current
power supply input of an alternating input voltage VAC =
230 V corresponding to an AC 200 V system with respect to
the load variation from the maximum load power Pomax =
200 W to the minimum load power Pomin = 0 W. AfsA is 75
kHz (= 173.5 kHz - 98.5 kHz).
Since such characteristics are obtained, the fourth
embodiment also achieves a wide range capability by only
switching frequency control.
As for AC—>DC power conversion efficiency (rjAC->DC),
T) AC—»DC = 93.1% at the alternating input voltage VAC =
100 V and the maximum load power Pomax = 200 W. This
represents an improvement of 0.8 percentage points over
the power conversion efficiency of the conventional power
supply circuit. Alternating-current input power is
reduced by 1.9 W. At the alternating input voltage VAC =
230 V and the maximum load power Pomax = 200 W, TJ AC~>DC =
90.2%.
FIG. 11 shows an example of configuration of a
power supply circuit according to a fifth embodiment.
The secondary side rectifier circuit of the power
supply circuit shown in FIG. 11 is formed as a voltage
doubler full-wave rectifier circuit.
A secondary winding N2 in this case is provided
with a center tap to be divided into secondary winding
parts N2A and N2B with the center tap as a boundary. The
secondary winding parts N2A and N2B have the same number
of turns.
A secondary side parallel resonant capacitor C2 for
forming a secondary side parallel resonant circuit is
connected in parallel with the whole of the secondary
winding N2. In addition, rectifier diodes Dol, Do2, Do3,
and Do4, a capacitor CIO, and a smoothing capacitor Co
are connected to the secondary winding N2 as follows.
An end part on the secondary winding part N2A side
of the secondary winding N2 is connected to a point of
connection between the anode of the rectifier diode Dol
and the cathode of the rectifier diode Do2. An end part
on the secondary winding part N2B side of the secondary
winding N2 is connected to a point of connection between
the anode of the rectifier diode Do3 and the cathode of
the rectifier diode Do4.
The center tap of the secondary winding N2 is
connected to the positive electrode terminal of the
capacitor CIO. The negative electrode terminal of the
capacitor CIO is connected to a point of connection
between the respective anodes of the rectifier diodes Dol
and Do3 as well as a secondary side ground. The cathodes
of the rectifier diodes Dol and Do3 are connected to the
positive electrode terminal of the smoothing capacitor Co.
The negative electrode terminal of the smoothing
capacitor Co is connected to the secondary side ground.
The thus formed voltage doubler full-wave rectifier
circuit can be considered to be a combination of a first
voltage doubler half-wave rectifier circuit including the
secondary winding part N2A, the rectifier diodes Dol and
Do2, the capacitor CIO, and the smoothing capacitor Co
and a second voltage doubler half-wave rectifier circuit
including the secondary winding part N2B, the rectifier
diodes Do3 and Do4, the capacitor CIO, and the smoothing
capacitor Co. The first voltage doubler half-wave
rectifier circuit and the second voltage doubler halfwave
rectifier circuit in this case share the capacitor
CIO and the smoothing capacitor Co.
In a half-wave period in which a secondary winding
voltage V2 is of negative polarity at the end part on the
secondary winding part N2A side, the first voltage
doubler half-wave rectifier circuit passes a rectified
current through a path from the secondary winding part
N2A to the capacitor CIO to the rectifier diode Do2 to
the secondary winding part N2A to charge the capacitor
CIO. The first voltage doubler half-wave rectifier
circuit thereby generates a potential corresponding to
once the level of the voltage induced in the secondary
winding part N2A as a voltage across the capacitor CIO.
Then, in a next half-wave period in which the secondary
winding voltage V2 is of positive polarity at the end
part on the secondary winding part N2A side, the first
voltage doubler half-wave rectifier circuit passes a
rectified current through a path from the secondary
winding part N2A to the rectifier diode Dol to the
smoothing capacitor Co to the capacitor CIO to the
secondary winding part N2B. At this time, the smoothing
capacitor Co is charged by a potential obtained by
superimposing the voltage across the capacitor CIO on the
potential (V2) of the secondary winding part N2A.
The second voltage doubler half-wave rectifier
circuit passes a rectified current through a path from
the secondary winding part N2B to the capacitor CIO to
the rectifier diode Do4 to the secondary winding part N2B
to charge the capacitor CIO in the half-wave period in
which the secondary winding voltage V2 is of negative
polarity at the end part on the secondary winding part
N2B side. Then, in the next half-wave period in which the
secondary winding voltage V2 is of positive polarity at
the end part on the secondary winding part N2B side, the
second voltage doubler half-wave rectifier circuit passes
a rectified current through a path from the secondary
winding part N2B to the rectifier diode Do3 to the
smoothing capacitor Co to the capacitor CIO to the
secondary winding part N2B. Thereby the smoothing
capacitor Co is charged by a potential obtained by
superimposing the voltage across the capacitor CIO on the
potential (equal to V2) of the secondary winding part N2B,
As a result of such an operation, the charging of
the smoothing capacitor Co by the superimposed potential
of the secondary winding part N2A and the capacitor CIO
and the charging of the smoothing capacitor Co by the
superimposed potential of the secondary winding part N2B
and the capacitor CIO are alternately performed in the
positive and negative half cycles of the secondary
winding voltage V2. Thereby a secondary side directcurrent
output voltage Eo as a rectified and smoothed
voltage having a level corresponding to twice the level
of the voltage induced in the secondary winding part N2A
or N2B is obtained as a voltage across the smoothing
capacitor Co.
The thus formed power supply circuit according to
the fifth embodiment is formed by selecting substantially
the same principal parts as in the fourth embodiment, for
example, so that the coupling coefficient k of an
isolated converter transformer PIT is set to k ^ 0.7 or
lower. Thereby a wide range capability is realized by
only switching frequency control. The resonant frequency
fol of a primary side parallel resonant circuit and the
resonant frequency fo2 of the secondary side parallel
resonant circuit are set so as to obtain a certain value
or higher as AC—>DC power conversion efficiency. As a
result, as in the foregoing embodiments, the waveform of
a switching current IQ1 at heavy load has a peak level
suppressed at a time of turning off a switching device Ql,
It is to be noted that the present invention is not
limited to the configurations shown as the foregoing
embodiments. For example, different configurations are
conceivable for details of the circuit configuration of
the primary side voltage resonant converter and the
configuration of the secondary side rectifier circuit
including the secondary side parallel resonant circuit.
In addition, a device other than a MOS-FET may be
selected as the switching device. Further, while the
foregoing embodiments include an externally excited
switching converter, the present invention is applicable
to cases where the switching converter is formed as a
self-excited switching converter.
It should be understood by those skilled in the art
that various modifications, combinations, subcombinations
and alterations may occur depending on
design requirements and other factors insofar as they are
within the scope of the appended claims or the
equivalents thereof.

We claim:
1. A switching power supply circuit comprising:
rectifying and smoothing means (Di, Ci) for obtaining a direct-current input voltage by rectifying and smoothing an alternating input voltage;
switching means formed with a switching device (Ql) supplied with said direct-current input voltage and performing switching operation;
switching-driving means (2) for switching-driving said switching device;
an isolated converter transformer (PIT) formed by at least winding a primary winding (Nl) supplied with a switching output obtained by the switching operation of said switching means and a secondary winding (N2) in which an alternating voltage is induced by the switching output obtained in said primary wining (Nl);
a primary side parallel resonant circuit formed by a leakage inductance component (LI) of the primary winding (Nl) of said isolated converter transformer (PIT) and a capacitance of a primary side parallel resonant capacitor (Cr), the primary side parallel resonant circuit converting operation of said switching means into a voltage resonant type operation;
a secondary side parallel resonant circuit formed by a leakage inductance component (L2) of said secondary winding of said isolated converter transformer (PIT) and a capacitance of a secondary side parallel resonant capacitor (C2) connected in parallel with the secondary winding
(N2);

secondary side direct-current output voltage generating means for generating a secondary side direct-current output voltage by performing a rectifying operation on the alternating voltage induced in the secondary winding (N2) of said isolated converter transformer (PIT) and input to the secondary side direct-current output voltage generating means and smoothing a rectified output resulting from the rectifying operation by a secondary side smoothing capacitor (Co); and
constant-voltage control means (1) for performing constant- voltage control on said secondary side direct-current output voltage by varying switching frequency of said switching means by controlling said switching-driving means according to level of said secondary side direct-current output voltage;
characterized in that a coupling coefficient fog loose coupling between a primary side and a secondary side of said isolated converter transformer (PIT) is set such that an electromagnetic coupling type resonant circuit formed with said primary side parallel resonant circuit and said secondary side parallel resonant circuit has a unimodal characteristic as characteristic of output in response to an input of a frequency signal having said switching frequency, and
a resonant frequency of said primary side parallel resonant circuit and a resonant frequency of said secondary side parallel resonant circuit are set so as to obtain power conversion efficiency at a certain level and higher under predetermined load conditions at least.