SWITCHING POWER SUPPLY CIRCUIT
CROSS REFERENCES TO RELATED APPLICATIONS
The present invention contains subject matter
related to Japanese Patent Application JP 2005-064482
filed with the Japanese Patent Office on March 8, 2005,
the entire contents of which being incorporated herein by
reference.
BACKGROUND OF THE INVENTION
The present invention relates to a switching power
supply circuit including a voltage resonant converter.
As types of a so-called soft switching power supply
that employs a resonant converter, a current resonant
type and a voltage resonant type have been widely known.
Currently, half-bridge connected current resonant
converters formed of a two-transistor switching element
have been widely employed since they can easily be put
into practical use.
However, since characteristics of high-breakdownvoltage
switching elements are currently being improved
for example, problems about breakdown voltage associated
with putting voltage resonant converters into practical
use are being cleared up. Furthermore, it is known that a
single-ended voltage resonant converter formed of onetransistor
switching element is advantageous over a onetransistor
current resonant forward converter with regard
to input feedback noises and noise components of a DC
output voltage line.
Fig. 9 illustrates one configuration example of a
switching power supply circuit including a single-ended
voltage resonant converter.
In the switching power supply circuit of Fig. 9, a
voltage from a commercial alternating-current power
supply AC is rectified and smoothed by a rectifying and
smoothing circuit formed of a bridge rectifier circuit Di
and a smoothing capacitor Ci, to thereby produce a
rectified and smoothed voltage Ei as the voltage across
the smoothing capacitor Ci.
The lines from the commercial power supply AC are
provided with a noise filter that includes a pair of
common mode choke coils CMC and two across-line
capacitors CL, and removes common mode noises.
The rectified and smoothed voltage Ei is input to
the voltage resonant converter as a DC input voltage. The
voltage resonant converter has a single-ended
configuration including one-transistor switching element
Ql as described above. The voltage resonant converter in
this circuit is separately excited. Specifically, th«
switching element Ql formed of a MOS-FET is switch-driven
by an oscillation and drive circuit 2.
A body diode DD of the MOS-FET is connected in
parallel to the switching element Ql. In addition, a
primary-side parallel resonant capacitor Cr is connected
in parallel to the channel between the source and drain
of the switching element Ql.
The primary-side parallel resonant capacitor Cr and
the leakage inductance LI of a primary winding Nl in an
isolation converter transformer PIT form a primary-side
parallel resonant circuit (voltage resonant circuit).
This primary-side parallel resonant circuit offers
voltage resonant operation as the switching operation of
the switching element Ql.
In order to switch-drive the switching element Ql,
the oscillation and drive circuit 2 applies a gate
voltage as a drive signal to the gate of the switching
element Ql. Thus, the switching element Ql implements
switching operation with the switching frequency
dependent upon the cycle of the drive signal.
The isolation converter transformer PIT transmits
switching outputs from the switching element Ql to the
secondary side.
The isolation converter transformer PIT is
constructed of an EE-core that is formed by combining Ecores
composed of a ferrite material for example.
Furthermore, the primary winding Nl and a secondary
winding N2 are wound around the center magnetic leg of
the EE-core/ with the winding part being divided into the
primary side and secondary side.
In addition, a gap with a length of about 1.0 mm is
provided in the center magnetic leg of the EE-core in the
isolation converter transformer PIT, so that a coupling
coefficient k of about 0.80 to 0.85 is obtained between
the primary side and the secondary side. When the
coupling coefficient k has such a value, the coupling
degree between the primary and secondary sides may be
regarded as loose coupling, and thus it is difficult to
obtain the saturation state. The value of the coupling
coefficient k is a factor in setting the leakage
inductance (11).
One end of the primary winding Nl in the isolation
converter transformer PIT is interposed between the
switching element Ql and the positive electrode of the
smoothing capacitor Ci. Thus, the transmission of
switching outputs from the switching element Ql is
allowed. In the secondary winding N2 of the isolation
converter transformer PIT, an alternating voltage induced
by the primary winding Nl is generated.
In this circuit, a secondary-side parallel resonant
capacitor C2 is connected in parallel to the secondary
winding N2. Thus, the leakage inductance L2 of the
secondary winding N2 and the capacitance of the
secondary-side parallel resonant capacitor C2 form a
secondary-side parallel resonant circuit (voltage
resonant circuit).
Furthermore, as shown in Fig. 9, connected to this
secondary-side parallel resonant circuit are a rectifier
diode Ool and a smoothing capacitor Co, to thereby form a
half-wave rectifier circuit. This half-wave rectifier
circuit produces, as the voltage across the smoothing
capacitor Co, a secondary-side DC output voltage Eo with
the same level as that of an alternating voltage V2
induced in the secondary winding N2 (secondary-side
parallel resonant circuit). The secondary-side DC output
voltage Eo is supplied to a load, and is input to a
control circuit 1 as a detected voltage for constantvoltage
control.
The control circuit 1 detects the level of the
secondary-side DC output voltage Eo input as a detected
voltage, and then inputs the obtained detection output to
the oscillation and drive circuit 2.
According to the level of the secondary-side DC
output voltage Eo indicated by the input detection output/
the oscillation and drive circuit 2 controls the
switching operation of the switching element Ql so that
the secondary-side DC output voltage Eo is kept constant
at a certain level. That is, the oscillation and drive
circuit 2 produces and outputs a drive signal for
achieving intended switching operation. Thus,
stabilization control of the secondary-side DC output
voltage Eo is achieved.
Figs. 10A, 10B and 11 show results of experiments
on the power supply circuit in Fig. 9. In the experiments,
major parts of the power supply circuit of Fig. 9 were
designed to have the following characteristics, as
conditions for an AC input voltage VAC of 100 V, which
corresponds to an AC 100 v-system input.
As the switching element Ql, a product of which
breakdown voltage was 900 V was selected. As the
secondary-side rectifier diode Dol, a product of which
breakdown voltage was 600 V was selected.
The core of the isolation converter transformer PIT
employed an EER-35 core, and a gap in the center magnetic
leg thereof was designed to have a gap length of 1 mm.
The numbers of turns T of the primary winding Nl and the
secondary winding N2 were both set to 43 T. The coupling
coefficient k of the isolation converter transformer PIT
was set to 0.81.
The capacitances of the primary-side parallel
resonant capacitor Cr and the secondary-side parallel
resonant capacitor C2 were set to 6800 pF and 0.01 uF,
respectively. Accordingly, the resonant frequency fol of
the primary-side parallel resonant circuit was set to 175
kHz, and the resonant frequency fo2 of the secondary-side
parallel resonant circuit was set to 164 kHz.
The rated level of the secondary-side DC output
voltage Eo was 135 V. The allowable load power range was
from the maximum load power Pomax of 200 W to the minimum
load power Pomin of 0 W.
Figs. 10A and 10B are waveform diagrams showing the
operation of major parts in the power supply circuit in
Fig. 9, while reflecting the switching cycle of the
switching element Ql. Fig. 10A shows a switching voltage
VI, a switching current IQ1, a primary winding current II,
a secondary winding voltage V2, a secondary winding
current 12, and a secondary-side rectified current ID1,
when the load power is the maximum load power Pomax of
200 W. Fig. 10B shows the switching voltage VI, the
switching current IQl, the primary winding current II,
the secondary winding voltage V2, the secondary winding
current 12, and the secondary-side rectified current ID1,
when the load power is the minimum load power Pomin of 0
W.
The switching voltage VI is the voltage obtained
across the switching element Ql. The switching voltage VI
has a waveform like those in Figs. IDA and 10B.
Specifically, the voltage level is at 0 level during the
period TON when the switching element Ql is in the onstate,
while a sinusoidal voltage resonant pulse is
obtained during the period TOFF when it is in the offstate.
This voltage resonant pulse waveform of the
switching voltage VI indicates that the operation of the
primary-side switching converter is voltage resonant
operation.
The peak level of the voltage resonant pulse of the
switching voltage VI was 550 Vp when the load power was
the maximum load power Pomax of 200 W and the input
voltage VAC was 100 V (AC 100 V-system), and was 800 Vp
when the load power was the maximum load power Pomax of
200 W and the input voltage VAC was 264 V (AC 200 Vsystem).
In order to respond to these peak levels of the
voltage resonant pulse, a product of which breakdown
voltage was 900 V was used as the switching element Ql as
described above.
The switching current IQ1 is the current flowing
through the switching element Ql (and the body diode DD).
During the period TOFF, the switching current IQ1 is at 0
level. During the period TON, the switching current IQ1
with a certain waveform like illustrated one is obtained.
Specifically, at the time of turn on of the switching
element Ql, the switching current IQ1 flows through the
body diode DD in the forward direction thereof, and thus
the switching current IQ1 has the negative polarity.
After the turn on, the polarity is inverted and the
switching current IQ1 flows between the drain and source
of the switching element Ql. The current value increases
with time until turn off of the switching element Ql.
Therefore, the peak level of the switching current IQ1 is
obtained at the timing of the turn off.
The primary winding current II flowing through the
primary winding Nl is the current resulting from the
synthesis between the current flowing as the switching
current IQ1 during the period TON and the current flowing
to the primary-side parallel resonant capacitor Cr during
the period TOFF. Thus, the primary winding current II has
a waveform like those shown in Figs. IDA and 10B.
As the operation of the secondary-side rectifier
circuit, the rectified current IDl flows through the
rectifier diode Dol with having a certain waveform like
that shown in Fig. 10A when the load power is the maximum
load power of 200 W. Specifically, the peak level of the
rectified current IDl is obtained at the time of turn on
of the rectifier diode Dol, and then the level gradually
decreases toward 0 as shown in the waveform of Fig. IDA.
During the period when the rectifier diode Dol is in the
off-state, the level of the rectifier diode Dol is at 0.
In contrast, when the load power is the minimum load
power Pomin of 0 W, the current level is invariably at 0
even during the ON period of the rectifier diode Dol.
The secondary winding voltage V2 is obtained in the
parallel circuit of the secondary winding N2 and the
secondary-side parallel resonant capacitor C2. During the
period when the secondary-side rectifier diode Dol
conducts, the secondary winding voltage v2 is clamped at
the level of the secondary-side DC output voltage Eo.
During the OFF period of the secondary-side rectifier
diode Dol, the secondary winding voltage V2 shows a
sinusoidal waveform of the negative polarity. The
secondary winding current 12 flowing through the
secondary winding N2 is the current resulting from the
synthesis between the rectified current IDl and the
current flowing through the secondary-side parallel
resonant circuit (N2 (L2)//C2). The secondary winding
current 12 has the illustrated waveform for example.
Fig. 11 shows, as a function of load, the switching
frequency fs, the ON period TON, the OFF period TOFF, and
the AC to DC power conversion efficiency (nAC -.DC) of
the power supply circuit shown in Fig. 9.
As for the AC to DC power conversion efficiency
{nAC - DC), an efficiency of 90% or more is achieved when
the load power Po is in the range of 100 W to 200 W. It
is known that a particularly single-ended voltage
resonant converter, of which switching element Ql is
formed of one transistor, offers favorable power
conversion efficiencies.
In addition, the switching frequency fs, the ON
period TON, and the OFF period TOFF in Fig. 11 indicate
the switching operation of the power supply circuit of
Fig. 9 as the characteristic of constant-voltage control
against load variation. In the circuit, the switching
frequency fs is controlled so that the switching
frequency increases as the load becomes lighter. As for
the ON and OFF periods TON and TOFF, the period TOFF is
almost constant independently of load variation. In
contrast, the period TON is shortened as the load becomes
lighter. That is, the power supply circuit in Fig. 9
varies and controls the switching frequency so as to
shorten the ON period TON as the load becomes lighter
while keeping the OFF period TOFF constant.
This variation control of the switching frequency
allows variation of the inductive impedance that is due
to the existence of the primary-side parallel resonant
circuit and secondary-side parallel resonant circuit.
This inductive impedance variation leads to a change of
the amount of transmitted power from the primary side to
the secondary side, and a change of the amount of
transmitted power from the secondary-side parallel
resonant circuit to a load. As a result, the level of the
secondary-side DC output voltage Eo is varied. Thus, the
secondary-side DC output voltage Eo is stabilized.
Fig. 12 schematically shows the constant-voltage
control characteristic of the power supply circuit shown
in Fig. 9, based on the relationship between the
switching frequency fs (kHz) and the secondary-side DC
output voltage Eo.
When the resonant frequencies of the primary-side
and secondary-side parallel resonant circuits are defined
as fol and fo2, respectively, the secondary-side parallel
resonant frequency fo2 is lower than the primary-side
parallel resonant frequency fol in the circuit of Fig. 9
as described above.
The characteristic curves in Fig. 12 are based on
these resonant frequencies/ and based on an assumption of
constant-voltage control characteristics relating to the
switching frequency fs, obtained for a certain constant
AC input voltage VAC. Specifically, Characteristic curves
A and B indicate the constant-voltage control
characteristics obtained when the load power is the
maximum load power Pomax and the minimum load power Pomin,
respectively, based on the resonant impedance
corresponding to the resonant frequency fol of the
primary-side parallel resonant circuit. Characteristic
curves C and D indicate the constant-voltage control
characteristics obtained when the load power is the
maximum load power Pomax and the minimum load power Pomin,
respectively, based on the resonant impedance
corresponding to the resonant frequency fo2 of the
secondary-side parallel resonant circuit.
When a circuit includes a primary-side parallel
resonant circuit and a secondary-side parallel resonant
circuit like the circuit in Fig. 9, the center resonant
frequency fo exists between the resonant frequencies fol
and fo2. Characteristic curves E and F indicate the
resonant impedance characteristics, based on the
relationship between the center resonant frequency to and
the switching frequency fs, obtained when the load power
is the maximum load power Pomax and the minimum load
power Pomin, respectively.
In a voltage resonant converter including a
secondary-side parallel resonant circuit, the level of
the secondary-side DC output voltage Eo is determined
depending on the resonant impedance characteristic with
respect to the center resonant frequency fo, as a
function of the switching frequency fs. Furthermore, the
voltage resonant converter in Fig. 9 employs a lower-side
control method, in which the switching frequency fs is
varied and controlled in a frequency range lower than the
center resonant frequency fo.
When, under the characteristics corresponding to
the center resonant frequency fof indicated by
Characteristic curves E and F in Fig. 12, constantvoltage
control in which the target value of the output
voltage is the rated level of the secondary-side DC
output voltage Eo (135 v, in the circuit of Fig. 9) is
intended with use of switching frequency control based on
lower-side control, the variable range of the switching
frequency fs required for the constant-voltage control
(requisite control range) is the range indicated by Afs.
That is, in the frequency range indicated by Afs, the
switching frequency is varied to a requisite value
according to load variation. Thus, the secondary-side DC
output voltage Eo is controlled so that it is kept at a
rated level tg.
An example of the conventional power supply
circuits is disclosed in Japanese Patent Laid-open Mo.
2000-152617.
In step with diversification of various electronic
apparatuses, demands have been increasing for so-called
wide-range compatible power supply circuits that operate
in response both to commercial AC voltage inputs of the
AC 100-V system and AC 200-V system.
The power supply circuit in Fig. 9 operates so as
to stabilize the secondary-side DC output voltage Eo by
switching frequency control as described above. The
variable range {requisite control range) of the switching
frequency required for the voltage stabilization is
indicated by Afs described in Fig. 12.
The power supply circuit of Fig. 9 is designed to
respond to load variation of a comparatively wide
variation range from 200 W to 0 W. In the power supply
circuit of Fig. 9, an actual requisite control range of
the switching frequency fs for this load variation
condition is from 117.6 kHz to 208.3 kHz. That is, the
range Afs is 96.7 kHz, and this range is comparatively
wide.
It is obvious that when the level of the AC input
voltage VAC is changed/ the level of the secondary-side
DC output voltage Eo also varies. That is, the level of
the secondary-side DC output voltage Eo varies depending
on the level of the AC input voltage VAC.
Therefore, the level variation of the secondaryside
DC output voltage Eo is larger when the AC input
voltage varies in a wide range including both the AC 100-
v system and AC 200-V system, compared with the level
variation when the AC input voltage varies only in a
single range of the AC 100-v system or AC 200-V system
for example. In order to ensure constant-voltage control
operation compatibly with the wide level variation of the
secondary-side DC output voltage Eo, the requisite
control range of the switching frequency needs to be
widened from the above-described range of 117.6 kHz to
208.3 kHz so that higher frequencies are also covered.
However, in a present 1C (the oscillation and drive
circuit 2} for driving switching elements, the upper
i e.
limit of a possible drive frequency is about 200 kHz.
Even if an 1C is developed so that an 1C can drive
switching elements with such a high frequency, the
driving of switching elements with a high frequency leads
to a significant decrease of the power conversion
efficiency. It therefore is substantially impossible to
put this 1C into practical use for a power supply circuit.
Ae described above, it is very difficult to achieve
a wide-range compatible power supply circuit by use of
the configuration shown in Fig. 9 for example.
The power supply circuit shown in Fig. 9 includes a
single-ended voltage resonant converter on its primary
side. The power supply circuit with such a configuration
tends to offer advantages for achieving a high power
conversion efficiency as described above. However, in
consideration of recent energy circumstances and
environmental circumstances for example, electronic
apparatuses have been required to have a further higher
power conversion efficiency characteristic. Accordingly,
a power supply circuit itself incorporated in an
electronic apparatus has been required to have a further
improved power conversion efficiency.
SUMMARY OF THE INVENTION
In consideration of the above-described problems/
one embodiment of the present invention provides a
switching power supply circuit having the following
configuration.
Specifically, the switching power supply circuit
includes a switching unit that includes a main switching
element supplied with a direct-current (DC) input voltage
and implementing switching, and a switching drive unit
that switch-drives the main switching element.
The circuit also includes an isolation converter
transformer that includes at least a primary winding and
a secondary winding that are wound around the isolation
converter transformer. The primary winding is supplied
with a switching output arising from switching operation
of the switching unit. The switching output supplied to
the primary winding induces an alternating voltage in the
secondary winding.
In addition, the circuit also includes a primaryside
parallel resonant circuit that is formed by at least
a leakage inductance component including the primary
winding of the isolation converter transformer and the
capacitance of a primary-side parallel resonant capacitor.
The primary-side parallel resonant circuit offers voltage
resonant operation as operation of the switching unit.
The circuit further includes a secondary-side
parallel resonant circuit that arises from connecting of
a secondary-side parallel resonant capacitor in parallel
to the secondary winding of the isolation converter
transformer, and is formed of a leakage inductance
component including the secondary winding and the
capacitance of the secondary-side parallel resonant
capacitor.
Furthermore, the circuit also includes a secondaryside
series resonant circuit that arises from connecting
of a secondary-side series resonant capacitor in series
to the secondary winding of the isolation converter
transformer, and is formed of a leakage inductance
component including the secondary winding and the
capacitance of the secondary-side series resonant
capacitor.
The circuit also includes a secondary-side
rectifying and smoothing unit that is supplied with the
alternating voltage induced in the secondary winding of
the isolation converter transformer, and implements
rectifying operation to thereby produce a secondary-side
DC output voltage, and a constant-voltage control unit
that implements constant-voltage control of the
secondary-side DC output voltage by controlling the
switching drive unit according to the level of the
secondary-side DC output voltage to thereby vary the
switching frequency of the switching unit.
Moreover, the circuit further includes an active
clamp circuit having an auxiliary switching element and
setting an ON period within a period during which the
main switching element is in the off-state. The auxiliary
switching element is in the on-state in the ON period.
The active clamp circuit causes charging and discharging
currents to flow through the auxiliary switching element
during the ON period. The charging and discharging
currents originally flow to the primary-side parallel
resonant capacitor in the absence of the active clamp
circuit.
In the circuit, the isolation converter transformer
is designed to provide such a certain coupling
coefficient between a primary side and a secondary side
as to offer loose coupling therebetween so that an
electromagnetically coupled resonant circuit formed by at
least the primary-side parallel resonant circuit and the
secondary-side parallel resonant circuit has a unimodal
characteristic as an output characteristic with respect
to an input of a frequency signal having the switching
frequency. In addition, at least the resonant frequency
of the primary-side parallel resonant circuit, the
resonant frequency of the secondary-side parallel
resonant circuit and the resonant frequency of the
secondary-side series resonant circuit are designed so
that a power conversion efficiency above a certain level
is obtained under a certain load condition.
The power supply circuit according to the
embodiment has a basic configuration in which a voltage
resonant converter is provide on the primary side and a
secondary-side parallel resonant circuit and a secondaryside
series resonant circuit are provided on the
secondary side. This configuration can be regarded as a
configuration including a parallel resonant circuit on
each of the primary and secondary sides. From this
viewpoint, the configuration has a coupled resonant
circuit due to electromagnetic coupling by an isolation
converter transformer. In addition, the isolation
converter transformer is designed to offer loose coupling
based on a certain coupling coefficient. Thus, a sharp
unimodal characteristic can be obtained as an output
characteristic with respect to a frequency signal
(switching output) having a switching frequency, which is
an input to the coupled resonant circuit. As a result,
the switching frequency variable range (requisite control
range) required for stabilizing a secondary-side DC
output voltage can be narrowed.
Moreover, the power supply circuit according to the
embodiment can be regarded also as a voltage resonant
converter that is provided with a series resonant circuit
on the secondary side. Typically/ the voltage resonant
converter having a series resonant circuit on the
secondary side has a favorable power conversion
efficiency characteristic, but tends to cause abnormal
operation in which zero volt switching (2VS) operation
fails to be implemented, when the load is an intermediate
load. In order to address this respect, the embodiment
designs the isolation converter transformer to offer a
loose coupling state above a certain extent, to thereby
allow suppression and elimination of the abnormal
operation. Therefore, ZV$ operation can be maintained in
the entire allowable load power region.
Furthermore, an active clamp circuit is provided to
thereby cause charging and discharging currents that
originally should flow to a primary-side parallel
resonant capacitor to flow through an auxiliary switching
element. This operation of the active clamp circuit
suppresses the peak level of a resonant voltage pulse
that is the voltage across the primary-side parallel
resonant capacitor.
As described above/ the present invention narrows
the variable control range (requisite control range) of a
switching frequency of a voltage resonant converter,
required for constant-voltage control. Thus, regarding a
voltage resonant switching converter, a wide-range
compatible configuration can easily be achieved simply by
implementing switching frequency control.
In order to achieve such a wide-range compatible
configuration, it is enough to form a basic configuration
in which the isolation converter transformer of the
voltage resonant converter having a secondary-side
resonant circuit is designed to offer a requisite
coupling coefficient. Therefore, a wide-range compatible
configuration can be achieved without involving increases
of costs, circuit size, circuit weight and so on due to
an increase of the number of parts.
In addition, since the power supply circuit is
formed of a combination between a voltage resonant
converter and a secondary-side series resonant circuit,
ZVS operation is maintained in the entire allowable load
power range, which provides a favorable power conversion
efficiency characteristic suitable for practical use of
the power supply circuit.
Moreover, since the peak level of a resonant
voltage pulse is suppressed by an active clamp circuit,
low-breakdown-voltage products can be selected as parts
elements typified by a main switching element. Thus, the
performances of the parts elements can be improved, which
allows enhancement of reliability of a power supply
circuit, a cost decrease, and reduction of size and
weight of the circuit.
BRIEF DESCRIPTION OF THE DRAWINGS
Fig. 1 is a circuit diagram illustrating a
configuration example of a power supply circuit according
to a first embodiment of the present invention;
Fig. 2 is a diagram illustrating a configuration
example of an isolation converter transformer included in
a power supply circuit of one embodiment of the
invention;
Figs. 3A and 3B are waveform diagrams showing the
operation of major parts in the power supply circuit of
the first embodiment, with reflecting the corresponding
switching frequency;
Fig. 4 is a diagram showing, as a function of load,
the variation characteristics of the AC to DC power
conversion efficiency, the switching frequency, and the
peak level of a voltage resonant pulse, regarding the
power supply circuit of the first embodiment;
Fig. 5 is a diagram illustrating the constantvoltage
control characteristic of a power supply circuit
of one embodiment of the invention;
Tig. 6 is a diagram illustrating, as the constantvoltage
control operation of a power supply circuit of
one embodiment of the invention, the switching frequency
control range (requisite control range) for addressing AC
input voltage variation and load variation;
Fig. 7 is a circuit diagram illustrating a
configuration example of a power supply circuit according
to a second embodiment of the present invention*
Fig. 8 is a circuit diagram illustrating a
configuration example of a power supply circuit according
to a third embodiment of the present invention;
Fig. 9 is a circuit diagram illustrating a
configuration example of a power supply circuit as a
conventional circuit;
Figs. IDA and 10B are waveform diagrams showing the
operation of major parts in the power supply circuit
shown in Fig. 9;
Fig. 11 is a diagram showing, as a function of load,
the variation characteristics of the AC to DC power
conversion efficiency, the switching frequency, and the
ON period and OFF period of a switching element,
regarding the power supply circuit in Fig. 9; and
Fig. 12 is a diagram conceptually showing the
constant-voltage control characteristic of a conventional
power supply circuit.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
Fig. 1 is a circuit diagram illustrating a
configuration example of a power supply circuit according
to a first embodiment of the present invention, as one of
best modes (embodiments) for carrying out the invention.
The power supply circuit in Fig. 1 includes, as its basic
configuration, a single-ended voltage resonant switching
converter.
In the switching power supply circuit in Fig. 1,
the lines from a commercial alternating-current power
supply AC are provided with a pair of common mode choke
coils CMC and two across-line capacitors CL. These common
mode choice coils CMC and the across-line capacitors CL
form a noise filter that removes common mode noises
included in the lines from the commercial alternatingcurrent
power supply AC.
A voltage from the commercial alternating-current
power supply AC (an AC input voltage VAC) is rectified by
9. bridge rectifier circuit Di, and the rectified output
is charged in the smoothing capacitor Ci. Thus, a
rectified and smoothed voltage Ei is obtained as the
voltage across the smoothing capacitor Ci. The rectified
and smoothed voltage Ei serves as a DC input voltage for
the switching converter at the subsequent stage.
In Fig. 1, the switching converter that receives
the rectified and smoothed voltage Ei as a DC input
voltage and implements switching operation is formed as a
single-ended voltage resonant converter that includes/
for example, one-transistor switching element Ql. In this
circuit, a high-breakdown-voltage MOS-FET is selected as
the switching element Ql. The voltage resonant converter
in this circuit is separately excited. Specifically, an
oscillation and drive circuit 2 switch-drives the
switching element.
Applied to the gate of the switching element Ql is
a switching drive signal (voltage) output from the
oscillation and drive circuit 2.
The drain of the switching element Ql is coupled to
the winding-start end of a primary winding Nl in an
isolation converter transformer PIT to be described later.
The winding-finish end of the primary winding Nl is
coupled to the positive electrode of the smoothing
capacitor Therefore, the DC input voltage (Ei) is
supplied via the serially connected primary winding Nl to
the switching element Ql. The source of the switching
element Ql is coupled to the primary-side ground.
Since a MOS-FET is used as the switching element Ql,
the switching element Ql incorporates a body diode DD so
that the body diode DD is connected in parallel to the
channel between the source and drain of the switching
element Ql. The anode of the body diode DD is connected
to the source of the switching element Ql, and the
cathode thereof is connected to the drain of the
switching element Ql. The body diode DD forms the path of
a switching current in the reverse direction, arising due
to the ON/OFF operation (switching operation) of the
switching element Ql.
In addition, a primary-side parallel resonant
capacitor Cr is connected in parallel to the channel
between the drain and source of the switching element Ql.
The capacitance of the primary-side parallel
resonant capacitor Cr and the leakage inductance LI of
the primary winding Nl in the isolation converter
transformer PIT form a primary-side parallel resonant
circuit (voltage resonant circuit) for a switching
current flowing through the switching element Ql. The
resonant operation of this primary-side parallel resonant
circuit offers voltage resonant operation as the
switching operation of the switching element Ql. In
response to this operation, during the OFF period of the
switching element Ql, a sinusoidal voltage resonant pulse
is obtained as a switching voltage VI, which is the
voltage across the switching element Ql (voltage between
the drain and source).
In order to drive the switching element Ql by
separate excitation for example, the oscillation and
drive circuit 2 includes an oscillation circuit that
generates an oscillation signal. Based on the oscillation
signal, the oscillation and drive circuit 2 produces a
drive signal that is a gate voltage for switch-driving
the MOS-FET, and applies it to the gate of the switching
element Ql, Thus, the switching element Ql continuously
implements ON/OFF operation with the switching frequency
dependent upon the cycle of the drive signal. That is,
the switching element Ql implements switching operation.
The isolation converter transformer PIT transmits
switching outputs from the primary-side switching
converter to the secondary side, while isolating the
primary side from the secondary side in terms of DC
voltage transmission therebetween.
Fig. 2 is a sectional view illustrating a
configuration example of the isolation converter
transformer PIT included in the power supply circuit of
fig. 1.
As shown in Fig. 2, the isolation converter
transformer PIT includes an EE-core (EE-shaped core)
formed by combining E-cores CR1 and CR2 made of a ferrite
material in such a manner that their magnetic legs face
each other.
Furthermore, a bobbin B is provided that is formed
of resin or the like and has such a divided shape that
winding parts on the primary side and secondary side are
independent of each other. The primary winding Nl is
wound around one winding part of the bobbin B. The
secondary winding N2 is wound around the other winding
part. The bobbin B around which the primary-side and
secondary-side windings have been thus wound is fitted to
the EE-core (CR1, CR2), which results in the state in
which the primary-side and secondary-side windings in the
different winding regions are wound around the center leg
of the EE-core. In this manner, the entire structure of
the isolation converter transformer PIT is completed.
In the center leg of the EE-cor«, a gap G having a
gap length of about 2 mm or more is formed as shown in
the drawing. Thus, a loose coupling state is obtained in
which the coupling coefficient k is about 0.7 or less for
example. That is, the degree of loose coupling is further
increased in the isolation converter transformer PIT in
Fig. 2, compared with that in the power supply circuit
shown in Fig. 9 as a conventional technique. The gap G
can be formed by designing the center legs of the E-cores
CR1 and CR2 to be shorter than two outer legs thereof.
As described above, one end of the primary winding
Nl in the isolation converter transformer PIT is coupled
to the drain of the switching element Ql. Thus, switching
outputs from the switching element Ql are transmitted to
the primary winding Nl, and an alternating voltage arises
in the primary winding Nl.
On the secondary side of the isolation converter
transformer PIT, an alternating voltage induced by the
primary winding Nl is generated in the secondary winding
N2.
Connected in parallel to the secondary winding N2
is a secondary-side parallel resonant capacitor C2. Thus,
the leakage inductance L2 of the secondary winding N2 and
the capacitance of the secondary-side parallel resonant
capacitor C2 form a secondary-side parallel resonant
circuit. This secondary-side parallel resonant circuit
implements resonant operation in response to rectifying
operation of a secondary-side rectifier circuit to be
described later. That is, voltage resonant operation is
achieved not only on the primary side but also on the
secondary side.
Furthermore, the present embodiment includes a
secondary-side series resonant capacitor C3. One
electrode of the secondary-side series resonant capacitor
C3 is coupled to the connecting node between the windingfinish
end of the secondary winding N2 and the secondaryside
parallel resonant capacitor C2. The other electrode
of the secondary-side series resonant capacitor C3 is
coupled to the connecting node between the anode and the
cathode of rectifier diodes Dol and Do2f respectively,
that form a secondary-side bridge rectifier circuit to be
described later. This connecting structure leads to the
formation of a series circuit of the secondary winding N2
and the secondary-side series resonant capacitor C3, in a
rectified current path of the secondary-side rectifier
circuit to be described later. Due to this series circuit
configuration, the leakage inductance L2 of the secondary
winding N2 and the capacitance of the secondary-side
series resonant capacitor C3 form a secondary-side series
resonant circuit. This secondary-side series resonant
circuit also implements resonant operation (current
resonant operation) in response to rectifying operation
of the secondary-side rectifier circuit to be described
later.
Thus, on the secondary side of the power supply
circuit in Fig. 1, series resonant operation (current
resonant operation) as well as the above-described
parallel resonant operation (voltage resonant operation)
is achieved.
The secondary-side rectifier circuit in the present
embodiment is formed as a full-wave rectifier circuit
(bridge full-wave rectifier circuit). For that purpose/ a
bridge rectifier circuit formed of four rectifier diodes
Dol, Do2, Do3 and Do4, and one smoothing capacitor Co are
coupled, in the following manner, to the secondary
winding N2 to which the secondary-side parallel resonant
capacitor C2 and the secondary-side series resonant
capacitor C3 are connected as described above.
Specifically, the winding-finish end of the
secondary winding N2 is coupled via the secondary-side
series resonant capacitor C3 to the connecting node
between the anode of the rectifier diode Dol and the
cathode of the rectifier diode Do2. The winding-start end
of the secondary winding N2 is coupled to the connecting
node between the anode of the rectifier diode Do3 and the
cathode of the rectifier diode Do4. The cathodes of the
rectifier diodes Dol and Do3 are coupled to th« positive
electrode of the smoothing capacitor Co. The negative
electrode of the smoothing capacitor Co is connected to
the secondary-side ground. The anodes of the rectifier
diodes Do2 and Do4 are also connected to the secondaryside
ground.
In the thus formed full-wave rectifier circuit,
during the periods of half cycles of one polarity
{hereinafter, sometimes referred to as one half cycles)
of the alternating voltage induced (excited) in the
secondary winding N2, the pair of the rectifier diodes
Dol and Do4 in the bridge rectifier circuit conduct to
thereby charge the rectified current in the smoothing
capacitor Co In contrast, during the periods of half
cycles of the other polarity (hereinafter, sometimes
referred to as the other half cycles) of the alternating
voltage induced in the secondary winding N2, the pair of
the rectifier diodes Do2 and Oo3 conduct to thereby
charge the rectified current in the smoothing capacitor
Co.
Due to these operations, generated across the
smoothing capacitor Co is the secondary-side DC output
voltage Eo having the level equal to that of the
alternating voltage induced in the secondary winding N2.
The thus obtained secondary-side DC output voltage
Eo is supplied to a load (not shown), and is branched and
input to a control circuit I to be described later as a
detected voltage.
Moreover, in association with the rectifying
operation of the full-wave rectifier circuit, voltage
resonant operation by the secondary-side parallel
resonant circuit and current resonant operation by the
secondary-side series resonant circuit are achieved. In
this configuration, the secondary-side rectifier circuit
can be regarded as a circuit that rectifies and smoothes
both resonant outputs from the secondary-side parallel
resonant circuit and secondary-side series resonant
circuit.
The control circuit 1 supplies to the oscillation
and drive circuit 2 a detection output dependent upon a
level change of the input secondary-side DC output
voltage Eo. The oscillation and drive circuit 2 drives
the switching element Ql while varying the switching
frequency according to the detection output input from
the control circuit 1.
This variation control of switching frequency of
the switching element Ql leads to changes of resonant
impedances of the primary and secondary sides of the
power supply circuit. These impedance changes result in a
change of the amount of power transmitted from the
primary winding Nl to the secondary winding N2 in the
isolation converter transformer PIT, and in a change of
the amount of power to be supplied from the secondaryside
rectifier circuit to a load. Thus, operation is
achieved in which the level of the secondary-side DC
output voltage Eo is controlled so that the level
variation thereof is cancelled. That is, stabilization of
the secondary-side DC output voltage Eo is allowed.
On the primary side of the power supply circuit in
Fig. 1, an active clamp circuit 10 is provided.
The active clamp circuit 10 includes an auxiliary
switching element Q2, a clamp capacitor CCL, and a clamp
diode DD2. As the auxiliary switching element Q2, a MOSFET
is selected. The clamp diode DD2 is a body diode
included in the auxiliary switching element Q2. The anode
thereof is connected to the source of the auxiliary
switching element Q2, while the cathode is connected to
the drain of the auxiliary switching element Q2.
Furthermore, the active clamp circuit 10 includes,
as drive circuitry for driving the auxiliary switching
element Q2, a drive winding Ng, a capacitor Cg, a gate
resistor Rg, and a resistor Rl between the gate and
source.
Note that hereinafter, the switching element Ql of
the single-ended voltage resonant converter is referred
to also as a main switching element Ql, for
differentiation from the auxiliary switching element Q2.
The drain of the auxiliary switching element Q2 is
connected to one electrode of the clamp capacitor CCL.
The other electrode of the clamp capacitor CCL is coupled
to the connecting node between the line of the rectified
and smoothed voltage Ei and the winding-finish end of the
primary winding Ml. The source of the auxiliary switching
element Q2 is coupled to the winding-start end of the
primary winding Nl.
That is, in the active clamp circuit 10 of the
present embodiment, the clamp capacitor CCL is connected
in series to the parallel circuit of the auxiliary
switching element Q2 and the clamp diode DD2. In addition,
the thus formed circuit is connected in parallel to the
primary winding Nl of the isolation converter transformer
PIT.
In the drive circuitry for the auxiliary switching
element Q2, the resistor Rl is interposed between the
gate and source of the auxiliary switching element Q2 as
shown in the drawing. In addition, the gate of the
auxiliary switching element Q2 is coupled to the series
circuit of the resistor Rg, the capacitor Cg and the
drive winding Ng. This series circuit and the resistor Rl
form a self-excited oscillation and drive circuit for the
auxiliary switching element Q2. The drive winding Ng is
formed by further turning up the winding-start end of the
primary winding Nl in the isolation converter transformer
PIT. The number of turns of the drive winding Ng is for
example 1 T (turn). Thus, generated in the drive winding
Ng is a voltage induced by an alternating voltage arising
in the primary winding Nl. Due to the relationship
between the winding directions of the primary winding Nl
and the drive winding Ng, the voltage arising in the
drive winding Ng has the polarity opposite to that of the
voltage arising in the primary winding Nl. The actual
operation of the drive winding Ng is ensured as long as
the nuinber of turns thereof is at least 1 T. However, the
number of turns is not limited to 1 T.
As described later in detail/ the active clamp
circuit 10 operates so as to suppress the peak level of a
voltage resonant pulse that is generated during the OFF
period of the main switching element Ql as the switching
voltage (parallel resonant voltage) VI arising across the
parallel circuit of the main switching element Ql and the
primary-side parallel resonant capacitor Cr.
The characteristics of major parts in an actual
power supply circuit having the configuration of Fig.
are as follows for example. Note that 300 W and 0 W (no
load} are the maximum load power Pomax and the minimum
load power Pomin, respectively, in the allowable load
power range of this power supply circuit.
The core of the isolation converter transformer PIT
employs an EER-35 core, and a gap G is designed to have a
gap length of 2.2 nan. The numbers of turns of the primary
winding Nl and the secondary winding N2 are set to 55 T
and 50 T, respectively. The induced voltage per one turn
in the secondary winding N2 is set to about 2.5 V / T or
higher. The leakage inductance LI of the primary winding
Nl is 350 uH, and the leakage inductance L2 of the
secondary winding N2 is 332 uH. According to these
conditions, the coupling coefficient k of the isolation
converter transformer PIT is 0.685,
As is well known, the above-described EER type is
one of types and standards of product cores. It is known
that the types also include an EE type. The sectional
shapes of cores of the EER and EE types are both an EEcharacter
shape. Therefore, the term EE-core in the
present specification encompasses both cores of the EER
and EE types.
As the main switching element Ql, a product of
which breakdown current and voltage are 10 A and 600 V,
respectively, is selected.
The capacitance of the primary-side parallel
resonant capacitor Cr is set to 2200 pF. The capacitance
of the secondary-side parallel resonant capacitor C2 is
set to 8200 pF. The capacitance of the secondary-side
series resonant capacitor C3 is set to 0.022 uF.
The parts in the active clamp circuit 10 are
designed as follows: the clamp capacitor CCL has a
capacitance of 0.056 uF; the number of turns of the drive
winding Ng is 1 T; the capacitor Cg has a capacitance of
0.033 uF; the gate resistor Rg has a resistance of 4.7 CJ;
the resistor Rl between the gate and source has a
resistance of 1 kC3. As the auxiliary switching element Q2,
a product of which breakdown current and voltage are 10 A
and 600 V, respectively, is selected.
In the present embodiment, the resonant frequency
fol of the power supply circuit in Fig. 1 is treated as
follows.
As shown in the waveform diagram of Figs. 3A and 3B
to be described later, the operation of the primary-side
switching converter in the power supply circuit in Fig. 1
can be regarded as complex operation between the
switching operation of the voltage resonant converter
(the main switching element Ql) and that of the active
clamp circuit 10. In addition, this complex operation can
be regarded as equivalent to the state in which one
primary-side parallel resonant circuit (hereinafter, a
primary-side parallel resonant circuit a) and another
parallel resonant circuit {hereinafter, a primary-side
parallel resonant circuit b) operate in a complex manner
on the primary side. The circuit a is formed of the
primary-side parallel resonant capacitor Cr and the
leakage inductance LI of the primary winding Nl. The
circuit b is formed of the clamp capacitor CCL and the
leakage inductance LI of the primary winding Nl. In this
case/ the primary-side parallel resonant circuit a can be
treated as a circuit that implements resonant operation
during the period when the main switching element Ql is
in the on-state. The primary-side parallel resonant
circuit b can be treated as a circuit that implements
resonant operation during the period when the auxiliary
switching element Q2 is in the on-state.
A 1
The resonant frequency fola of the primary-side
parallel resonant circuit a is 181.5 kHz since the
capacitance of the primary-side parallel resonant
capacitor Cr is 2200 pF and the leakage inductance LI of
the primary winding Nl is 350 pH. The resonant frequency
folb of the primary-side parallel resonant circuit b is
36 kHz since the capacitance of the clamp capacitor CCL
is 0.056 uF and the leakage inductance LI of the primary
winding Nl is 350 uH. In this manner, the primary side of
the circuit in Fig. 1 involves two different resonant
frequencies of two primary-side parallel resonant
circuits. Therefore, in the present embodiment, the
parallel resonant frequency fol when these primary-side
parallel resonant circuits are regarded as one integrated
circuit, is expressed by the equation fol (fola + folb)
hat is, the average value of the resonant
frequencies fola and folb of the primary-side parallel
resonant circuits a and b, respectively, is treated as
the parallel resonant frequency fol of the primary-side
parallel resonant circuit. In the present embodiment, fol
is 108.8 kHz (108.75 kHz) since fol » (181.5 kHz +
That is, the resonant frequency fol in the present
embodiment is the resonant frequency of the primary-side
parallel resonant circuit, obtained due to the operation
of the primary-side switching converter, including the
operation of the active clarap circuit 10. Therefore, the
following concept regarding resonant frequency design
available. Specifically, the resonant frequency fola of
the primary-side parallel resonant circuit a, which
includes the capacitance of the primary-side parallel
resonant capacitor Cr as its component, is designed so
that the resonant frequency fol, which is due to the
existence of the active clamp circuit 10 and is expressed
by the equation fol = (fola + folb) / 2, has a requisite
value.
The resonant frequency fo2 of the secondary-side
parallel resonant circuit is 96.5 kHz since the
capacitance of the secondary-side parallel resonant
capacitor C2 is 8200 pF and the leakage inductance L2 of
the secondary winding N2 is 332 uH.
The resonant frequency fo3 of the secondary-side
series resonant circuit is 58.9 kHz since the capacitance
of the secondary-side series resonant capacitor C3 is
0.022 pF and the leakage inductance L2 of the secondary
winding N2 is 332 uH.
In the present embodiment, the relative
relationship among the resonant frequencies fol, fo2 and
fo3 is as follows. Specifically, the resonant frequency
fol can be regarded as almost equal to the resonant
frequency fo2. In addition, the resonant frequency fo3 is
smaller than these resonant frequencies fol and fo2. That
is, it can be deemed that these resonant frequencies are
designed to have the relationship in which fol nearly
equals fo2 and is larger than fo3.
The waveform diagrams in Figs. 3A and 3B show the
operations of major parts in the power supply circuit of
Fig. I having the above-described configuration, with
reflecting the switching cycle of the switching element
Ql. Fig. 3A shows the switching voltage VI, a switching
current IQ1, a clamp current IQ2, a current Icr, a
primary winding current II, and a secondary winding
current 12 when the load power is the maximum load power
Pomax of 300 W. Fig. 3B shows the waveforms of these
currents and voltage when the load power is the minimum
load power Pomin of 0 W.
The switching voltage VI is the voltage between the
drain and source of the switching element Ql, and is also
the voltage across the parallel resonant capacitor Cr.
The switching current IQl is the current flowing through
the switching element Ql (and the body diode DD). The
switching voltage VI and the switching currant IQl
indicate the ON/OFF timings of the switching element Ql.
One switching cycle is divided into the period TON during
which the switching element Ql should be in the on-state,
and the period TOFF during which it should be in the offstate.
The switching voltage VI has a waveform in which
the voltage is at 0 level during the period TON, and is
obtained as a voltage resonant pulse during the period
TOFF. The voltage resonant pulse as the switching voltage
VI is originally obtained as a sinusoidal resonant
waveform since the operation of the primary-side
switching converter is voltage resonant operation.
However, in the present embodimentf the operation to be
described later of the active clamp circuit 10 offers a
modified waveform in which the peak of the voltage
resonant pulse is suppressed.
The switching current IQ1 is at 0 level during the
period TOFF. When the period TOFF ends and the period TON
starts, i.e. at the timing of turn on of the switching
element Ql, initially the switching current IQ1 flows
through the body diode DD and therefore has the negative
polarity. Subsequently, the switching current IQ1 flows
from the drain to the source, and thus the polarity
thereof is inverted to the positive polarity. This
waveform of the switching current IQ1 indicates that zero
volt switching (ZVS) and zero current switching (2CS) are
adequately implemented. In addition, the switching
current IQ1 can be regarded as a current that flows to
the main switching element Ql via the leakage inductance
LI arising in the primary winding Nl in the isolation
converter transformer PIT.
The primary winding current II is the current
flowing through the primary winding Nl, and results from
the synthesis between the current flowing through the
switching element Ql and the current flowing to the
primary-side parallel resonant capacitor Cr. The waveform
of the primary winding current II during the period TOFF
corresponds to the waveform of the current flowing to the
primary-side parallel resonant capacitor Cr.
An alternating voltage is induced in the secondary
winding N2. Thus, during the periods when the secondary
winding voltage has a positive level larger than a
certain level, i.e. during part of the periods of one
half cycles of the secondary winding voltage, the pair of
the rectified diodes Dol and Do4 conduct. In these
conductive periods, the rectified current flows to the
smoothing capacitor Co. In contrast, during the periods
when the secondary winding voltage has a negative level
of which absolute value is larger than a certain value,
i.e. during part of the periods of the other half cycles
of the secondary winding voltage, the pair of the
rectified diodes Do2 and Do3 conduct. In these conductive
periods, the rectified current flows to the smoothing
capacitor Co.
The secondary winding current 12 is obtained as a
result of the synthesis between the current flowing to
the secondary-side parallel resonant capacitor C2, and
the rectified currents of the both half cycles, flowing
through the pair of the rectified diodes Dol and Do4, and
the pair of the rectified diodes Do2 and Do3. Although
not shown in Fig. 3B, when the load power is the minimum
load power Pomin of 0 W, the rectified current through
the rectified diodes is at 0 level even in the periods
during which the rectified diodes Dol and Do4, or the
rectified diodes Do2 and Do3 conduct.
In the waveform diagrams of Figs. 3A and 3Bf the
operation of the active clamp circuit 10 is indicated
with being classified into five-stage operation modes
from mode 1 to mode 5 that sequentially proceed in one
switching cycle.
During the period TON when the main switching
element Ql is in the on-state, the active clamp circuit
10 implements the operation mode 1. During the period TON,
the auxiliary switching element Q2 is in the off-state.
That is, the operation mode 1 implements control so that
the auxiliary switching element Q2 is kept at the off state
In the operation mode 1 (during the period TON),
the switching current IQ1 has the above-described
waveform. Specifically, immediately after turn on of the
switching element Ql, the switching current IQ1 flows
through the clamp diode DD with the negative polarity.
Thereafter, the polarity is inverted, and thus the
switching current IQ1 flows in the direction from the
drain to the source of the main switching element Ql with
the positive polarity.
During the period when the switching current IQ1
flows with the negative polarity, the clamp diode DD
conducts since discharging of the primary-side parallel
resonant capacitor Cr has finished at the end of a period
td2, which is just anterior to the period of •the negative
switching current IQ1. Thus, the switching output current
IQl flows via the clamp diode DD and the primary winding
Nl in that order, which regenerates power for the power
supply. That is, the period of the negative switching
current corresponds to a mode of regenerating power
for the power supply. After the completion of the power
regeneration operation/ a current is supplied from the
smoothing capacitor via the primary winding Nl, so
that the switching current IQ1 flows between the source
and drain of the main switching element Ql.
When the period TON corresponding to the operation
mode 1 ends and the period TOFF starts, a period tdl
corresponding to operation mode 2 is initialized.
During the period tdl/ the main switching element
Ql is turned off, and therefore the current through the
primary winding Nl flows as a current Icr shown in Figs.
3A and 3B to charge the primary-side parallel resonant
capacitor Cr. At this time/ the charging current that
flows to the primary-side parallel resonant capacitor Cr
has a pulse-like waveform of the positive polarity. This
pulse waveform corresponds to the operation of a partial
resonant mode. In addition, at this time, the main
switching element Ql is turned off by ZVS since the
primary-side parallel resonant capacitor Cr is connected
in parallel to the main switching element Ql.
Subsequently to the period tdl, a period starts in
which control is implemented so that the auxiliary
switching element Q2 is turned on while the main
switching element Ql is kept at the off-state. This
period corresponds to a period TON2 shown in Figs. 3A and
3B. In this manner, ON/OFF of the auxiliary switching
element Q2 is controlled so that it is in the on-state
within the period during which the main switching element
Ql is in the off-state.
This period TON2 is the period during which the
active clamp circuit 10 operates. During the period TON2,
initially operation mode 3 is carried out, which is then
followed by operation mode 4.
Due to the foregoing operation mode 2, the primaryside
parallel resonant capacitor Cr is charged by the
current Icr flowing via the primary winding Nl to the
primary-side parallel resonant capacitor Cr. Thus, in the
operation mode 3 subsequent to the operation mode 2, the
voltage level of the primary winding Nl is equal to or
higher than the voltage level across the clamp capacitor
CCL at the initial timing (at the start of the period
TON2) . This voltage level is high enough that the clamp
diode DD2 connected in parallel to the auxiliary
switching element Q2 conducts. Thus/ a current flows
through the clamp diode DD2 and the clamp capacitor CCL
in that order. Accordingly, the clamp current IQ2 has a
sawtooth waveform in which the voltage level rises with
time from a negative level toward 0 after the start of
the period TON2 shown in Figs. 3A and 3B.
As described above, the capacitance of the clamp
capacitor CCL is 0.056 uF, while the capacitance of the
primary-side parallel resonant capacitor Cr is 2200 pF,
for example. Therefore, the capacitance of the primaryside
parallel resonant capacitor Cr is considerably
smaller than that of the clamp capacitor CCL. Due to such
selection of the capacitances of the clamp capacitor CCL
and the primary-side parallel resonant capacitor Cr, most
of the current flows to the clamp capacitor CCL as the
clamp current IQ2 while almost no current flows to the
primary-side parallel resonant capacitor Cr, in the
operation mode 3. As a result, the amount of charging
current to the primary-side parallel resonant capacitor
Cr during the period TON2 is reduced. Therefore, the
slope of the voltage resonant pulse as the switching
voltage VI is made gentler, which suppresses the peak
level Vlp of the switching voltage VI. That is, clamp
operation for the voltage resonant pulse is achieved.
It should be noted that, if the configuration in
Fig. 1 has no active clamp circuit 10, the voltage
resonant pulse has for example a steep sinusoidal
waveform since charging and discharging currents flow
during the whole period TOFF. The peak level of the
voltage resonant pulse is higher than that in a
configuration including the active clamp circuit 10.
After the end of the operation mode 3 in the period
TON2, the operation sequence moves to the operation mode
4.
The start timing of the operation mode 4 is
equivalent to the timing at which the flow direction of
the clamp current IQ2 shown in Figs. 3A and 3B is
inverted from the negative direction to the positive
direction. At the timing of the polarity inversion of the
clamp current IQ2 from the negative polarity to the
positive polarity, the auxiliary switching element Q2 is
turned on by ZVS and ZCS. In the state in which the
auxiliary switching element Q2 thus conducts, the
resonant operation obtained at this time by the primaryside
parallel resonant circuit allows the clamp current
IQ2 to flow via the primary winding Nl and the clamp
capacitor CCL in that order, and thus flow from the drain
to the source of the auxiliary switching element Q2.
Therefore, the clamp current IQ2 has a waveform in which
the positive voltage level increases with time as shown
in Figs. 3A and 3B.
The voltage applied to the gate of the auxiliary
switching element Q2 is the voltage induced in the drive
winding Ng, although not shown in the drawings. This
voltage is a rectangular waveform pulse voltage.
The periods tdl and td2 correspond to threshold
periods during which both the main switching element Ql
and the auxiliary switching element Q2 are in the offstate.
These threshold periods are held by the flow of
The gate-flowing current.
The operation mode4 is completed at the timing of
start of rise up of the voltage across the auxiliary
switching element Q2. Before the rise up, this voltage is
at 0 level in the period TOFF since the auxiliary
switching element Q2 conducts. Subsequent to the
operation mode 4, the operation sequence moves to
operation mode 5 during the period td2.
In the operation mode 5, a discharging current
flows from the primary-side parallel resonant capacitor
Cr to the primary winding Nl. That is, partial resonant
operation is achieved. In Figs. 3A and 3B, this
discharging as the partial resonant operation is
indicated by the current Icr flowing to the primary-side
parallel resonant capacitor Cr, as a pulse waveform
current of the negative polarity flowing only during the
period td2.
During the period td2, the voltage resonant pulse
of the switching voltage VI applied to the main switching
element Ql has a steep slope since the capacitance of the
primary-side parallel resonant capacitor Cr is smaller
than that of the clamp capacitor CCL also as described
above. Therefore, in the period td2, the voltage level
rapidly falls down toward 0 as shown in the waveform
diagram.
At the timing of start of the operation mode 5
after the end of the operation mode 4, the turn off of
the auxiliary switching element Q2 is initialized. This
turn off operation is carried out based on ZVS since the
voltage resonant pulse (the switching voltage VI) falls
down with a certain slope.
The active clamp circuit 10 implements the abovedescribed
operations mode 1 to mode 5 in each one
switching cycle.
Based on the above-described operations of the
respective parts, a comparison will be made between the
waveforms when the load power is the maximum load power
Pomax of 300 W shown in Fig. 3A, and those when it is the
minimum load power Pomin of 0 W shown in Fig. 3B. First,
the comparison shows that, in the operation of the
primary-side switching converter, as the load becomes
lighter, the period length of one switching cycle (TOFF +
TON) becomes shorter, i.e., the switching frequency fs
becomes higher. This period length variation indicates
that the above-described switching frequency control
operation for varying the switching frequency fs
depending on load variation is achieved as constantvoltage
control operation.
In the present embodiment, the lighter the load is,
the higher the switching frequency becomes as described
above. In addition, as the load becomes lighter, the
length ratio of the period TOFF to the period TON in one
switching cycle becomes larger, i.e., the duty ratio of
the period TON to the period TOFF becomes smaller. This
duty ratio decrease is due to the suppression of the peak
level Vlp of the voltage resonant pulse (VI) by the
active clamp circuit 10. When the peak level Vlp of the
voltage resonant pulse (VI) is suppressed, the conduction
angle of the voltage resonant pulse is correspondingly
increased. As is also apparent from Figs. 3A and 3B, when
the load becomes lighter, the peak level Vlp of the
voltage resonant pulse (VI) is suppressed. In association
with this suppression, the conduction angle of the
voltage resonant pulse (VI) is increased. This increase
of the conduction angle is indicated as a change of the
duty ratio between the periods TON and TOFF in one
switching cycle.
As described above, the capacitance of the clamp
capacitor CCL is considerably larger than that of the
primary-side parallel resonant capacitor Cr. A larger
difference between the capacitances of the primary-side
parallel resonant capacitor Cr and the clamp capacitor
CCL results in a larger degree of suppression of the
voltage resonant pulse peak. This larger peak suppression
however offers a larger increase of the conduction angle
of the voltage resonant pulse (VI). When the conduction
angle of the voltage resonant pulse (VI) in one switching
cycle is increased, that is, when the period (TOFF)
during which the main switching element Ql is in the offstate
is extended, the ON period (TON) of the main
switching element Ql is correspondingly shortened. If the
extent of shortening of the ON period (TON) exceeds a
certain level, there arise problems, in some cases, of
switching loss in the main switching element Ql and
lowering of amount of power transmitted from the primary
side to the secondary side. In the present embodiment,
the capacitances of the clamp capacitor CCL and the
primary-side parallel resonant capacitor Cr are selected
(CCL - 0.056 uF and Cr = 2200 pF), based on results of
experiments and so on that were carried out in
consideration of the balance among the requisite
suppression level of the voltage resonant pulse peak,
switching loss, power transmission amount, and so forth.
By this capacitance selection, the relationship in which
CCL nearly equals Cr x 25 is achieved.
Fig. 4 shows, as a function of load (from Pomin of
0 W to Pomax of 300 W), the variation characteristics of
the AC to DC power conversion efficiency (nAC -» DC), the
switching frequency fs, and the peak level Vlp of the
voltage resonant pulse (the switching voltage VI)
regarding the power supply circuit of Fig. 1. These
Characteristic curves are based on the results of
experiments on the power supply circuit of Fig. 1, and
correspond to the AC input voltages VAC of 100 V and 230
V, respectively.
According to Fig. 4, the switching frequency fs
increases as the load becomes lighter also as described
with Fig. 3. Furthermore, the switching frequency
increases as the AC input voltage VAC becomes higher.
This frequency change tendency indicates that, in the
constant-voltage control operation, the switching
frequency fs is made higher when the secondary-side DC
output voltage Eo increases in response to a load
decrease and an AC input voltage increase.
The specific values about the switching frequency
fs are as follows. When the AC input voltage VAC is 100 V,
the range of the switching frequency fs for addressing
the load variation range from the maximum load power
Pomax of 300 W to the minimum load power Pomin of 0 W is
from 80.0 kHz to 135.0 kHz. Therefore, Afs is 55.0 kHz.
The variation ranges of lengths of the periods TON and
TOFF corresponding to this switching frequency range are
from 8.5 us to 2.9 us, and from 4.0 us to 4.5 us,
respectively.
When the AC input voltage VAC is 230 V, the range
of the switching frequency fs for addressing the saine
load variation range from 300 W to 0 W is from 136.0 kHz
to 181.8 kHz. Therefore, Afs is 45.8 kHz. The variation
ranges of lengths of the periods TON and TOFF
corresponding to this switching frequency range are from
2.9 us to 0.5 us, and from 4.5 us to 5.0 us, respectively.
As for the AC to DC power conversion efficiency
(nAC - DC), a lighter load offers a higher efficiency in
the load power range from the maximum load power Pomax of
300 W to load power Po of about 100 W. In contrast, in
the load power range lower than about 100 W, a lighter
load offers a lower efficiency.
The measurement results on the AC to DC power
conversion efficiencies {nAC — DC) when the load power
58
was the maximum load power Pomax of 300 W were obtained:
the efficiency nAC - DC was 92.3% when the AC input
voltage VAC was 100 V, and the efficiency nAC — DC was
91.2% when the AC input voltage VAC was 230 V.
The peak level Vlp of the voltage resonant pulse
(VI) increases as the load becomes heavier. This peak
level rise corresponds to the increase of amount of
current charged in the primary-side parallel resonant
capacitor Cr in association with the increase of amount
of current flowing through the switching converter due to
the load power increase. The measurement results were
obtained about the peak levels Vlp when the load power
was the maximum load power Pomax of 300 W, which offered
the maximum peak levels. Specifically, the peak level Vlp
was about 400 Vp when the AC input voltage VAC was 100 V,
and the peak level Vlp was about 550 Vp when the AC input
voltage VAC was 230 V.
Since the peak level Vlp of the voltage resonant
pulse (VI) takes these values, a product of which
breakdown voltage is 600 V (e.g. TO-220 package) can be
used as the main switching element Ql as described above.
In addition, also as the auxiliary switching element Q2,
a product of which breakdown voltage is 600 V can be used
similarly. In the power supply circuit in Fig. 9 for
example, although the maximum load power Potnax thereof is
200 W, which is lower than that of the present embodiment,
a product of a breakdown voltage of 900 V is required as
the switching element Ql. If the active clamp circuit 10
is removed from the power supply circuit of Fig. 1, of
which maximum load power Pomax is 300 W, a product of a
breakdown voltage of 1200 V (e.g. TO-3P package) needs to
be used as the switching element Ql. That is, in the
present embodiment, lower-breakdown-voltage parts can be
selected as the main switching element Ql (and the
auxiliary switching element Q2), the primary-side
parallel resonant capacitor Cr connected in parallel to
the main switching element Ql, and so on.
Since low-breakdown-voltage parts can be thus
selected as the respective parts, the characteristics of
these parts elements can be enhanced. For example, the
switching characteristic of the main switching element Ql
is further improved, which reduces power loss and
improves the circuit reliability. In addition, lowerbreakdown-
voltage parts have small sizes, which can
promote size and weight reductions of circuit boards.
Moreover, costs of parts can be reduced.
Of the characteristics of the power supply circuit
in Fig. 1, the characteristic of the switching frequency
fs will be compared with that of the power supply circuit
in Fig. 9.
In the power supply circuit of Fig. 9, when the AC
input voltage VAC was 100 V, the requisite range of the
switching frequency fs for th* load power variation from
the maximum load power Pomax of 300 W to the minimum load
power Pomin of 0 W was from 117.6 kHz to 208.3 kHz.
Therefore, Afs was 96.7 kHz.
In contrast, in the power supply circuit of Fig. 1,
when the AC input voltage VAC is 100 V, the requisite
range of the switching frequency fs for the load power
variation from the maximum load power Pomax of 300 W to
the minimum load power Pomin of 0 W is from 80.0 kHz to
135.0 kHz, and therefore Afs is 55.0 kHz. This requisite
control range is significantly smaller than that of the
power supply circuit of Fig. 9. In addition, in the power
supply circuit of Fig. 1, when the AC input voltage VAC
is 230 V, the requisite range of the switching frequency
fs for the load power variation front the maximum load
power Pomax of 300 W to the minimum load power Pomin of 0
W is from 136.0 kHz to 181.8 kHz, and therefore Afs is
45.8 kHz. That is, also when the AC input voltage VAC is
230 V, a requisite control range significantly smaller
than that of the power supply circuit of Fig. 9 is
achieved.
These characteristics of the switching frequency fs
of the power supply circuit in Fig. 1 indicate that a socalled
wide-range compatible circuit is allowed that can
stably operate compatibly with commercial AC voltage
inputs in the range of the AC 100 V-system to the AC 200
V-system (e.g. the voltage VAC in the range of 85 V to
264 V), under the condition of load power variation from
the maximum load power Poraax of 300 W to the minimum load
power Pomin of 0 W. This respect will be described below.
The power supply circuit in Fig. 1 has, as its
basic configuration, a voltage resonant converter that is
provided with both a secondary-side parallel resonant
circuit and a secondary-side series resonant circuit as a
secondary-side resonant circuit. In this configuration,
the combination of the primary-side parallel resonant
circuit and secondary-side parallel resonant circuit
dominantly contributes to achievement of a wide-range
compatible circuit.
When attention is focused on the combination of the
primary-side parallel resonant circuit and secondary-side
parallel resonant circuit as the configuration of the
power supply circuit in Fig. l, it is deemed that the
power supply circuit in Fig, 1 has a parallel resonant
circuit on each of the primary and secondary sides with
the intermediary of electromagnetic coupling by the
isolation converter transformer PIT between the parallel
resonant circuits. If this configuration is interpreted
from a viewpoint of the relationship between the primaryside
parallel resonant circuit and secondary-side
resonant circuit, the power supply circuit in Fig. I can
be regarded as equal to an electromagnetically coupled
resonant circuit that is fed with a frequency signal
dependent upon the switching frequency fs.
The constant-voltage control characteristic for the
secondary-side DC output voltage Eo of the power supply
circuit in Fig. 1, which can be thus regarded as
including an electromagnetically coupled resonant circuit,
differs depending on the coupling degree (the coupling
coefficient k) of the isolation converter transformer PIT.
This respect will be described with reference to Fig. 5.
Fig. 5 shows the output characteristic of the
electromagnetically coupled resonant circuit with respect
to the input (switching frequency signal) thereto.
Specifically, the characteristic of controlling the
secondary-side DC output voltage Eo is indicated based on
the relationship between the voltage Eo and the switching
frequency In Fig. 5, switching frequencies are
plotted on the abscissa while the levels of the
secondary-side DC output voltage Eo are plotted on the
ordinate.
As is described with Fig. 1, in the present
embodiment, the resonant frequency fol of the primaryside
parallel resonant circuit is set to about 1.5 times
the resonant frequency fo2 of secondary-side parallel
resonant circuit. Therefore, the resonant frequency fol
is higher than the resonant frequency fo2. Referring to
Fig. 5, the resonant frequencies fol and fo2 are shown on
the abscissa, on which the switching frequencies fs are
plotted. Also in Fig. 5, such indication is made that the
resonant frequency fol is higher than the resonant
frequency fo2 in association with the above-described
relationship therebetween.
When the isolation converter transformer PIT
employs tight coupling in which the coupling coefficient
k is 1, the leakage inductance LI of the primary winding
Nl and the leakage inductance L2 of the secondary winding
N2 are both 0.
When the primary aide and secondary side of the
isolation converter transformer PIT are thus tightly
coupled, the constant-voltage control characteristic
draws a so-called bimodal curve as shown by
Characteristic curve 1 in Fig. 5. In this curve, the
secondary-side DC output voltage Eo comes to peaks when
the switching frequency is at frequency values of fl and
f2 that are different from the resonant frequencies fol
and fo2 of the primary-side and secondary-side parallel
resonant circuits, respectively.
The frequency fl is expressed by Equation 1.
fl = fo / Vl +lc Equation 1
The frequency f2 is expressed by Equation 2.
f2 = f o / >/l - k Equation 2
In Equations 1 and 2, To, which is one of the terms,
denotes the center resonant frequency that exists at the
center between the resonant frequency fol of the primaryside
parallel resonant circuit and the resonant frequency
fo2 of the secondary-side parallel resonant circuit. The
center resonant frequency is determined depending on the
impedance of the primary side, the impedance of the
secondary side, and the impedance common to the primary
and secondary sides (mutual coupling inductance M).
The mutual coupling inductance M is expressed by
Equation 3.
M - kVLl*L2 Equation 3
If the coupling coefficient k is gradually
decreased from 1, i.e., if the coupling state is
gradually shifted from tight coupling toward loose
coupling, such a change arises in Characteristic curve 1
in Fig. 5 that the bimodal tendency gradually fades and a
curve near the center resonant frequency fo becomes flat.
When the coupling coefficient k is decreased to a certain
value, the coupling state reaches a so-called critical
coupling state. In the critical coupling state, as shown
by Characteristic curve 2, the bimodal characteristic
tendency has disappeared and the shape of a curve around
the center resonant frequency fo is flat.
If the coupling coefficient k is further decreased
from the critical coupling state, and thus the degree of
loose coupling is further increased, a unimodal
characteristic is obtained as shown by Characteristic
curve 3 of Fig. 5, in which one peak exists only at the
center frequency fo. A comparison of Characteristic curve
3 with Characteristic curves 1 and 2 makes it apparent
that Characteristic curve 3 has a steeper slope than that
of Characteristic curves 1 and 2 as its quadratic curve
shape, although the peak level itself of Characteristic
curve 3 is lower than that of Characteristic curves 1 and
2.
The isolation converter transformer PIT of the
present embodiment employs a loose coupling state in
66
which the coupling coefficient k is smaller than about
0.7. Such a coupling coefficient Jc provides operation
that is based on a unimodal characteristic shown by
Characteristic curve 3.
When the unimodal characteristic shown in Fig. 5 is
compared with the constant-voltage control characteristic
in Fig. 12 of a conventional power supply circuit (Fig.
9), it is apparent that the characteristic in Fig. 12
shows a considerably gentler slope as a quadratic curve
than that of Fig. 5.
Since the characteristic in Fig. 12 shows a gentle
curve, even when a power supply circuit has a singlerange
compatible configuration for an AC input voltage
VAC of 100 V for example, the requisite control range of
the switching frequency fs for constant-voltage control
of the secondary-side DC output voltage Eo is from 117.6
kHz to 208.3 kHz, and therefore Afs is 96.7 kHz.
Accordingly, as described above, it is very difficult to
provide a power supply circuit with a wide-range
compatible characteristic allowed simply by implementing
constant-voltage control by switching frequency control.
In contrast, the constant-voltage control
characteristic of the present embodiment is a unimodal
characteristic shown by Characteristic curve 3 in Fig. 5,
and therefore the constant-voltage control operation
thereof is expressed by the characteristic curves of Fig.
6,
Fig. 6 illustrates four Characteristic curves A to
D obtained from the power supply circuit in Fig. 1
according to the present embodiment. Characteristics
curves A and B correspond to the maximum load power Pomax
and the minimum load power Pomin, respectively, when the
AC input voltage VAC is 100V (AC 100 V-system)..
Characteristics curves C and D correspond to the maximum
load power Pomax and the minimum load power Pomin,
respectively, when the AC input voltage VAC is 230V {AC
200 V-system).
As is apparent from Fig. 6, when the AC input
voltage VAC is 100 V, which corresponds to an AC 100 Vsystem
input, the variable control range (requisite
control range) of the switching frequency required for
keeping the secondary-side DC output voltage Eo constant
at a requisite rated level tg, is expressed by Afsl.
Specifically, the requisite control range is equivalent
to the frequency range from the switching frequency fs
providing the level tg on Characteristic curve A to the
switching frequency fs providing the level tg on
Characteristic curve B.
In addition, when the AC input voltage VAC is 230 V,
which corresponds to an AC 200 V-system input, the
variable control range (requisite control range) of the
switching frequency required for keeping the secondaryside
DC output voltage Eo constant at the requisite rated
level tg, is expressed by Afs2. Specifically/ the
requisite control range is equivalent to the frequency
range from the switching frequency fs providing the level
tg on Characteristic curve C to the switching frequency
fs providing the level tg on Characteristic curve D.
As described above, the unimodal characteristic as
the characteristic of controlling the secondary-side DC
output voltage Eo in the present embodiment draws a
considerably steep quadratic curve compared with the
control characteristic shown in Pig. 12.
Therefore, the requisite control ranges fcfsl and
Afs2 obtained when the AC input voltage VAC is 100 V and
230 V, respectively, are considerably smaller than the
range Afs in Fig. 12.
In addition, the frequency variable range (AfsA)
between the minimum switching frequency in the range Afsl
(the switching frequency fs providing the level tg on
Characteristic curve A) and the maximum switching
frequency in the range Afs2 (the switching frequency fs
providing the level tg on Characteristic curve D), is
also considerably small.
Actually measured values of the frequency ranges
Afsl, Afs2 and AfsA about the power supply circuit in Fig.
1 were as follows: Afsl - 55.0 kHz (= 135.0 kHz - 80.0
kHz), Afs2 - 45.8 kHz (- 181.8 kHz - 136.0 kHz), and AfsA
- 101.8 kHz (- 181.8 kHz - 80.0 kHz).
The frequency variable range AfsA sufficiently
falls within the switching frequency variable range of
present switching drive ICs (the oscillation and drive
circuit 2) , That is, the power supply circuit in Fig. 1
can vary and control its switching frequency within the
frequency variable range AfsA actually.
Thus, the power supply circuit of the present
embodiment in Pig. 1 can adequately stabilize the
secondary-side DC output voltage Eo, which is the main DC
voltage, both for a commercial AC voltage input of the AC
100 V-system and for one of the AC 200 V-system. That is,
a wide-range compatible configuration is allowed simply
by implementing switching frequency control.
Note that coupled resonant circuits employing
electromagnetic coupling have been already known as a
measure for, in communication techniques, widening the
amplification bandwidth of an amplification circuit
formed of a transistor. The measure is typified by
intermediate frequency transformer amplifiers for example.
In this technique field, however, a unimodal
characteristic due to loose coupling is not used but a
bimodal characteristic due to tight coupling or a flat
characteristic due to critical coupling is used. In the
present embodiment, regarding a technique of such a
coupled resonant circuit employing electromagnetic
coupling, a unimodal characteristic due to loose coupling,
which has been not used in communication technique fields,
is actively used in a field of a resonant switching
converter. Thus, the switching frequency variable range
(requisite control range) required for stabilizing the
secondary-side DC output voltage Eo is narrowed as
described above, which can form a wide-range compatible
configuration allowed simply by constant-voltage control
by switching frequency control.
As a configuration for achieving a wide-range
compatible circuit as a switching power supply circuit
including a resonant converter, besides the configuration
of the present embodiment, a configuration has been known
in which the configuration of a primary-side switching
converter is switched between a half-bride configuration
and a full-bridge configuration depending on which of an
AC 100 V-system input and an AC 200 V-system input a
commercial AC voltage input is. In addition, another
configuration is also known in which the operation of a
rectifier circuit for rectifying a commercial AC voltage
is switched between full-wave rectification and voltagedoubling
rectification depending on which of an AC 100 Vsystem
input and an AC 200 V-system input a commercial AC
voltage input is.
However, the switch over of a circuit configuration
in response to a change between the AC 100 V-system and
AC 200 V-system involves the following problem.
In order to switch the circuit configuration
according to a commercial AC voltage level, for example,
the threshold value (e.g. 150 V) of an input voltage is
set in advance, and the circuit configuration is switched
to a configuration for the AC 200 V-system when an input
voltage is higher than the threshold value, and to one
for the AC 100 V-system when it is lower than the
threshold value. However, such simple switch possibly
causes a problem. For example, when an AC 200 V-system
input is used, in response even to temporary lowering of
an AC input voltage level due to an instantaneous power
failure or the like, switch to a configuration for the AC
100 V-system is possibly carried out. Specifically, in a
configuration for switching rectifying operation for
example, there is a possibility that, even when an AC 200
V-system input is used, a determination is made that an
AC 100 V-system input is used and thus switch to a
voltage-doubler rectifier circuit is carried out, which
breaks switching elements and so on due to overvoltage.
Therefore, in practice, in order to prevent the
occurrence of the above-described malfunction, a
configuration is employed that detects not only a DC
input voltage to a main switching converter but also a DC
input voltage to a converter circuit on the standby power
supply side.
However, in order to detect a DC input voltage to a
converter circuit on the standby power supply side, for
example, a comparator 1C for comparing a reference
voltage with an input voltage needs to be provided.
Accordingly, the number of parts increases, which
facilitates an increase of circuit fabrication costs and
the size of a circuit board.
In addition, since a DC input voltage of a
converter on the standby power supply side needs to be
detected for preventing malfunction, practical use of a
power supply circuit is limited to use for an electronic
apparatus that has a standby power supply in addition to
a main power supply. That is, the kind of electronic
apparatuses for which a power supply circuit can be used
is limited to one including a standby power supply, which
problematically narrows the application range of a power
supply circuit correspondingly.
Furthermore, in a configuration that implements
switch over between half-bridge and full-bridge
configurations, at least four switching elements need to
be provided in order to allow the full-bridge
configuration. If this switch over is unnecessary, it is
enough for the circuit to include only a half-bridge
configuration, which needs only two switching elements.
In contrast, if the circuit employs this switch over,
additional two switching elements are required.
In addition, a configuration that implements switch
over of rectifying operation needs to include two
smoothing capacitors Ci in order to ensure voltagedoubling
rectifying operation. That is, compared with a
configuration that implements only full-wave
rectification, additional one smoothing capacitor Ci is
required.
Also from a viewpoint of these additional parts
necessities, a wide-range compatible configuration in
association with the above-described circuit switch over
causes an increase of circuit fabrication costs and the
size of & power supply circuit board. In particular,
since the smoothing capacitor Ci falls in the category of
large-size parts among parts of a power supply circuit,
the configuration implementing switch over of rectifying
operation further facilitates an increase of board size.
A wide control range of switching frequency also
causes another problem that the high-speed response
characteristic of stabilizing the secondary-side DC
output voltage Eo is lowered.
In particular/ some recent electronic apparatuses
involve a load condition referred to as so-called
switching load, in which load power is instantaneously
switched between the maximum load and no load in response
to ON/OFF of each drive part for example. Therefore, a
power supply circuit needs to implement constant-voltage
control of the secondary-side DC output voltage Eo with
responding to such rapid and large variation of load
power.
However, when a power supply circuit involves a
wide control range of the switching frequency as
described above, it takes a long time period for the
circuit to vary its switching frequency to the frequency
required for ensuring constant-voltage control in
response to a load that varies between the maximum and
minimum values. That is, the responsivity of constantvoltage
control is lowered.
In contrast, if a wide-range compatible
configuration is allowed simply by implementing switching
frequency control like the present embodiment, there is
no need to employ such a configuration like the abovedescribed
ones as switch, regarding rectifier circuitry
for producing a DC input voltage (Ei), rectifying
operation or the type of the switching converter between
half-bridge connection and full-bridge connection,
according to the rated level of a commercial AC voltage.
If the configuration for switch over of circuitry
is unnecessary, the number of e.g. the smoothing
capacitors Ci can be only one, and the number of
switching elements can be at least only two, requisite
for ensuring half-bridge connection. Accordingly,
reduction of circuit component parts and circuit scale,
lowering of switching noises, and so on can be achieved.
Moreover, if the configuration for switch over of
circuitry is unnecessary, the need to provide a special
configuration for preventing malfunction associated with
the switch over is also eliminated. This respect also
suppresses an increase of component parts and costs.
Furthermore/ since on electronic apparatus does not need
to have a standby power supply to prevent malfunction,
the range of apparatuses allowing use of a power supply
circuit therefore can be widened.
The number of minimum necessary parts that should
be added to a conventional voltage resonant converter,
which includes a parallel resonant circuit only on its
primary side, in order to achieve the advantages of the
present embodiment, is only a secondary-side parallel
resonant capacitor. Therefore, a wide-range compatible
configuration can be achieved with a much lower number of
additional parts than that when a conventional
configuration employing switch over of circuitry is used.
In addition, due to the narrowing of the requisite
control range (Afs) of the switching frequency fs for
constant-voltage control even under the condition of
commercial AC voltage inputs of both the AC 100 V-system
and AC 200 V-system, the responsivity and control
sensitivity of the constant-voltage control are greatly
improved.
Some electronic apparatuses implement operation in
which the load power Po is varied so that the load
condition is switched at comparatively high-speed between
the maximum load and no load. This is referred to as
called switching load. Examples of the apparatuses that
implement such operation associated with the switching
load include printers, which are a peripheral device of
personal computers, and plasma displays.
If the apparatus that involves operation associated
with the switching load is provided with a power supply
circuit having a comparatively wide requisite control
range Afs like that shown in Fig. 9 for example, the
switching frequency fs needs to be varied by a large
variation amount in association with a sharp change of
the load power also as described above. Accordingly, it
is difficult to ensure a high responsivity of constantvoltage
control.
In contrast, in the present embodiment, the
requisite control range Afs is significantly narrowed for
each of the single ranges of the AC 100 V-system and AC
200 V-system in particular. This control range reduction
allows stabilization of the secondary-side DC output
voltage Eo with high speed response to a sharp variation
of the load power Po between the maximum load and no load.
That is, the responsivity of the constant-voltage control
against switching load is significantly enhanced.
As described above with reference to Fig. 4, the
power conversion efficiency nAC -* DC of the power supply
circuit in Fig. 1 is 92.3% when the AC input voltage VAC
is 100 V and the load power is the maximum load power
Pomax of 300 W. In contrast, the power conversion
efficiency nAC - DC of the conventional power supply
circuit in Fig. 9 is about 92% when the AC input voltage
VAC is 100 V and the load power is the maximum load power
Pomax of 200 W. That is, the efficiency of the circuit in
Fig. 1 under the above-described condition is almost
equal to or larger than that of the circuit in Fig. 9
under this condition. The allowable maximum load power of
th« present embodiment is 300 W, while that of the power
supply circuit in Fig. 9 is 200 W. Therefore, when the
power supply circuit of the present embodiment and the
circuit of Pig. 9 operate under the same load condition,
the power supply circuit of the present embodiment offers
a considerably higher power conversion efficiency than
the circuit of Fig. 9. In other words, in the power
supply circuit of the present embodiment, the increase of
the allowable maximum load power from 200 W to 300 W is
allowed since a favorable power conversion efficiency
characteristic is achieved.
In the present embodiment, a primary basic factor
in the improvement of the power conversion efficiency is
that a series resonant circuit is formed on the secondary
side.
The inventor of the present application has
confirmed in advance/ from experiments and so on, that a
combination of a secondary-side series resonant circuit
with a primary-side parallel resonant circuit, as a
voltage resonant converter, is particularly advantageous
for obtaining a high power conversion efficiency. For
example, the existence of a secondary-side series
resonant circuit allows supply of power as the secondaryside
DC output voltage Eo including an energy increase
due to the resonant operation of the secondary-side
series resonant circuit. This energy increase allows a
corresponding decrease of amount of power transmitted
from the primary side to the secondary side. Thus, power
loss on the primary side is reduced, which achieves a
high power conversion efficiency.
In addition, the voltage resonant converter
provided with a secondary-side series resonant circuit
has a. characteristic in that the power conversion
efficiency tends to increase as the load becomes lighter
from the maximum load power. The power supply circuit of
the present embodiment shown in Fig. 1 has a similar
characteristic as shown in Fig. 4 since it has a
secondary-side series resonant circuit. In contrast, a
configuration obtained by combining a secondary-side
parallel resonant circuit with a voltage resonant
converter, like the power supply circuit of Fig. 9 for
example, has a tendency that the power conversion
efficiency decreases as the load becomes lighter.
Compared with such a characteristic, a voltage resonant
converter having a secondary-side series resonant circuit
exhibits a favorable characteristic as a power conversion
efficiency characteristic against load variation.
In addition to the above-described respect, the
improvement of power conversion efficiency in the present
embodiment is attributed also to setting of each resonant
frequency.
Specifically, in the present embodiment, the power
conversion efficiency nAC — DC is invariably at least 90%
when the load power Po is within the load variation range
from 200 W to 25 W. This power conversion efficiency
characteristic under such a load condition is eventually
due to adjustment of the resonant frequencies fol, fo2
and fo3. More specifically, experiments were implemented
with the resonant frequencies fol, fo2 and fo3 being set
to various values. Thus, the above-described power
conversion efficiency characteristic is eventually
obtained by establishing the relationship in which fol
nearly equals fo2 and is larger than fo3 by setting the
resonant frequencies as follows: fol » 108.8 kHz, fo2
96.5 kHz, and fo3 =58.9 kHz.
Furthermore, the improvement of the power
conversion efficiency due to this setting of the resonant
frequencies is indicated also by the waveform of the
switching current IQ1 shown in Fig. 3A.
Specifically, as is apparent from a comparison
between the switching current IQ1 of Pig. 3A
corresponding to the embodiment and that of Fig. IDA
corresponding to a conventional circuit-, in the waveform
of the switching current IQ1 of Fig. 3A corresponding to
the present embodiment, the peak level is obtained at
timing before the turn off timing of the switching
element Ql, at which the period TON ends and the period
TOFF starts. At the turn off timing/ the switching
current IQ1 has a level lower than this peak level.
This waveform of the switching current IQI is
associated with the waveform of the secondary winding
current 12. Specifically, the secondary winding current
12 has a waveform component corresponding to the current
that flows through a secondary-side resonant circuit
including a combination between a parallel resonant
circuit and a series resonant circuit. The waveform of
the secondary winding current 12 is determined by setting
of the resonant frequencies fo2 and fo3 relative to the
resonant frequency fol.
Thus/ the waveform of the switching current IQ1
shown in Fig. 3A results from appropriate setting of the
resonant frequencies fol, fo2 and fo3 of the primary-side
parallel resonant circuit, secondary-side parallel
resonant circuit and secondary-side series resonant
circuit, respectively.
The waveform of the switching current IQ1 of Fig.
3A indicates that the level of the switching current IQ1
at the time of turn off the switching element Ql is
suppressed. If the level of the switching current 1Q1 at
the time of turn off is suppressed, the switching loss
and conductive loss at the time of turn off is
correspondingly reduced.
The peak levels of the switching current IQl of Fig.
3A and Fig. JOA are S Ap and 4 Ap, respectively. The
level of Fig. 10A is smaller. This peak level difference,
however, is due to the difference of the maximum load
power condition between Fig. 3A and Fig. 10A: 300 W and
200 W, respectively. Under the same load condition, the
present embodiment offers lower switching loss and
conductive loss since it provides such operation that the

switching current IQ1 comes to the peak before turn off
of the switching element Ql.
This decrease of switching loss and conductive loss
in the switching element is one of factors in the
achievement of a high power conversion efficiency in the
power supply circuit of the present embodiment.
As described above, the basic configuration of the
power supply circuit of the present embodiment for
obtaining a favorable power conversion efficiency is a
voltage resonant converter having a series resonant
circuit on the secondary side. However, in the
configuration that merely includes the secondary-side
series resonant circuit, malfunction arises when the load
power is intermediate load power.
Specifically, in the configuration obtained by
simply combining a secondary-side series resonant circuit
with a voltage resonant converter (primary-side parallel
resonant circuit), the switching current IQ1 has the
waveform shown in Fig. 3A when the power load is the
maximum load power for example. In the waveform, the
switching current IQ1 is at 0 level until the end of the
period TOFF, which is the turn on timing of the switching
element Ql. When the period TON starts, initially a
current of the negative polarity flows through the body
diode DD, and then the polarity is inverted and the
switching current IQ1 flows between the drain and source
of the switching element Ql. That is, zero voltage
switching (ZVS) is adequately implemented. However, in a
load range of intermediate loads, such operation arises
that the switching current IQ1 flows as a noise at timing
before the end of the period TOFF, which is the turn on
timing. This operation is abnormal operation in which ZVS
is not implemented adequately.
In such abnormal operation, a corresponding peak
current flows at the time of turn on of the switching
element Ql, which causes an increase of a switching loss.
In addition, the occurrence of such abnormal operation
anyway yields an offset of the phase-gain characteristic
of the constant-voltage control circuitry for example,
which leads to switching operation in an abnormal
oscillation state. Therefore, it is difficult to put into
practical use a power supply circuit that has a
configuration in which a secondary-side parallel resonant
circuit is combined with a conventional voltage resonant
converter.
Such abnormal operation in response to an
intermediate load is attributed to the interaction
between the primary-side parallel resonant circuit that
forms the voltage resonant converter and the secondaryside
series resonant circuit, due to the simultaneous
operation thereof.
The present embodiment employs a wide gap G of the
isolation converter transformer PIT in the structure
shown in Fig. 2. Thus, the coupling coefficient k between
the primary and secondary sides is set lower than that of
a conventional circuit. Accordingly, the primary~side
parallel resonant circuit and secondary-side series
resonant circuit form an electromagnetically coupled
resonant circuit that has a sharp unimodal characteristic.
Thus, a wide-range compatible configuration is allowed.
From a viewpoint of the relationship between the
primary-side parallel resonant circuit and secondary-side
series resonant circuit, such setting of the coupling
coefficient k as to increase the degree of loose coupling
serves to attenuate the interaction between the primaryside
parallel resonant circuit and secondary-side series
resonant circuit. This attenuation of the interaction
eliminates the occurrence of a noise current as the
operation of the switching current IQ1 at the time of
turn on the switching element Ql, which offers a waveform
based on normal ZVS operation. Furthermore/ in
association with this, abnormal oscillation operation is
also eliminated. That is, in the entire allowable load
power region, stable switching operation in which ZVS
operation is maintained is ensured.
Therefore, the present embodiment of Fig. I allows
practical use of the configuration obtained by combining
the secondary-side series resonant circuit with the
primary-side parallel resonant circuit.
As described above, in the present embodiment/
loose coupling above a certain extent is ensured in the
isolation converter transformer PIT in order to achieve
two objects: achievement of a wide-range compatible
configuration allowed simply by switching frequency
control, and securing of ZVS operation in the entire
allowable load power range.
Avoided in conventional voltage resonant converters
is the enhancement of degree of loose coupling to such an
extent as to yield a coupling coefficient k like one of
the isolation converter transformer PIT of the present
embodiment. This is because such loose coupling causes
lowering of the power conversion efficiency due to an
increase of power transmission loss from the primary side
to the secondary side. In the power supply circuit of the
present embodiment, however, significantly favorable
power conversion efficiency characteristics are achieved
in almost the entire allowable load power region also as
shown in Fig. 4. The reasons and grounds therefore are as
described above. As to achievement of a wide-range
compatible configuration, as described above, the
coupling coefficient between the primary and secondary
sides of an isolation converter transformer is designed
to be lower than that of conventional power supply
circuits, to thereby form an electromagnetically coupled
resonant circuit that is formed of primary-side and
secondary-side parallel resonant circuits and has a sharp
unimodal output characteristic. Thus, a power supply
circuit can achieve a wide-range compatible configuration.
In addition, the provision of an active clamp circuit
offers changes of duty ratio of the periods Ton and Toff
within one switching period in association with
enlargement of the conduction angle. These duty ratio
changes lead to a small frequency range requisite for
constant-voltage control against load variation, which
permits the maximum load power to be 300 W.
Therefore, the maximum load power of a power supply
circuit without an active clamp circuit is 200 W, which
is the same value as that of the power supply circuit in
Fig. 9.
Variations of a secondary-side rectifier circuit as
other embodiments of the present invention will be
described below with reference to Figs. 7 and 8.
Fig. 1 illustrates the configuration of a power
supply circuit according to a second embodiment of the
invention.
Note that Fig. 7 shows only the configuration of
the secondary side from the isolation converter
transformer PIT, Since parts other than these illustrated
parts are the same as those of Fig. 1, the illustration
thereof is omitted in Fig. 7. The same parts in Fig. 7 as
those in Fig. 1 are given the sane numerals and will not
be described in detail below. This respect also applies
to Fig. 8 similarly.
In the power supply circuit in Fig. 7, the
secondary-side parallel resonant capacitor C2 is
connected in parallel to the whole secondary winding N2.
Thus, the leakage inductance L2 of the secondary winding
N2 (N2A + N2B) and the capacitance of the secondary-side
parallel resonant capacitor C2 form a secondary-side
parallel resonant circuit. In addition, a secondary-side
series resonant circuit is formed in the secondary-side
rectifier circuit as described above.
The second embodiment includes a voltage-doubler
full-wave rectifier circuit as the secondary-side
rectifier circuit.
In the voltage-doubler full-wave rectifier circuit/
the secondary winding N2 is provided with a center tap,
and thus the secondary winding N2 is divided into
secondary winding portions N2A and N2B on either side of
the center tap. The secondary winding portions N2A and
N2B have the same certain number of turns.
One end of the secondary winding N2 in the
secondary winding portion N2A is connected in series to a
secondary-side series resonant capacitor C3A. The other
end of the secondary winding N2 in the secondary winding
portion N2B is connected in series to a secondary-side
series resonant capacitor C3B. Thus, a first secondaryside
series resonant circuit is formed of the leakage
inductance component of the secondary winding portion N2A
and the capacitance of the secondary-side series resonant
capacitor C3A, while a second secondary-side series
resonant circuit is formed of the leakage inductance
component of the secondary winding portion N2B and the
capacitance of the secondary-side series resonant
capacitor C3B.
The one end of the secondary winding N2 in the
secondary winding portion N2A is coupled via the
secondary-side series resonant capacitor C3A to the
connecting node between the anode of the rectifier diode
Dol and the cathode of the rectifier diode Do2. The other
end of the secondary winding N2 in the secondary winding
portion N2B is coupled via the secondary-side series
resonant capacitor C3B to the connecting node between the
anode of the rectifier diode Do3 and the cathode of the
rectifier diode Do4.
The cathodes of the rectifier diodes Dol and Do3
are coupled to the positive electrode of the smoothing
capacitor Co. The negative electrode of the smoothing
capacitor Co is connected to the secondary-side ground.
The connecting node between the anodes of.the
rectifier diodes Do2 and Do4, and the center tap of the
secondary winding N2 are also connected to the secondaryside
ground.
This connecting structure forms first and second
voltage-doubler half-wave rectifier circuits. The first
rectifier circuit is formed of the secondary winding
portion N2A, the secondary-side series resonant capacitor
C3A, the rectifier diodes Dol and Do2, and the smoothing
capacitor Co. The second rectifier circuit is formed of
the secondary winding portion N2B, the secondary-side
series resonant capacitor C3B, the rectifier diodes Do3
and Do4, and the smoothing capacitor Co. In this case,
the first voltage-doubler half-wave rectifier circuit
includes the first secondary-side series resonant circuit,
and the second voltage-doubler half-wave rectifier
circuit includes the second secondary-side series
resonant circuit.
In the first voltage-doubler half-wave rectifier
circuit, in the periods of half cycles of one polarity of
the alternating voltage induced in the secondary winding
N2, rectifying operation is implemented in which the
rectified current follows through the secondary winding
portion N2A, the rectifier diode Do2r the secondary-side
series resonant capacitor C3A, and the secondary winding
portion N2A in that order. Thus, the secondary-side
series resonant capacitor C3A is charged by the potential
of the alternating voltage (V2) of the secondary winding
portion N2A. la the periods of half cycles of the other
polarity, rectifying operation is implemented in which
the rectified current follows through the secondary
winding portion N2A, the secondary-side series resonant
capacitor C3A, the rectifier diode Dol, and the smoothing
capacitor Co, and the secondary winding portion N2A in
that order. Thus, the smoothing capacitor Co is charged
by the potential resulting from superposition of the
voltage across the secondary-side series resonant
capacitor C3A and the alternating voltage of the
secondary winding portion N2A. In addition, in response
to the operation of the first voltage-doubler half-wave
rectifier circuit, resonant operation of the first
secondary-side series resonant circuit arises.
As for the second voltage-doubler half-wave
rectifier circuit, in the periods of half cycles of the
other polarity of the alternating voltage induced in the
secondary winding N2, rectifying operation is implemented
in which the rectified current follows through the
secondary winding portion N2B, the rectifier diode Do4,
the secondary-side series resonant capacitor C3B, and the
secondary winding portion N2B in that order. Thus, the
secondary-side series resonant capacitor C3B is charged
by the potential of the alternating voltage (equivalent
to the voltage V2) of the secondary winding portion N2B.
In the periods of half cycles of the one polarity,
rectifying operation is implemented in which the
rectified current follows through the secondary winding
portion N2B, the secondary-side series resonant capacitor
C3B, the rectifier diode Do3, and the smoothing capacitor
Co, and the secondary winding portion N2B in that order.
Thus, the smoothing capacitor Co is charged by the
potential resulting from superposition of the voltage
across the secondary-side series resonant capacitor C3B
and the alternating voltage of the secondary winding
portion N2B. In response to the operation of the second
voltage-doubler half-wave rectifier circuit, resonant
operation of the second secondary-side series resonant
circuit arises.
Furthermore/ resonant operation of the secondaryside
parallel resonant circuit arises in response to the
operation of the first and second voltage-doubler halfwave
rectifier circuits.
According to the above-described rectifying
operation, in the periods of half cycles of one polarity
of the alternating voltage of the secondary winding N2,
the charging of the smoothing capacitor Co due to the
rectified current is carried out with the potential
resulting from superposition of the voltage induced in
the secondary winding portion N2B and the voltage across
the secondary-side series resonant capacitor C3B. In
addition, in the periods of half cycles of the other
polarity, the charging of the smoothing capacitor Co due
to the rectified current is carried out with the
potential resulting from superposition of the voltage
induced in the secondary winding portion N2A and the
voltage across the secondary-side series resonant
capacitor C3A. Thus, the secondary-side DC output voltage
Eo, which is the voltage across the smoothing capacitor
Co, has the level equal to twice the level of the voltage
(V2) induced in the secondary winding portions N2A and
N2B. That is, operation of a voltage-doubler full-wave
rectifier circuit is achieved.
Fig. 8 illustrates a configuration example of a
power supply circuit according to a third embodiment of
the invention.
In the power supply circuit of Fig. 8, one
secondary-side parallel resonant capacitor C2 and one
secondary-side series resonant capacitor C3 are coupled
to the secondary winding N2 with the same connecting
structure as that of Fig. 1. Thus, on the secondary side
from the isolation converter transformer PIT, a
secondary-side parallel resonant circuit is formed of the
leakage inductance L2 of the secondary winding N2 and the
capacitance of the secondary-side parallel resonant
capacitor C2, and a secondary-side series resonant
circuit is formed of the leakage inductance L2 of the
secondary winding N2 and the capacitance of the
secondary-side series resonant capacitor C3. In addition,
a voltage-doubler half-wave rectifier circuit is provided
as a secondary-side rectifier circuit.
95
This voltage-doubler half-wave rectifier circuit is
formed by coupling two rectifier diodes Dol and Do2 and
one smoothing capacitor Co to the secondary winding N2,
to which the secondary-side parallel and series resonant
capacitors C2 and C3 are coupled as described above. The
connecting structure of the voltage-doubler half-wave
rectifier circuit is as follows. The winding-finish end
of the secondary winding N2 is coupled via the secondaryside
series resonant capacitor C3 to the anode of the
rectifier diode Dol and the cathode of the rectifier
diode Do2. The cathode of the rectifier diode &ol is
connected to the positive electrode of the smoothing
capacitor Co. Connected to the secondary-side ground are
the winding-start end of the secondary winding N2, the
anode of the rectifier diode Do2, and the negative
electrode of the smoothing capacitor Co.
Rectifying operation of the thus formed voltagedoubler
half-wave rectifier circuit is as follows.
In the periods of half cycles corresponding to one
polarity of the secondary winding voltage V2, a forward
voltage is applied to the rectifier diode Do2, which
leads to conduction of the rectifier diode Do2. Therefore,
the rectified current is charged in the secondary-side
series resonant capacitor C3. Thus, generated across the
secondary-side series resonant capacitor C3 is a voltage
with the same level as that of the alternating voltage
induced in the secondary winding N2. In the periods of
half cycles of the other polarity of the secondary
winding voltage V2, the rectifier diode Dol is provided
with a forward voltage and thus conducts. At this time,
the smoothing capacitor Co is charged by the potential
resulting from superposition of the secondary winding
voltage V2 and the voltage across the secondary-side
series resonant capacitor C3.
ThuSf generated across the smoothing capacitor Co
is the secondary-aide DC output voltage Eo having the
level equal to twice the level of the alternating voltage
excited in the secondary winding N2. In this rectifying
operation, the charging of the smoothing capacitor Co is
implemented only in the periods of half cycles of one
polarity of the alternating voltage excited in the
secondary winding N2. That is, rectifying operation of a
voltage-doubler half-wave rectifier circuit is achieved.
Moreover/ in addition to the above-described
rectifying operation, resonant operation of the
secondary-side parallel and series resonant circuits
arises.
It should be noted that the present invention is
not limited to the above-described configurations as the
embodiments. For example, other configurations are also
available as the circuit configuration of details of the
primary-side voltage resonant converter, and the
configuration of the secondary-side rectifier circuit
including a secondary-side parallel resonant circuit.
In addition, as the main switching element (and the
auxiliary switching element), for example, an insulated
gate bipolar transistor (IGBT) or a bipolar transistor
may be used instead of a MOS-FET. Moreover, although the
above-described embodiments employ a separately-excited
switching converter, the present invention can also be
applied to a configuration employing a self-excited
switching converter.

WHAT IS CLAIMED IS:
1. A switching power supply circuit comprising: a switching unit that includes a main switching element supplied with a direct-current (DC) input voltage and implementing switching;
a switching drive unit that switch-drives the main switching element;
an isolation converter transformer that includes at least a primary winding and a secondary winding that are wound around the isolation converter transformer, the primary winding being supplied with a switching output arising from switching operation of the switching unit, an alternating voltage being induced in the secondary winding by the switching output supplied to the primary winding?
a primary-side parallel resonant circuit that is formed by at least a leakage inductance component including the primary winding of the isolation converter transformer and a capacitance of a primary-side parallel resonant capacitor, the primary-side parallel resonant circuit offering voltage resonant operation as operation of the switching unit;
a secondary-side parallel resonant circuit that arises from connecting of a secondary-side parallel
resonant capacitor in parallel to the secondary winding of the isolation converter transformer, and is formed of a leakage inductance component including the secondary winding and a capacitance of the secondary-side parallel resonant capacitor;
a secondary-side series resonant circuit that arises from connecting of a secondary-side series resonant capacitor in series to the secondary winding of the isolation converter transformer, and is formed of a leakage inductance component including the secondary winding and a capacitance of the secondary-side series resonant capacitor;
a secondary-side rectifying and smoothing unit that is supplied with the alternating voltage induced in the secondary winding of the isolation converter transformer, and implements rectifying operation to thereby produce a secondary-side DC output voltage;
a constant-voltage control unit that implements constant-voltage control of the secondary-side DC output voltage by controlling the switching drive unit according to a level of the secondary-side DC output voltage to thereby vary a switching frequency of the switching unit, wherein;
the isolation converter transformer is designed to
provide such a certain coupling coefficient between a primary side and a secondary side as to offer loose coupling therebetween so that an electromagnetically coupled resonant circuit formed by at least the primary-side parallel resonant circuit and the secondary-side parallel resonant circuit has a unimodal characteristic as an output characteristic with respect to an input of a frequency signal having the switching frequency; and
at least a resonant frequency of the primary-side parallel resonant circuit, a resonant frequency of the secondary-side parallel resonant circuit and a resonant frequency of the secondary-side series resonant circuit are designed so that a power conversion efficiency above a certain level is obtained under a certain load condition.
2. The switching power supply circuit according to claim 1, further comprising:
an active clamp circuit that includes an auxiliary switching element and sets an ON period within a period during which the main switching element is in an off-state/ the auxiliary switching element being in an on-state in the ON period, the active clamp circuit causing charging and discharging currents to flow through the auxiliary switching element during the ON period, the
charging and discharging currents originally flowing to the primary-side parallel resonant capacitor in the absence of the active clamp circuit.
3. The switching power supply circuit according to
claim 1, wherein
the secondary winding is divided by a tap.
4. The switching power supply circuit according to
claim 3, wherein
each of ends of the divided secondary winding is connected in series to the secondary-side series resonant capacitor, to thereby form the secondary-side series resonant circuit.
5. The switching power supply circuit according to
claim I, wherein
the secondary-side rectifying and smoothing unit includes a bridge rectifier circuit to thereby implement full-wave rectifying operation.
6. The switching power supply circuit according to
claim 1, wherein
said secondary-side rectifying and smoothing unit includes a voltage-doubler full-wave rectifier circuit that charges at least one of the secondary-side series resonant capacitor and a secondary-side smoothing capacitor in each half cycle of the alternating voltage
excited in the divided secondary winding, to thereby produce the secondary-side DC output voltage having a level equal to twice a level of the alternating voltage.
7. The switching power supply circuit according to
claim 1, wherein
the secondary-side rectifying and smoothing unit includes a voltage-doubler half-wave rectifier circuit that charges the secondary-side series resonant capacitor or a secondary-side smoothing capacitor in each half cycle of the alternating voltage excited in the secondary winding, to thereby produce the secondary-side DC output voltage having a level equal to twice a level of the alternating voltage.
8. A switching power supply circuit comprising:
a rectifying and smoothing unit that is supplied
with an alternating-current input voltage and implements rectifying and smoothing operation;
a switching unit that includes a main switching element supplied with a direct-current (DC) voltage output from the rectifying and smoothing unit, and implementing switching;
a switching drive unit that switch-drives the main' switching element;
an isolation converter transformer that includes at
least a primary winding and a secondary winding that are wound around the isolation converter transformer, the primary winding being supplied with a switching output arising from switching operation of the switching unit/ an alternating voltage being induced in the secondary winding by the switching output supplied to the primary winding;
a primary-side parallel resonant circuit that is formed by at least a leakage inductance component including the primary winding of the isolation converter transformer and a capacitance of a primary-side parallel resonant capacitor, the primary-side parallel resonant circuit offering voltage resonant operation as operation of the switching unit;
a secondary-side parallel resonant circuit that arises from connecting of a secondary-side parallel resonant capacitor in parallel to the secondary winding of the isolation converter transformer, and is formed of a leakage inductance component including the secondary winding and a capacitance of the secondary-side parallel resonant capacitor;
a secondary-side series resonant circuit that arises from connecting of a secondary-side series resonant capacitor in series to the secondary winding of
the isolation converter transformer, and is formed of a leakage inductance component including the secondary winding and a capacitance of the secondary-side series resonant capacitor;
a secondary-side rectifying and smoothing unit that is supplied with the alternating voltage induced in the secondary winding of the isolation converter transformer, and implements rectifying operation to thereby produce a secondary-side DC output voltage; and
a constant-voltage control unit that implements constant-voltage control of the secondary-side DC output voltage by controlling the switching drive unit according to a level of the secondary-side DC output voltage to thereby vary a switching frequency of the switching unit, wherein:
the isolation converter transformer is designed to provide such a certain coupling coefficient between a primary side and a secondary side as to offer loose coupling therebetween so that an electromagnetically coupled resonant circuit formed by at least the primary-side parallel resonant circuit and the secondary-side parallel resonant circuit has a unimodal characteristic as an output characteristic with respect to an input of a frequency signal having the switching frequency; and
at least a resonant frequency of the primary-side parallel resonant circuit, a resonant frequency of the secondary-side parallel resonant circuit and a resonant frequency of the secondary-side series resonant circuit are designed so that a power conversion efficiency above a certain level is obtained under a certain load condition.
9. The switching power supply circuit according to claim 8, further comprising:
an active clamp circuit that includes an auxiliary switching element and sets an ON period within a period during which the main switching element is in an off-state, the auxiliary switching element being in an on-state in the ON period, the active clamp circuit causing charging and discharging currents to flow through the auxiliary switching element during the ON period, the charging and discharging currents originally flowing to the primary-side parallel resonant capacitor in the absence of the active clamp circuit.
10. The switching power supply circuit according
to claim 8, wherein
the secondary winding is divided by a tap.
11. The switching power supply circuit according
to claim 10, wherein
each of ends of the divided secondary winding is connected in series to the secondary-side series resonant capacitor, to thereby form the secondary-side series resonant circuit-
12. The switching power supply circuit according
to claim 8, wherein
the secondary-side rectifying and smoothing unit includes a bridge rectifier circuit to thereby implement full-wave rectifying operation.
13. The switching power supply circuit according
to claim 8, wherein
said secondary-side rectifying and smoothing unit includes a voltage-doubler full-wave rectifier circuit that charges at least one of the secondary-side series resonant capacitor and a secondary-side smoothing capacitor in each half cycle of the alternating voltage excited in the divided secondary winding/ to thereby produce the secondary-side DC output voltage having a level equal to twice a level of the alternating voltage.
14. The switching power supply circuit according
to claim 8, wherein
the secondary-side rectifying and smoothing unit includes a voltage-doubler half-wave rectifier circuit that charges the secondary-side series resonant capacitor
or a secondary-side smoothing capacitor in each half cycle of the alternating voltage excited in the secondary winding, to thereby produce the secondary-side DC output voltage having a level equal to twice a level of the alternating voltage.