DESCRlPTION
[Title of Invention]
TRANSMISSION METHOD, TRANSMISSION DEVICE, RECEPTION
METHOD, AND RECEPTION DEVICE
5 [Technical Field]
[OOOl]
(Cross-Reference to Related Application) The disclosures of Japanese
Patent Application No. 2010-234061, filed on October 18, 2010 and No.
20 10-275 164, filed on December 9, 20 10, including the claims, specifications,
10 drawings, and abstracts thereof, are incorporated herein by reference in their
entirety.
[0002]
The present invention relates to a precoding method, a precoding device, a
transmission method, a transmission device, a reception method, and a reception
15 device that in particular perform communication using a multi-antenna.
[Background Art]
[0003]
Multiple-Input Multiple-Output (MIMO) is a conventional example of a
communication method using a multi-antenna. In multi-antenna communication, of
20 which MIMO is representative, multiple transmission signals are each modulated,
and each modulated signal is transmitted fiom a different antenna simultaneously in
order to increase the transmission speed of data.
[0004]
Fig. 28 shows an example of the structure of a transmission and reception
25 device when the number of transmit antennas is two, the number of receive antennas
is two, and the number of modulated signals for transmission (transmission streams)
is two. In the transmission device, encoded data is interleaved, the interleaved data is
modulated, and fiequency conversion and the like is performed to generate
transmission signals, and the transmission signals are transmitted fiom antennas. In
this case, the method for simultaneously transmitting different modulated signals
fi-om different transmit antennas at the same time and at the same frequency is
spatial multiplexing MIMO.
5 [0005]
In this context, it has been suggested in Patent Literature 1 to use a
transmission device provided with a different interleave pattern for each transmit
antenna. In other words, the transmission device in Fig. 28 would have two different
interleave patterns with respective interleaves (my xb). As shown in Non-Patent
10 Literature 1 and Non-Patent Literature 2, reception quality is improved in the
reception device by iterative performance of a phase detection method that uses soft
values (the MIMO detector in Fig. 28).
[0006]
Models of actual propagation environments in wireless communications
15 include non-line of sight (NLOS), of which a Rayleigh fading environment is
representative, and line of sight (LOS), of which a Rician fading environment is
representative. When the transmission device transmits a single modulated signal,
and the reception device performs maximal ratio combining on the signals received
by a plurality of antennas and then demodulates and decodes the signal resulting
20 fiom maximal ratio combining, excellent reception quality can be achieved in an
LOS environment, in particular in an environment where the Rician factor is large,
which indicates the ratio of the received power of direct waves versus the received
power of scattered waves. However, depending on the transmission system (for
example, spatial multiplexing MIMO system), a problem occurs in that the reception
25 quality deteriorates as the Rician factor increases (see Non-Patent Literature 3).
Figs. 29A and 29B show an example of simulation results of the Bit Error
Rate PER) characteristics (vertical axis: BER, horizontal axis: signal-to-noise
power ratio (SNR)) for data encoded with low-density parity-check (LDPC) code
2
and transmitted over a 2 x 2 (two transmit antennas, two receive antennas) spatial
multiplexing MIMO system in a Rayleigh fading environment and in a Rician fading
environment with Rician factors of K = 3, 10, and 16 dB. Fig. 29A shows the BER
characteristics of Max-log A Posteriori Probability (APP) without iterative detection
5 (see Non-Patent Literature 1 and Non-Patent Literature 2), and Fig. 29B shows the
BER characteristics of Max-log-APP with iterative detection (see Non-Patent
Literature 1 and Non-Patent Literature 2) (number of iterations: five). As is clear
fiom Figs. 29A and 29B, regardless of whether iterative phase detection is
performed, reception quality degrades in the spatial multiplexing MIMO system as
10 the Rician factor increases. It is thus clear that the unique problem of "degradation
of reception quality upon stabilization of the propagation environment in the spatial
multiplexing MIMO system", which does not exist in a conventional single
modulation signal transmission system, occurs in the spatial multiplexing MIMO
system.
15 [0007]
Broadcast or multicast communication is a service directed towards
line-of-sight users. The radio wave propagation environment between the
broadcasting station and the reception devices belonging to the users is often an
LOS environment. When using a spatial multiplexing MIMO system having the
20 above problem for broadcast or multicast communication, a situation may occur in
which the received electric field strength is high at the reception device, but
degradation in reception quality makes it impossible to receive the service. In other
words, in order to use a spatial multiplexing MIMO system in broadcast or multicast
communication in both an NLOS environment and an LOS environment, there is a
25 desire for development of a MIMO system that offers a certain degree of reception
quality.
[OOOS]
Non-Patent Literature 8 describes a method to select a codebook used in
precoding (i.e. a precoding matrix, also referred to as a precoding weight matrix)
based on feedback information fiom a communication partner. Non-Patent
Literature 8 does not at all disclose, however, a method for precoding in an
5 environment in which feedback information cannot be acquired fiom the
communication partner, such as in the above broadcast or multicast communication.
[0009]
On the other hand, Non-Patent Literature 4 discloses a method for switching
the precoding matrix over time. This method can be applied even when no feedback
10 information is available. Non-Patent Literature 4 discloses using a unitary matrix as
the matrix for precoding and switching the unitary matrix at random but does not at
all disclose a method applicable to degradation of reception quality in the
above-described LOS environment. Non-Patent Literature 4 simply recites hopping
between precoding matrices at random. Obviously, Non-Patent Literature 4 makes
15 no mention whatsoever of a precoding method, or a structure of a precoding matrix,
for remedying degradation of reception quality in an LOS environment.
[Citation List]
[Patent Literature]
[OO 1 01
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won-Patent Literature]
[OOl 11
won-Patent Literature 11
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[Non-Patent Literature 21
"Performance analysis and design optimization of LDPC-coded MIMO
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I DVB Document A122, Framing structure, channel coding and modulation
I
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T. Ohgane, T. Nishimura, and Y. Ogawa, "Application of space division
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! [Summary of Invention]
[Technical Problem]
[OO 121
It is an object of the present invention to provide a MIMO system that
improves reception quality in an LOS environment.
[Solution to Problem]
[00 131
5 In order to solve the above problems, an aspect of the present invention is a
precoding method for generating, fiom a plurality of baseband signals, a plurality of
precoded signals to be transmitted over the same fiequency bandwidth at the same
time, comprising the steps of: selecting a matrix F[i] fiom among N matrices while
switching between the N matrices, the N matrices defining precoding performed on
10 the plurality of baseband signals, i being an integer fiom 0 to N - 1, and N being an
integer at least two; and generating a first precoded signal zl and a second precoded
signal 22 by precoding, in accordance with the selected matrix F[i], a first baseband
signal sl generated fiom a first plurality of bits and a second baseband signal s2
generated from a second plurality of bits, a first encoded block and a second
15 encoded block being generated respectively as the first plurality of bits and the
second plurality of bits using a predetermined error correction block encoding
method, the first baseband signal sl and the second baseband signal s2 being
generated respectively fiom the first encoded block and the second encoded block to
have M symbols each, the first precoded signal zl and the second precoded signal 22
20 being generated to have M slots each by precoding a combination of the first
baseband signal sl and the second baseband signal s2, M being an integer at least
two, the first precoded signal zl and the second precoded signal 22 satisfling the
equation (zl, 221T = F[i](sl, s21T.
[00 1 41
25 Another aspect of the present invention is a precoding device for generating,
fiom a plurality of baseband signals, a plurality of precoded signals to be transmitted
over the same fiequency bandwidth at the same time, comprising: a weighting
information generation unit configured to select a matrix F[i] fiom among N
matrices while switching between the N matrices, the N matrices defining precoding
performed on the plurality of baseband signals, i being an integer fiom 0 to N - 1,
and N being an integer at least two; a weighting unit configured to generate a first
precoded signal zl and a second precoded signal 22 by precoding, in accordance
5 with the selected matrix F[i], a first baseband signal sl generated fiom a first
plurality of bits and a second baseband signal s2 generated fiom a second plurality
of bits; an error correction coding unit configured to generate a first encoded block
as the first plurality of bits and a second encoded block as the second plurality of
bits using a predetermined error correction block encoding method; and a mapper
10 configured to generate a baseband signal with M symbols fiom the first encoded
block and a baseband signal with M symbols fiom the second encoded block, M
being an integer at least two, the first precoded signal zl and the second precoded
signal 22 satisfying the equation (zl, = F[i](sl, s21T, and the weighting unit
generating precoded. signals with M slots by precoding a combination of the
15 baseband signal generated fiom the first encoded block and the baseband signal
generated fiom the second encoded block.
[00 1 51
With the above aspects of the present invention, a modulated signal is
generated by performing precoding while hopping between precoding matrices so
20 that among a plurality of precoding matrices, a precoding matrix used for at least
one data symbol and precoding matrices that are used for data symbols that are
adjacent to the data symbol in either the frequency domain or the time domain all
differ. Therefore, reception quality in an LOS environment is improved in response
to the design of the plurality of precoding matrices.
25 [Advantageous Effects of Invention]
[00 1 61
With the above structure, the present invention provides a transmission
method, a reception method, a transmission device, and a reception device that
remedy degradation of reception quality in an LOS environment, thereby providing
high-quality service to LOS users during broadcast or multicast communication.
prief Description of Drawings]
[00 171
5 Fig. 1 is an example of the structure of a transmission device and a
reception device in a spatial multiplexing MIMO system.
Fig. 2 is an example of a fiame structure.
Fig. 3 is an example of the structure of a transmission device when adopting
a method of hopping between precoding weights.
10 Fig. 4 is an example of the structure of a transmission device when adopting
a method of hopping between precoding weights.
Fig. 5 is an example of a fiame structure.
Fig. 6 is an example of a method of hopping between precoding weights.
Fig. 7 is an example of the structure of a reception device.
Fig. 8 is an example of the structure of a signal processing unit in a
reception device.
Fig. 9 is an example of the structure of a signal processing unit in a
reception device.
Fig. 10 shows a decoding processing method.
Fig. 11 is an example of reception conditions.
Figs. 12A and 12B are examples of BER characteristics.
Fig. 13 is an example of the structure of a transmission device when
adopting a method of hopping between precoding weights.
Fig. 14 is an example of the structure of a transmission device when
25 adopting a method of hopping between precoding weights.
Figs. 15A and 15B are examples of a fiame structure.
Figs. 16A and 16B are examples of a fiame structure.
Figs. 17A and 17B are examples of a fiame structure.
9
Figs. 18A and 18B are examples of a fiame structure.
Figs. 19A and 19B are examples of a £kame structure.
Fig. 20 shows positions of poor reception quality points.
Fig. 21 shows positions of poor reception quality points.
Fig. 22 is an example of a hme structure.
Fig. 23 is an example of a fiame structure.
Figs. 24A and 24B are examples of mapping methods.
Figs. 25A and 25B are examples of mapping methods.
Fig. 26 is an example of the structure of a weighting unit.
Fig. 27 is an example of a method for reordering symbols.
Fig. 28 is an example of the structure of a transmission device and a
reception device in a spatial multiplexing MIMO system.
Figs. 29A and 29B are examples of BER characteristics.
Fig. 30 is an example of a 2 x 2 MIMO spatial multiplexing MIMO system.
Figs. 3 1A and 3 1B show positions of poor reception points.
Fig. 32 shows positions of poor reception points.
Figs. 33A and 33B show positions of poor reception points.
Fig. 34 shows positions of poor reception points.
Figs. 35A and 35B show positions of poor reception points.
Fig. 36 shows an example of minimum distance characteristics of poor
reception points in an imaginary plane.
Fig. 37 shows an example of minimum distance characteristics of poor
reception points in an imaginary plane.
.Figs. 38A and 38B show positions of poor reception points.
Figs. 39A and 39B show positions of poor reception points.
Fig. 40 is an example of the structure of a transmission device in
Embodiment 7.
Fig. 41 is an example of the frame structure of a modulated signal
transmitted by the transmission device.
Figs. 42A and 42B show positions of poor reception points.
Figs. 43A and 43B show positions of poor reception points.
5 Figs. 44A and 44B show positions of poor reception points.
Figs. 45A and 45B show positions of poor reception points.
Figs. 46A and 46B show positions of poor reception points.
Figs. 47A and 47B are examples of a frame structure in the time and
frequency domains.
10 Figs. 48A and 48B are examples of a frame structure in the time and
frequency domains.
Fig. 49 shows a signal processing method.
Fig. 50 shows the structure of modulated signals when using space-time
block coding.
15 Fig. 51 is a detailed example of a fiame structure in the time and frequency
domains.
Fig. 52 is an example of the structure of a transmission device.
Fig. 53 is an example of a structure of the modulated signal generating units
#I-#M in Fig. 52.
20 Fig. 54 shows the structure of the OFDM related processors (5207-1 and
5207-2) in Fig. 52.
Figs. 55A and 55B are detailed examples of a frame structure in the time
and frequency domains.
Fig. 56 is an example of the structure of a reception device.
Fig. 57 shows the structure of the OFDM related processors (5600-X and
5600 - Y) in Fig. 56.
Figs. 58A and 58B are detailed examples of a frame structure in the time
and frequency domains.
11
Fig. 59 is an example of a broadcasting system.
Figs. 60A and 60B show positions of poor reception points.
Fig. 61 is an example of the structure of a transmission device when
adopting hierarchical transmission.
5 Fig. 62 is an example of the structure of a transmission device when
adopting hierarchical transmission.
Fig. 63 is an example of precoding of a base stream.
Fig. 64 is an example of precoding of an enhancement stream.
Figs. 65A and 65B are examples of arrangements of symbols in modulated
10 signals when adopting hierarchical transmission.
Fig. 66 is an example of the structure of a signal processing unit in a
transmission device when adopting hierarchical transmission.
Fig. 67 is an example of the structure of a transmission device when
adopting hierarchical transmission.
15 Fig. 68 is an example of the structure of a transmission device when
adopting hierarchical transmission.
Fig. 69 is an example of a structure of symbols in a baseband signal.
Figs. 70A and 70B are examples of arrangements of symbols in modulated
signals when adopting hierarchical transmission.
20 Fig. 71 is an example of the structure of a transmission device when
adopting hierarchical transmission.
Fig. 72 is an example of the structure of a transmission device when
adopting hierarchical transmission.
Fig. 73 is an example of a structure of symbols in space-time block coded
25 baseband signals.
Figs. 74A and 74B are examples of arrangements of symbols in modulated
signals when adopting hierarchical transmission.
Figs. 75A and 75B are examples of arrangements of symbols in modulated
signals when adopting hierarchical transmission.
Fig. 76 is an example of a modification of the number of symbols and of
slots necessary for one encoded block when using block coding.
5 Fig. 77 is an example of a modification of the number of symbols and of
slots necessary for two encoded blocks when using block coding.
Fig. 78 shows the overall structure of a digital broadcasting system.
Fig. 79 is a block diagram showing an example of the structure of a
reception device.
10 Fig. 80 shows the structure of multiplexed data.
Fig. 81 schematically shows how each stream is multiplexed in the
multiplexed data.
Fig. 82 shows in detail how a video stream is stored in a sequence of PES
packets.
15 Fig. 83 shows the structure of a TS packet and a source packet in
multiplexed data.
Fig. 84 shows the data structure of a PMT.
Fig. 85 shows the internal structure of multiplexed data information.
Fig. 86 shows the internal structure of stream attribute information.
20 Fig. 87 is a structural diagram of a video display / audio output device.
Fig. 88 shows the structure of a baseband signal switching unit.
[Description of Embodiments]
[00 1 81
The following describes embodiments of the present invention with
25 reference to the drawings.
(Embodiment 1)
The following describes the transmission method, transmission device,
reception method, and reception device of the present embodiment.
13
[00 1 91
Prior to describing the present embodiment, an overview is provided of a
transmission method and decoding method in a conventional spatial multiplexing
MIMO system.
5 Fig. 1 shows the structure of an Nt x N, spatial multiplexing MIMO system.
An information vector z is encoded and interleaved. As output of the interleaving, an
encoded bit vector u = (u,, . . ., uNt) is acquired. Note that ui = (uil, . . ., uM) (where M
is the number of transmission bits per symbol). Letting the transmission vector s =
(sl, . . .. , sNtlTa nd the transmission signal from transmit antenna #1 be represented as
10 si = map(u,), the normalized transmission energy is represented as ~ { l s ~=f } E s/Nt
(E, being the total energy per channel). Furthermore, letting the received vector be y
= (yl, . . ., yN,)T, the received vector is represented as in Equation 1.
[0020]
Math 1
15 Equation 1
roo2 11
In this Equation, HNNr is the channel matrix, n = (nl, ..., nNJT is the noise
20 vector, and ni is the i.i.d. complex Gaussian random noise with an average value 0
and variance 2. From the relationship between transmission symbols and reception
symbols that is induced at the reception device, the probability for the received
vector may be provided as a multi-dimensional Gaussian distribution, as in Equation
Math 2
Equation 2
5 [0023]
Here, a reception device that performs iterative decoding composed of an
outer soft-inlsoft-out decoder and a MIMO detector, as in Fig. 1, is considered. The
vector of a log-likelihood ratio (L-value) in Fig. 1 is represented as in Equations
3-5.
10 [0024]
Math 3
Equation 3
15 [0025]
Math 4
Equation 4
20 [0026]
Math 5
Equation 5
[0027]

5 The following describes iterative detection of MIMO signals in the Nt x N,
spatial multiplexing MIMO system.
The log-likelihood ratio of u, is defined as in Equation 6.
[0028]
Math 6
10 Equation 6
[0029]
' From Bayes' theorem, Equation 6 can be expressed as Equation 7.
15 [0030]
Math 7
Equation 7
Let Urn,*, = {ulu, = &I). When approximating Idaj - ma. In aj, an
5 approximation of Equation 7 can be sought as Equation 8. Note that the above
symbol "-" indicates approximation.
[0032]
Math 8
Equation 8
[0033]
P(ulu,) and In P(ulu,) in Equation 8 are represented as follows.
[0034]
15 Math 9
Equation 9
[0035]
Math 10
5 Equation 10
[0036]
Math 11
10 Equation 1 1
Incidentally, the logarithmic probability of the equation defined in Equation
2 is represented in Equation 12.
[0038]
Math 12
5 Equation 12
[0039]
Accordingly, fiom Equations 7 and 13, in MAP or A Posteriori Probability
10 (APP), the a posteriori L-value is represented as follows.
[0040]
Math 13
Equation 13
Cume.x+P,{ - l2i0l y- ~s(u)l'+XI / znp(u~}
L(u,. I Y) = In { XU,., exp - -2-0--2 I IY-HS(U,I~+C.PL,)} !I
15
[0041]
Hereinafter, this is referred to as iterative APP decoding. From Equations 8
and 12, in the log-likelihood ratio utilizing Max-Log approximation (Max-Log APP),
the a posteriori L-value is represented as follows.
20 [0042]
Math 14
Equation 14
[0043]
Math 15
5 Equation 15
[0044]
Hereinafter, this is referred to as iterative Max-log APP decoding. The
10 extrinsic information required in an iterative decoding system can be sought by
subtracting prior inputs fiom Equations 13 and 14.

Fig. 28 shows the basic structure of the system that is related to the
subsequent description. This system is a 2 x 2 spatial multiplexing MIMO system.
15 There is an outer encoder for each of streams A and B. The two outer encoders are
identical LDPC encoders. (Here, a structure using LDPC encoders as the outer
encoders is described as example, but the error correction coding used by the
outer encoder is not limited to LDPC coding. The present invention may similarly be
embodied using other error correction coding such as turbo coding, convolutional
20 coding, LDPC convolutional coding, and the like. Furthermore, each outer encoder
is described as having a transmit antenna, but the outer encoders are not limited to
this structure. A plurality of transmit antennas may be used, and the number of outer
encoders may be one. Also, a greater number of outer encoders may be used than the
number of transmit antennas.) The streams A and B respectively have interleavers
(n, nb). Here, the modulation scheme is 2 h -(wi~th h~ bits~ tra nsmitted in one
symbol).
[0045]
The reception device performs iterative detection on the above MlMO
5 signals (iterative APP (or iterative Max-log APP) decoding). Decoding of LDPC
codes is performed by, for example, sum-product decoding.
[0046]
Fig. 2 shows a fiame structure and lists the order of symbols after
interleaving. In this case, (i, j3, (ib, jb) are represented by the following Equations.
10 [0047]
-
Math 16
Equation 16
15 [0048]
Math 17
Equation 17
20 [0049]
In this case, ia, ib indicate the order of symbols after interleaving, ja, jb
indicate the bit positions (ja,j b = 1, .. . , h) in the modulation scheme, na, nb indicate
the interleavers for the streams A and B, and nai,,, indicate the order of data
in streams A and B before interleaving. Note that Fig. 2 shows the fiame structure
25 for ia = ib.

The following is a detailed description of the algorithms for sum-product
decoding used in decoding of LDPC codes and for iterative detection of MIMO
signals in the reception device.
[0050]
5 Sum-Product Decoding
Let a two-dimensional M x N matrix H = {Hm) be the check matrix for
LDPC codes that are targeted for decoding. Subsets A(m), B(n) of the set 11, N] = (1,
2, . . ., N) are defined by the following Equations.
[005 11
10 Math 18
Equation 18
LO0521
15 Math19
Equation 19
100531
20 In these Equations, A(m) represents the set of column indices of 1's in the
mth column of the check matrix H, and B(n) represents the set of row indices of 1's
in the nfi row of the check matrix H. The algorithm for sum-product decoding is as
follows.
Step Am1 (initialization): let a priori value logarithmic ratio P, = 0 for all
25 combinations (m, n) satisming H,, = 1. Assume that the loop variable (the number
of iterations) l,, = 1 and the maximum number of loops is set t o. ,l,Su,
22
Step A-2 (row processing): the extrinsic value logarithmic ratio & is updated for all
combinations (m, n) satisfying H,, = 1 in the order of m = 1, 2, ..., M, using the
following updating Equations.
[0054]
5 Math 20
Equation 20
Equation 21
[0056]
15 Math 22
Equation 22
exp(x) + 1
f (x) 3 in
exp(x) - 1
[0057]
20 In these Equations, f represents a Gallager function. Furthermore, the
method of seeking h, is described in detail later.
Step A.3 (column processing): the extrinsic value logarithmic ratio P, is updated
for all combinations (m, n) satisfiing H,, = 1 in the order of n = 1, 2, . . ., N, using
the following updating Equation.
[0058]
5 Math23
Equation 23
[0059]
10 Step A-4 (calculating a log-likelihood ratio): the log-likelihood ratio L, is sought for
n E [I, NJ by the following Equation.
[0060]
Math 24
Equation 24
[0061]
Step A-5 (count of the number of iterations): if I, < I, ,, then I, is
incremented, and processing returns to step A-2. If 1,- = I, the sum-product
20 decoding in this round is finished.
[0062]
The operations in one sum-product decoding have been described.
Subsequently, iterative MIMO signal detection is performed. In the variables m, n,
, ,hp , &, and L,, used in the above description of the operations of sum-product
decoding, the variables in stream A are m, n, aam,, pam,, La, and L,, and the
variables in stream B are mb,nb, ab m bnbp,b mbnLb,b ,a nd Lnb.

The following describes the method of seeking A, in iterative MIMO signal
5 detection in detail.
[0063]
The following Equation holds fiom Equation 1.
[0064]
Math 25
10 Equation 25
[0065]
The following Equations are defined fiom the fiame structures of Fig. 2 and
1 5 fiom Equations 16 and 17.
[0066]
Math 26
Equation 26
a n, = R i a , ja
20
[0067]
Math 27
Equation 27
[0068]
In this case, n,nb E [I, N]. Hereinafter, b, L,, Lb, and b, where the
5 number of iterations of iterative MIMO signal detection is k, are represented as h, ,,
Lk, na, hk, nb, and Ll, nb-
[0069]
Step B- 1 (initial detection; k = 0): b, and &, ,,b are sought as follows in the
case of initial detection.
10 In iterative APP decoding:
[0070]
Math 28
Equation 28
15
[0071]
In iterative Max-log APP decoding:
[0072]
Math 29
20 Equation 29
[0073]
Math 30
Equation 30
[0074]
Here, let X = a, b. Then, assume that the number of iterations of iterative
MIMO signal detection is lmim=o 0 and the maximum number of iterations is set to
10 lm,mo, m a r
[0075]
Step B-2 (iterative detection; the number of iterations k): hk, na and hk .b,
where the number of iterations is k, are represented as in Equations 31-34, fiom
Equations 1 1, 13-1 5, 16, and 17. Let (X, Y) = (a, b)(b, a).
15 In iterative APP decoding:
[0076]
Math 3 1
Equation 3 1
20
[0077]
Math 32
Equation 32
[0078]
In iterative Max-log APP decoding:
5 [0079]
Math 33
Equation 33
10 [OOSO]
Math 34
Equation 34
15 [0081]
Step B-3 (counting the number of iterations and estimating a codeword):
increment I, if lmim, < lmimo,,, and return to step B-2. Assuming that lmim=o I-,,
,,, the estimated codeword is sought as in the following Equation.
[0082]
20 Math 35
Equation 35
[0083]
Here, let X = a, b.
5 [0084]
Fig. 3 is an example of the structure of a transmission device 300 in the
present embodiment. An encoder 302A receives information (data) 301A and a
fiame structure signal 313 as inputs and, in accordance with the fiame structure
signal 313, performs error correction coding such as convolutional coding, LDPC
10 coding, turbo coding, or the like, outputting encoded data 303A. (The flame
structure signal 3 13 includes information such as the error correction method used
for error correction coding of data, the encoding ratio, the block length, and the like.
The encoder 302A uses the error correction method indicated by the fi-ame structure
signal 3 13. Furthermore, the error correction method may be switched.)
[0085]
An interleaver 304A receives the encoded data 303A and the fiame
structure signal 313 as inputs and performs interleaving, i.e. changing the order of
the data, to output interleaved data 305A. (The method of interleaving may be
switched based on the h e structure signal 3 13.)
A mapper 306A receives the interleaved data 305A and the fiame structure
signal 313 as inputs, performs modulation such as Quadrature Phase Shift Keying
(QPSK), 16 Quadrature Amplitude Modulation (16QAM), 64 Quadrature Amplitude
Modulation (64QAM), or the like, and outputs a resulting baseband signal 307A.
(The method of modulation may be switched based on the fiame structure signal
313.)
Figs. 24A and 24B are an example of a mapping method over an IQ plane,
having an in-phase component I and a quadrature component Q, to form a baseband
signal in QPSK modulation. For example, as shown in Fig. 24A, if the input data is
"OO", the output is I = 1.0, Q = 1 .O. Similarly, for input data of "01 ", the output is I =
5 -1.0, Q = 1.0, and so forth. Fig. 24B is an example of a different method of mapping
in an IQ plane for QPSK modulation than Fig. 24A. The difference between Fig.
24B and Fig. 24A is that the signal points in Fig. 24A have been rotated around the
origin to yield the signal points of Fig. 24B. Non-Patent Literature 9 and Non-Patent
Literature 10 describe such a constellation rotation method, and the Cyclic Q Delay
10 described in Non-Patent Literature 9 and Non-Patent Literature 10 may also be
adopted. As another example apart fiom Figs. 24A and 24B, Figs. 25A and 25B
show signal point layout in the IQ plane for 16QAM. The example corresponding to
Fig. 24A is shown in Fig. 25A, and the example corresponding to Fig. 24B is shown
in Fig. 25B.
15 [0086]
An encoder 302B receives information (data) 301B and the fiame structure
signal 3 13 as inputs and, in accordance with the fiame structure signal 3 13, performs
error correction coding such as convolutional coding, LDPC coding, turbo coding,
or the like, outputting encoded data 303B. (The fiame structure signal 3 13 includes
20 information such as the error correction method used, the encoding ratio, the block
length, and the like. The error correction method indicated by the fiame structure
signal 3 13 is used. Furthermore, the error correction method may be switched.)
An interleaver 304B receives the encoded data 303B and the fiame structure
signal 313 as inputs and performs interleaving, i.e. changing the order of the data, to
25 output interleaved data 305B. (The method of interleaving may be switched based
on the fiame structure signal 3 13 .)
A mapper 306B receives the interleaved data 305B and the fiame structure
signal 313 as inputs, performs modulation such as Quadrature Phase Shift Keying
30
(QPSK), 16 Quadrature Amplitude Modulation (1 6QAM), 64 Quadrature Amplitude
Modulation (64QAM), or the like, and outputs a resulting baseband signal 307B.
(The method of modulation may be switched based on the fiame structure signal
313.)
5 [0087]
A weighting information generating unit 314 receives the fiame structure
signal 3 13 as an input and outputs information 3 15 regarding a weighting method
based on the fiame structure signal 313. The weighting method is characterized by
regular hopping between weights.
10 [0088]
A weighting unit 308A receives the baseband signal 307A, the baseband
signal 307B, and the information 3 15 regarding the weighting method, and based on
the information 315 regarding the weighting method, performs weighting on the
baseband signal 307A and the baseband signal 307B and outputs a signal 309A
15 resulting fiom the weighting. Details on the weighting method are provided later.
[0089]
A wireless unit 3 10A receives the signal 309A resulting fiom the weighting
is an input and performs processing such as orthogonal modulation, band limiting,
frequency conversion, amplification, and the like, outputting a transmission signal
20 3 1 1A. A transmission signal 5 1 1A is output as a radio wave fiom an antenna 3 12A.
[0090]
A weighting unit 308B receives the baseband signal 307A, the baseband
signal 307B, and the information 315 regarding the weighting method, and based on
the information 315 regarding the weighting method, performs weighting on the
25 baseband signal 307A and the baseband signal 307B and outputs a signal 309B
resulting fiom the weighting.
[0091]
Fig. 26 shows the structure of a weighting unit. The baseband signal 307A
is multiplied by wl l(t), yielding wl l(t)sl (t), and is multiplied by w21(t), yielding
w2l(t)sl(t). Similarly, the baseband signal 307B is multiplied by w12(t) to generate
w12(t)s2(t) and is multiplied by w22(t) to generate w22(t)s2(t). Next, zl(t) =
5 wl 1 (t)sl (t) + w 12(t)s2(t) and z2(t) = w2l (t)sl (t) + w22(t)s2(t) are obtained.
[0092]
Details on the weighting method are provided later.
[0093]
A wireless unit 310B receives the signal 309B resulting fiom the weighting
10 as an input and performs processing such as orthogonal modulation, band limiting,
frequency conversion, amplification, and the like, outputting a transmission signal
3 1 1B. A transmission signal 5 1 1B is output as a radio wave fiom an antenna 3 12B.
[0094]
Fig. 4 shows an example of the structure of a transmission device 400 that
15 differs fiom Fig. 3. The differences in Fig. 4 fiom Fig. 3 are described.
[0095]
An encoder 402 receives information (data) 401 and the fiame structure
signal 3 13 as inputs and, in accordance with the h e structure signal 3 13, performs
error correction coding and outputs encoded data 402.
20 [0096]
A distribution unit 404 receives the encoded data 403 as an input, distributes
the data 403, and outputs data 405A and data 405B. Note that in Fig. 4, one encoder
is shown, but the number of encoders is not limited in this way. The present
invention may similarly be embodied when the number of encoders is m (where m is
25 an integer greater than or equal to one) and the distribution unit divides encoded data
generated by each encoder into two parts and outputs the divided data.
[0097]
Fig. 5 shows an example of a frame structure in the time domain for a
transmission device according to the present embodiment. A symbol 500-1 is a
symbol for notifying the reception device of the transmission method. For example,
the symbol 500-1 conveys information such as the error correction method used for
5 transmitting data symbols, the encoding ratio, and the modulation method used for
transmitting data symbols.
[0098]
The symbol 501-1 is for estimating channel fluctuation for the modulated
signal zl(t) (where t is time) transmitted by the transmission device. The symbol
10 502-1 is the data symbol transmitted as symbol number u (in the time domain) by
the modulated signal zl(t), and the symbol 503-1 is the data symbol transmitted as
symbol number u + 1 by the modulated signal zl (t).
[0099]
The symbol 501-2 is for estimating channel fluctuation for the modulated
15 signal z2(t) (where t is time) transmitted by the transmission device. The symbol
502-2 is the data symbol transmitted as symbol number u by the modulated signal
z2(t), and the symbol 503-2 is the data symbol transmitted as symbol number u + 1
by the modulated signal z2(t).
[O 1001
20 The following describes the relationships between the modulated signals
zl(t) and z2(t) transmitted by the transmission device and the received signals rl(t)
and r2(t) received by the reception device.
[OlOl]
In Fig. 5, 504#1 and 504#2 indicate transmit antennas in the transmission
25 device, and 505#1 and 505#2 indicate receive antennas in the reception device. The
transmission device transmits the modulated signal zl(t) from transmit antenna
504#1 and transmits the modulated signal z2(t) from transmit antenna 504#2. In this
case, the modulated signal zl(t) and the modulated signal z2(t) are assumed to
3 3
occupy the same (a shared/common) frequency (bandwidth). Letting the channel
fluctuation for the transmit antennas of the transmission device and the antennas of
the reception device be hll(t), hI2(t), h21(t), and h22(t), the signal received by the
receive antenna 505#1 of the reception device be rl (t), and the signal received by the
5 receive antenna 505#2 of the reception device be r2(t), the following relationship
holds.
[O 1 021
Math 36
Equation 36
[0 1 031
Fig. 6 relates to the weighting method (preceding method) in the present
embodiment. A weighting unit 600 integrates the weighting units 308A and 308B in
15 Fig. 3. As shown in Fig. 6, a stream sl(t) and a stream s2(t) correspond to the
baseband signals 307A and 307B in Fig. 3. In other words, the streams sl(t) and
s2(t) are the baseband signal in-phase components I and quadrature components Q
when mapped according to a modulation scheme such as QPSK, 16QAM, 64QAM,
or the like. As indicated by the fiame structure of Fig. 6, the stream sl(t) is
20 represented as sl (u) at symbol number u, as sl (u + 1) at symbol number u + 1, and
so forth. Similarly, the stream s2(t) is represented as s2(u) at symbol number u, as
s2(u + 1) at symbol number u + 1, and so forth. The weighting unit 600 receives the
baseband signals 307A (sl(t)) and 307B (s2(t)) and the information 3 15 regarding
weighting information in Fig. 3 as inputs, performs weighting in accordance with the
25 information 315 regarding weighting, and outputs the signals 309A (zl(t)) and 309B
(z2(t)) after weighting in Fig. 3. In this case, zl(t) and z2(t) are represented as
follows.
For symbol number 4i (where i is an integer greater than or equal to zero):
[0 1 041
5 Math37
Equation 37
[0 1051
10 Here, j is an imaginary unit.
For symbol number 4i + 1 :
[0 1 061
Math 38
Equation 3 8
[0 1071
For symbol number 4i + 2:
[0108]
20 Math 39
Equation 39
[0 1 091
For symbol number 4i + 3:
5 [OllO]
Math 40
Equation 40
In this way, the weighting unit in Fig. 6 regularly hops between precoding
weights over a four-slot period (cycle). (While precoding weights have been
described as being hopped between regularly over four slots, the number of slots for
regular hopping is not limited to four.)
15 Incidentally, Non-Patent Literature 4 describes switching the precoding
weights for each slot. This switching of precoding weights is characterized by being
random. On the other hand, in the present embodiment, a certain period (cycle) is
provided, and the precoding weights are hopped between regularly. Furthermore, in
each 2 x 2 precoding weight matrix composed of four precoding weights, the
20 absolute value of each of the four precoding weights is equivalent to (l/sqrt(2)), and
hopping is regularly performed between precoding weight matices having this
characteristic.
[0112]
In an LOS environment, if a special precoding matrix is used, reception
quality may greatly improve, yet the special precoding matrix differs depending on
the conditions of direct waves. In an LOS environment, however, a certain tendency
5 exists, and if precoding matrices are hopped between regularly in accordance with
this tendency, the reception quality of data greatly improves. On the other hand,
when precoding matrices are hopped between at random, a precoding matrix other
than the above-described special precoding matrix may exist, and the possibility of
performing precoding only with biased precoding matrices that are not suitable for
10 the LOS environment also exists. Therefore, in an LOS environment, excellent
reception quality may not always be obtained. Accordingly, there is a need for a
precoding hopping method suitable for an LOS environment. The present invention
proposes such a precoding method.
[0113]
15 Fig. 7 is an example of the structure of a reception device 700 in the present
embodiment. A wireless unit 703-X receives, as an input, a received signal 702-X
received by an antenna 701-X, performs processing such as fkequency conversion,
quadrature demodulation, and the like, and outputs a baseband signal 704-X.
[0114]
20 A channel fluctuation estimating unit 705-1 for the modulated signal zl
transmitted by the transmission device receives the baseband signal 704-X as an
input, extracts a reference symbol 501-1 for channel estimation as in Fig. 5,
estimates a value corresponding to hll in Equation 36, and outputs a channel
estimation signal 706-1.
25 [0115]
A channel fluctuation estimating unit 7052 for the modulated signal 22
transmitted by the transmission device receives the baseband signal 70- as an
input, extracts a reference symbol 501-2 for channel estimation as in Fig. 5,
estimates a value corresponding to h12 in Equation 36, and outputs a channel
estimation signal 706-2.
[0116]
A wireless unit 703-Y receives, as input, a received signal 702-Y received
5 by an antenna 701-Y, performs processing such as frequency conversion, quadrature
demodulation, and the like, and outputs a baseband signal 704-Y.
[0117]
A channel fluctuation estimating unit 707-1 for the modulated signal zl
transmitted by the transmission device receives the baseband signal 704-Y as an
10 input, extracts a reference symbol 501-1 for channel estimation as in Fig. 5,
estimates a value corresponding to h2, in Equation 36, and outputs a channel
estimation signal 708-1.
[01 181
A channel fluctuation estimating unit 707-2 for the modulated signal 22
15 transmitted by the transmission device receives the baseband signal 704-Y as an
input, extracts a reference symbol 501-2 for channel estimation as in Fig. 5,
estimates a value corresponding to h22 in Equation 36, and outputs a channel
estimation signal 708-2.
[0119]
20 A control information decoding unit 709 receives the baseband signal
704-X and the baseband signal 704-Y as inputs, detects the symbol 500-1 that
indicates the transmission method as in Fig. 5, and outputs a signal 710 regarding
information on the transmission method indicated by the transmission device.
[O 1201
25 A signal processing unit 711 receives, as inputs, the baseband signals
704-X and 704-Y, the channel estimation signals 706-1,706-2,708-1, and 708 - 2,
and the signal 710 regarding information on the transmission method indicated by
the transmission device, performs detection and decoding, and outputs received data
712-1 and 712-2.
[0121]
Next, operations by the signal processing unit 71 1 in Fig. 7 are described in
5 detail. Fig. 8 is an example of the structure of the signal processing unit 71 1 in the
present embodiment. Fig. 8 shows an INNER MIMO detector, a soft-inlsoft-out
decoder, and a weighting coefficient generating unit as the main elements.
Non-Patent Literature 2 and Non-Patent Literature 3 describe the method of iterative
decoding with this structure. The MIMO system described in Non-Patent Literature
10 2 and Non-Patent Literature 3 is a spatial multiplexing MIMO system, whereas the
present embodiment differs fiom Non-Patent Literature 2 and Non-Patent Literature
3 by describing a MIMO system that changes precoding weights with time. Letting
the (channel) matrix in Equation 36 be H(t), the precoding weight matrix in Fig. 6 be
W(t) (where the precoding weight matrix changes over t), the received vector be R(t)
15 = (rl(t),r2(t)lT, and the stream vector be S(t) = (sl(t),s2(t)lT, the following Equation
holds.
[0122]
Math 4 1
Equation 41
[0 1231
In this case, the reception device can apply the decoding method in
Non-Patent Literature 2 and Non-Patent Literature 3 to the received vector R(t) by
25 considering H(t)W(t) as the channel matrix.
Therefore, a weighting coefficient generating unit 819 in Fig. 8 receives, as
input, a signal 81 8 regarding information on the transmission method indicated by
the transmission device (corresponding to 710 in Fig. 7) and outputs a signal 820
regarding information on weighting coeflicients.
An INNER MIMO detector 803 receives the signal 820 regarding
information on weighting coefficients as input and, using the signal 820, performs
the calculation in Equation 41. Iterative detection and decoding is thus performed.
The following describes operations thereof.
10 [0126]
In the signal processing unit in Fig. 8, a processing method such as that
shown in Fig. 10 is necessary for iterative decoding (iterative detection). First, one
codeword (or one frame) of the modulated signal (stream) sl and one codeword (or
one frame) of the modulated signal (stream) s2 are decoded. As a result, the
15 Log-Likelihood Ratio (LLR) of each bit of the one codeword (or one fiame) of the
modulated signal (stream) sl and of the one codeword (or one fiarne) of the
modulated signal (stream) s2 is obtained from the soft-inlsoft-out decoder. Detection
and decoding is performed again using the LLR. These operations are performed
multiple times (these operations being referred to as iterative decoding (iterative
20 detection)). Hereinafter, description focuses on the method of generating the
log-likelihood ratio (LLR) of a symbol at a particular time in one M e .
[0 1271
In Fig. 8, a storage unit 815 receives, as inputs, a baseband signal 801X
(corresponding to the baseband signal 704-X in Fig. 7), a channel estimation signal
25 group 802X (corresponding to the channel estimation signals 706-1 and 706-2 in
Fig. 7), a baseband signal 801Y (corresponding to the baseband signal 704-Y in Fig.
7), and a channel estimation signal group 802Y (corresponding to the channel
estimation signals 708-1 and 708-2 in Fig. 7). In order to achieve iterative decoding
(iterative detection), the storage unit 81 5 calculates H(t)W(t) in Equation 41 and
stores the calculated matrix as a transformed channel signal group. The storage unit
815 outputs the above signals when necessary as a baseband signal 816X, a
transformed channel estimation signal group 81 7X, a baseband signal 816Y, and a
5 transformed channel estimation signal group 817Y.
[0128]
Subsequent operations are described separately for initial detection and for
iterative decoding (iterative detection).
[0 1291
10
The INNER MIMO detector 803 receives, as inputs, the baseband signal
801X, the channel estimation signal group 802X, the baseband signal 801Y, and the
channel estimation signal group 802Y. Here, the modulation method for the
modulated signal (stream) sl and the modulated signal (stream) s2 is described as
15 16QAM.
[0130]
The INNER MIMO detector 803 first calculates H(t)W(t) fiom the channel
estimation signal group 802X and the channel estimation signal group 802Y to seek
candidate signal points corresponding to the baseband signal 801X. Fig. 11 shows
20 such calculation. In Fig. 1 1, each black dot (a) is a candidate signal point in the IQ
plane. Since the modulation method is 16QAM, there are 256 candidate signal
points. (Since Fig. 11 is only for illustration, not all 256 candidate signal points are
shown.) Here, letting the four bits transferred by modulated signal sl be bO, bl, b2,
and b3, and the four bits transferred by modulated signal s2 be b4, b5, b6, and b7,
25 candidate signal points corresponding to (bO, bl, b2, b3, b4, b5, b6, b7) in Fig. 11
exist. The squared Euclidian distance is sought between a received signal point 1101
(corresponding to the baseband signal 801X) and each candidate signal point. Each
squared Euclidian distance is divided by the noise variance 02. Accordingly, Ex(bO,
bl, b2, b3, b4, b5, b6, b7), i.e. the value of the squared Euclidian distance between a
candidate signal point corresponding to (bO, bl, b2, b3, b4, b5, b6, b7) and a
received signal point, divided by the noise variance, is sought. Note that the
baseband signals and the modulated signals sl and s2 are each complex signals.
[0131]
Similarly, H(t)W(t) is calculated from the channel estimation signal group
802X and the channel estimation signal group 802Y, candidate signal points
corresponding to the baseband signal 801Y are sought, the squared Euclidian
distance for the received signal point (corresponding to the baseband signal 801Y) is
sought, and the squared Euclidian distance is divided by the noise variance 02.
Accordingly, Ey(bO, bl, b2, b3, b4, b5, b6, b7), i.e. the value of the squared
Euclidian distance between a candidate signal point corresponding to (bO, bl, b2, b3,
b4, b5, b6, b7) and a received signal point, divided by the noise variance, is sought.
[0 1321
Then Ex(bO, bl, b2, b3, b4, b5, b6, b7) + EaO, bl, b2, b3, b4, b5, b6, b7)
= E(b0, bl, b2, b3, b4, b5, b6, b7) is sought.
[0133]
The INNER MIMO detector 803 outputs E(b0, bl, b2, b3, b4, b5, b6, b7) as
a signal 804.
[0 1341
A log-likelihood calculating unit 805A receives the signal 804 as input,
calculates the log likelihood for bits bO, bl, b2, and b3, and outputs a log-likelihood
signal 806A. Note that during calculation of the log likelihood, the log likelihood for
"1" and the log likelihood for "0" are calculated. The calculation method is as shown
in Equations 28, 29, and 30. Details can be found in Non-Patent Literature 2 and
Non-Patent Literature 3.
[0135]
Similarly, a log-likelihood calculating unit 805B receives the signal 804 as
input, calculates the log likelihood for bits b4, b5, b6, and b7, and outputs a
log-likelihood signal 806B.
[0136]
5 A deinterleaver (807A) receives the log-likelihood signal 806A as an input,
performs deinterleaving corresponding to the interleaver (the interleaver (304A) in
Fig. 3), and outputs a deinterleaved log-likelihood signal 808A.
[0137]
Similarly, a deinterleaver (807B) receives the log-likelihood signal 806B as
10 an input, performs deinterleaving corresponding to the interleaver (the interleaver
(304B) in Fig. 3), and outputs a deinterleaved log-likelihood signal 808B.
[0138]
A log-likelihood ratio calculating unit 809A receives the interleaved
log-likelihood signal 808A as an input, calculates the log-likelihood ratio (LLR) of
15 the bits encoded by the encoder 302A in Fig. 3, and outputs a log-likelihood ratio
signal 810A.
[0139]
Similarly, a log-likelihood ratio calculating unit 809B receives the
interleaved log-likelihood signal 808B as an input, calculates the log-likelihood ratio
20 (LLR) of the bits encoded by the encoder 302B in Fig. 3, and outputs a
log-likelihood ratio signal 81 0B.
[0 1401
A soft-inlsoft-out decoder 81 1A receives the log-likelihood ratio signal
810A as an input, performs decoding, and outputs a decoded log-likelihood ratio
25 812A.
[0141]
Similarly, a soft-inlsoft-out decoder 81 1B receives the log-likelihood ratio
signal 810B as an input, performs decoding, and outputs a decoded log-likelihood
ratio 812B.
[0 1421
5
An interleaver (8 13A) receives the log-likelihood ratio 8 12A decoded by
the soft-inlsoft-out decoder in the (k- 1 )& iteration as an input, performs
interleaving, and outputs an interleaved log-likelihood ratio 814A. The interleaving
pattern in the interleaver (813A) is similar to the interleaving pattern in the
10 interleaver (304A) in Fig. 3.
[0 1431
An interleaver (8 13B) receives the log-likelihood ratio 8 12B decoded by the
soft-idsoft-out decoder in the (k - 1 )& iteration as an input, performs interleaving,
and outputs an interleaved log-likelihood ratio 814B. The interleaving pattern in the
15 interleaver (813B) is similar to the interleaving pattern in the interleaver (304B) in
Fig. 3.
[0 1441
The INNER MIMO detector 803 receives, as inputs, the baseband signal
8 16X, the transformed channel estimation signal group 817X, the baseband signal
20 816Y, the transformed channel estimation signal group 817Y, the interleaved
log-likelihood ratio 814A, and the interleaved log-likelihood ratio 814B. The reason
for using the baseband signal 816X, the transformed channel estimation signal group
817X, the baseband signal 816Y, and the transformed channel estimation signal
group 817Y instead of the baseband signal 801X, the channel estimation signal
25 group 802X, the baseband signal 801Y, and the channel estimation signal group
802Y is because a delay occurs due to iterative decoding.
[0 1451
The difference between operations by the INNER MIMO detector 803 for
iterative decoding and for initial detection is the use of the interleaved log-likelihood
ratio 814A and the interleaved log-likelihood ratio 814B during signal processing.
The INNER MIMO detector 803 first seeks E(b0, bl, b2, b3, b4, b5, b6, b7), as
5 during initial detection. Additionally, coefficients corresponding to Equations 11
and 32 are sought fiom the interleaved log-likelihood ratio 814A and the interleaved
log-likelihood ratio 914B. The value E(b0, bl, b2, b3, b4, b5, b6, b7) is adjusted
using the sought coefficients, and the resulting value E'(b0, bl, b2, b3, b4, b5, b6,
b7) is output as the signal 804.
10 [0146]
The log-likelihood calculating unit 805A receives the signal 804 as input,
calculates the log likelihood for bits bO, bl, b2, and b3, and outputs the
log-likelihood signal 806A. Note that during calculation of the log likelihood, the
log likelihood for "1" and the log likelihood for "0" are calculated. The calculation
15 method is as shown in Equations 31, 32, 33, 34, and 35. Details can be found in
Non-Patent Literature 2 and Non-Patent Literature 3.
[0 1471
Similarly, the log-likelihood calculating unit 805B receives the signal 804
as input, calculates the log likelihood for bits b4, b5, b6, and b7, and outputs the
20 log-likelihood signal 806B. Operations by the deinterleaver onwards are similar to
initial detection.
[0 1481
Note that while Fig. 8 shows the structure of the signal processing unit
when performing iterative detection, iterative detection is not always essential for
25 obtaining excellent reception quality, and a structure not including the interleavers
813A and 813B, which are necessary only for iterative detection, is possible. In such
a case, the INNER MIMO detector 803 does not perform iterative detection.
[0 1491
45
The main part of the present embodiment is calculation of H(t)W(t). Note
that as shown in Non-Patent Literature 5 and the like, QR decomposition may be
used to perform initial detection and iterative detection.
[0150]
5 Furthermore, as shown in Non-Patent Literature 11, based on H(t)W(t),
linear operation of the Minimum Mean Squared Error (MMSE) and Zero Forcing
(ZF) may be performed in order to perform initial detection.
[0151]
Fig. 9 is the structure of a different signal processing unit than Fig. 8 and is
10 for the modulated signal transmitted by the transmission device in Fig. 4. The
difference with Fig. 8 is the number of soft-inlsoft-out decoders. A soft-inlsoft-out
decoder 901 receives, as inputs, the log-likelihood ratio signals 810A and 810B,
performs decoding, and outputs a decoded log-likelihood ratio 902. A distribution
unit 903 receives the decoded log-likelihood ratio 902 as an input and distributes the
15 log-likelihood ratio 902. Other operations are similar to Fig. 8.
[0152]
Figs. 12A and 12B show BER characteristics for a transmission method
using the precoding weights of the present embodiment under similar conditions to
Figs. 29A and 29B. Fig. 12A shows the BER characteristics of Max-log A Posteriori
20 Probability (APP) without iterative detection (see Non-Patent Literature 1 and
Non-Patent Literature 2), and Fig. 12B shows the BER characteristics of
Max-log-APP with iterative detection (see Non-Patent Literature 1 and Non-Patent
Literature 2) (number of iterations: five). Comparing Figs. 12A, 12B, 29A, and 29B
shows how if the transmission method of the present embodiment is used, the BER
25 characteristics when the Rician factor is large greatly improve over the BER
characteristics when using spatial multiplexing MIMO system, thereby confirming
the usefulness of the method in the present embodiment.
[0153]
46
As described above, when a transmission device transmits a plurality of
modulated signals fiom a plurality of antennas in a MIMO system, the advantageous
effect of improved transmission quality, as compared to conventional spatial
multiplexing MIMO system, is achieved in an LOS environment in which direct
5 waves dominate by hopping between precoding weights regularly over time, as in
the present embodiment.
[0 1 541
In the present embodiment, and in particular with regards to the structure of
the reception device, operations have been described for a limited number of
10 antennas, but the present invention may be embodied in the same way even if the
number of antennas increases. In other words, the number of antennas in the
reception device does not affect the operations or advantageous effects of the present
embodiment. Furthermore, in the present embodiment, the example of LDPC coding
has particularly been explained, but the present invention is not limited to LDPC
15 coding. Furthermore, with regards to the decoding method, the soft-inlsoft-out
decoders are not limited to the example of sum-product decoding. Another
soft-inlsoft-out decoding method may be used, such as a BCJR algorithm, a SOVA
algorithm, a Max-log-MAP algorithm, and the like. Details are provided in
Non-Patent Literature 6.
20 [0155]
Additionally, in the present embodiment, the example of a single carrier
method has been described, but the present invention is not limited in this way and
may be similarly embodied for multi-carrier transmission. Accordingly, when using
a method such as spread spectrum communication, Orthogonal Frequency-Division
25 Multiplexing (OFDM), Single Carrier Frequency Division Multiple Access
(SC-FDMA), Single Carrier Orthogonal Frequency-Division Multiplexing
(SC-OFDM), or wavelet OFDM as described in Non-Patent Literature 7 and the like,
for example, the present invention may be similarly embodied. Furthermore, in the
present embodiment, symbols other than data symbols, such as pilot symbols
(preamble, unique word, and the like), symbols for transmission of control
information, and the like, may be arranged in the fiame in any way.
[0156]
5 The following describes an example of using OFDM as an example of a
multi-cariier method.
[0157]
Fig. 13 shows the structure of a transmission device when using OFDM. In
Fig. 13, elements that operate in a similar way to Fig. 3 bear the same reference
10 signs.
[0158]
An OFDM related processor 1301A receives, as input, the weighted signal
309A, performs processing related to OFDM, and outputs a transmission signal
1302A. Similarly, an OFDM related processor 1301B receives, as input, the
15 weighted signal 309B, performs processing related to OFDM, and outputs a
transmission signal 1302B.
[0159]
Fig. 14 shows an example of a structure fiom the OFDM related processors
1301A and 1301B in Fig. 13 onwards. The part fiom 1401A to 1410A is related to
20 the part fiom 1301A to 312A in Fig. 13, and the part fiom 1401B to 1410B is related
to the part fiom 1301B to 312B in Fig. 13.
[0 1601
A seriallparallel converter 1402A performs seriaVparalle1 conversion on a
weighted signal 1401A (corresponding to the weighted signal 309A in Fig. 13) and
25 outputs a parallel signal 1403A.
[0161]
A reordering unit 1404A receives a parallel signal 1403A as input, performs
reordering, and outputs a reordered signal 1405A. Reordering is described in detail
later.
[0 1 621
5 An inverse fast Fourier transformer 1406A receives the reordered signal
1405A as an input, performs a fast Fourier transform, and outputs a fast Fourier
transformed signal 1407A.
[0 1 631
A wireless unit 1408A receives the fast Fourier transformed signal 1407A
10 as an input, performs processing such as fiequency conversion, amplification, and
the like, and outputs a modulated signal 1409A. The modulated signal 1409A is
output as a radio wave fiom an antenna 14 10A.
101 641
A seriallparallel converter 1402B performs seriaVparalle1 conversion on a
15 weighted signal 1401B (corresponding to the weighted signal 309B in Fig. 13) and
outputs a parallel signal 1403B.
[0 1 651
A reordering unit 1404B receives a parallel signal 1403B as input, performs
reordering, and outputs a reordered signal 1405B. Reordering is described in detail
20 later.
[0 1661
An inverse fast Fourier transformer 1406B receives the reordered signal
1405B as an input, performs a fast Fourier transform, and outputs a fast Fourier
transformed signal 1407B.
25 [0167]
A wireless unit 1408B receives the fast Fourier transformed signal 1407B as
an input, performs processing such as fiequency conversion, amplification, and the
like, and outputs a modulated signal 1409B. The modulated signal 1409B is output
as a radio wave fiom an antenna 14 10B.
[0 1 681
In the transmission device of Fig. 3, since the transmission method does not
5 use multi-carrier, precoding hops to form a four-slot period (cycle), as shown in Fig.
6, and the precoded symbols are arranged in the time domain. When using a
multi-carrier transmission method as in the OFDM method shown in Fig. 13, it is of
course possible to arrange the precoded symbols in the time domain as in Fig. 3 for
each (sub)carrier. In the case of a multi-carrier transmission method, however, it is
10 possible to arrange symbols in the fiequency domain, or in both the fiequency and
time domains. The following describes these arrangements.
[0 1691
Figs. 15A and 15B show an example of a method of reordering symbols by
reordering units 1401A and 1401B in Fig. 14, the horizontal axis representing
15 fiequency, and the vertical axis representing time. The fiequency domain runs fiom
(sub)carrier 0 through (sub)carrier 9. The modulated signals zl and 22 use the same
fiequency bandwidth at the same time. Fig. 15A shows the reorde;ing method for
symbols of the modulated signal zl, and Fig. 15B shows the reordering method for
symbols of the modulated signal 22. Numbers #1, #2, #3, #4, . . . are assigned to in
20 order to the symbols of the weighted signal 1401A which is input into the
seriaVparalle1 converter 1402A. At this point, symbols are assigned regularly, as
shown in Fig. 15A. The symbols #1, #2, #3, #4, ... are arranged in order starting
fiom carrier 0. The symbols #1 through #9 are assigned to time $1, and subsequently,
the symbols # 10 through # 19 are assigned to time $2.
25 [0170]
Similarly, numbers #1, #2, #3, #4, . . . are assigned in order to the symbols of
the weighted signal 1401B which is input into the seriaVparalle1 converter 1402B.
At this point, symbols are assigned regularly, as shown in Fig. 15B. The symbols #1,
50
#2, #3, #4, . . . are arranged in order starting fiom carrier 0. The symbols #I through
#9 are assigned to time $1, and subsequently, the symbols #10 through #19 are
assigned to time $2. Note that the modulated signals zl and 22 are complex signals.
[0171]
5 The symbol group 1501 and the symbol group 1502 shown in Figs. 15A and
15B are the symbols for one period (cycle) when using the precoding weight
hopping method shown in Fig. 6. Symbol #O is the symbol when using the precoding
weight of slot 4i in Fig. 6. Symbol #1 is the symbol when using the precoding
weight of slot 4i + 1 in Fig. 6. Symbol #2 is the symbol when using the precoding
10 weight of slot 4i + 2 in Fig. 6. Symbol #3 is the symbol when using the precoding
weight of slot 4i + 3 in Fig. 6. Accordingly, symbol #x is as follows. When x mod 4
is 0, the symbol #x is the symbol when using the precoding weight of slot 4i in Fig.
6. When x mod 4 is 1, the symbol #x is the symbol when using the precoding weight
of slot 4i + 1 in Fig. 6. When x mod 4 is 2, the symbol #x is the symbol when using
15 the precoding weight of slot 4i + 2 in Fig. 6. When x mod 4 is 3, the symbol #x is
the symbol when using the precoding weight of slot 4i + 3 in Fig. 6.
[0 1 721
In this way, when using a multi-carrier transmission method such as OFDM,
unlike during single carrier transmission, symbols can be arranged in the frequency
20 domain. Furthermore, the ordering of symbols is not limited to the ordering shown
in Figs. 15A and 15B. Other examples are described with reference to Figs. 16A,
16B, 17A, and 17B.
[0 1 731
Figs. 16A and 16B show an example of a method of reordering symbols by
25 the reordering units 1404A and 1404B in Fig. 14, the horizontal axis representing
fkequency, and the vertical axis representing time, that differs fiom Figs. 15A and
15B. Fig. 16A shows the reordering method for symbols of the modulated signal zl,
and Fig. 16B shows the reordering method for symbols of the modulated signal 22.
The difference in Figs. 16A and 16B as compared to Figs. 15A and 15B is that the
reordering method of the symbols of the modulated signal zl differs fiom the
reordering method of the symbols of the modulated signal 22. In Fig. 16B, symbols
#O through #5 are assigned to carriers 4 through 9, and symbols #6 through #9 are
5 assigned to carriers 0 through 3. Subsequently, symbols #10 through #19 are
assigned regularly in the same way. At this point, as in Figs. 15A and 15B, the
symbol group 1601 and the symbol group 1602 shown in Figs. 16A and 16B are the
symbols for one period (cycle) when using the precoding weight hopping method
shown in Fig. 6.
10 [0174]
Figs. 17A and 17B show an example of a method of reordering symbols by
the reordering units 1404A and 1404B in Fig. 14, the horizontal axis representing
fiequency, and the vertical axis representing time, that differs fkom Figs. 15A and
15B. Fig. 17A shows the reordering method for symbols of the modulated signal zl,
15 and Fig. 17B shows the reordering method for symbols of the modulated signal 22.
The difference in Figs. 17A and 17B as compared to Figs. 15A and 15B is that
whereas the symbols are arranged in order by carrier in Figs. 15A and 15B, the
symbols are not arranged in order by carrier in Figs. 17A and 17B. It is obvious that,
in Figs. 17A and 17B, the reordering method of the symbols of the modulated signal
20 zl may differ fiom the reordering method of the symbols of the modulated signal 22,
as in Figs. 16A and 16B.
[0 1751
Figs. 18A and 18B show an example of a method of reordering symbols by
the reordering units 1404A and 1404B in Fig. 14, the horizontal axis representing
25 fiequency, and the vertical axis representing time, that differs fiom Figs. 15A
through 17B. Fig. 18A shows the reordering method for symbols of the modulated
signal zl, and Fig. 18B shows the reordering method for symbols of the modulated
signal 22. In Figs. 15A through 17B, symbols are arranged in the fiequency domain,
52
whereas in Figs. 18A and 18B, symbols are arranged in both the fiequency and time
domains.
[0 1 761
In Fig. 6, an example has been described of hopping between precoding
5 weights over four slots. Here, however, an example of hopping over eight slots is
described. The symbol groups 1801 and 1802 shown in Figs. 18A and 18B are the
symbols for one period (cycle) when using the precoding weight hopping method
(and are therefore eight-symbol groups). Symbol #O is the symbol when using the
precoding weight of slot 8i. Symbol #1 is the symbol when using the precoding
10 weight of slot 8i + 1. Symbol #2 is the symbol when using the precoding weight of
slot 8i + 2. Symbol #3 is the symbol when using the precoding weight of slot 8i + 3.
Symbol #4 is the symbol when using the precoding weight of slot 8i + 4. Symbol #5
is the symbol when using the precoding weight of slot 8i + 5. Symbol #6 is the
symbol when using the precoding weight of slot 8i + 6. Symbol #7 is the symbol
15 when using the precoding weight of slot 8i + 7. Accordingly, symbol #x is as
follows. When x mod 8 is 0, the symbol #x is the symbol when using the precoding
weight of slot 8i. When x mod 8 is 1, the symbol #x is the symbol when using the
precoding weight of slot 8i + 1. When x mod 8 is 2, the symbol #x is the symbol
when using the precoding weight of slot 8i + 2. When x mod 8 is 3, the symbol #x is
20 the symbol when using the precoding weight of slot 8i + 3. When x mod 8 is 4, the
symbol #x is the symbol when using the precoding weight of slot 8i + 4. When x
mod 8 is 5, the symbol #x is the symbol when using the precoding weight of slot 8i
+ 5. When x mod 8 is 6, the symbol #x is the symbol when using the precoding
weight of slot 8i + 6. When x mod 8 is 7, the symbol #x is the symbol when using
25 the precoding weight of slot 8i + 7. In the symbol ordering in Figs. 18A and 18B,
four slots in the time domain and two slots in the fiequency domain for a total of 4 x
2 = 8 slots are used to arrange symbols for one period (cycle). In this case, letting
the number of symbols in one period (cycle) be m x n symbols (in other words, m x
n precoding weights exist), the number of slots (the number of carriers) in the
frequency domain used to arrange symbols in one period (cycle) be n, and the
number of slots used in the time domain be m, m should be greater than n. This is
because the phase of direct waves fluctuates more slowly in the time domain than in
5 the frequency domain. Therefore, since the precoding weights are changed in the
present embodiment to minimize the influence of steady direct waves, it is
preferable to reduce the fluctuation in direct waves in the period (cycle) for changing
the precoding weights. Accordingly, m should be greater than n. Furthermore,
considering the above points, rather than reordering symbols only in the frequency
10 domain or only in the time domain, direct waves are more likely to become stable
when symbols are reordered in both the frequency and the time domains as in Figs.
18A and 18B, thereby making it easier to achieve the advantageous effects of the
present invention. When symbols are ordered in the frequency domain, however,
fluctuations in the frequency domain are abrupt, leading to the possibility of yielding
15 diversity gain. Therefore, reordering in both the frequency and the time domains is
not necessarily always the best method.
[0 1771
Figs. 19A and 19B show an example of a method of reordering symbols by
the reordering units 1404A and 1404B in Fig. 14, the horizontal axis representing
20 frequency, and the vertical axis representing time, that differs from Figs. 18A and
18B. Fig. 19A shows the reordering method for symbols of the modulated signal zl,
and Fig. 19B shows the reordering method for symbols of the modulated signal 22.
As in Figs. 18A and 1 8B, Figs. 19A and 19B show arrangement of symbols using
both the frequency and the time axes. The difference as compared to Figs. 18A and
25 18B is that, whereas symbols are arranged first in the frequency domain and then in
the time domain in Figs. 18A and 18B, symbols are arranged first in the time
domain and then in the frequency domain in Figs. 19A and 19B. In Figs. 19A and
19B, the symbol group 1901 and the symbol group 1902 are the symbols for one
period (cycle) when using the precoding hopping method.
[0178]
Note that in Figs. 18A, 18B, 19A, and 19B, as in Figs. 16A and 16B, the
5 present invention may be similarly embodied, and the advantageous effect of high
reception quality achieved, with the symbol arranging method of the modulated
signal zl differing fiom the symbol arranging method of the modulated signal 22.
Furthermore, in Figs. 1 SA, 1 SB, 19A, and 19B, as in Figs. 17A and 17B, the present
invention may be similarly embodied, and the advantageous effect of high reception
10 quality achieved, without arranging the symbols in order.
[0 1 791
Fig. 27 shows an example of a method of reordering symbols by the
reordering units 1404A and 1404B in Fig. 14, the horizontal axis representing
fiequency, and the vertical axis representing time, that differs fiom the above
15 examples. The case of hopping between precoding matrix regularly over four slots,
as in Equations 37-40, is considered. The characteristic feature of Fig. 27 is that
symbols are arranged in order in the fiequency domain, but when progressing in the
time domain, symbols are cyclically shifted by n symbols (in the example in Fig. 27,
n = 1). In the four symbols shown in the symbol group 2710 in the fiequency
20 domain in Fig. 27, precoding hops between the precoding matrices of Equations
3740.
[0 1 801
In this case, symbol #O is precoded using the precoding matrix in Equation
37, symbol #1 is precoded using the precoding matrix in Equation 38, symbol #2 is
25 precoded using the precoding matrix in Equation 39, and symbol #3 is precoded
using the precoding matrix in Equation 40.
[0181]
Similarly, for the symbol group 2720 in the frequency domain, symbol #4 is
precoded using the precoding matrix in Equation 37, symbol #5 is precoded using
the precoding matrix in Equation 38, symbol #6 is precoded using the precoding
matrix in Equation 39, and symbol #7 is precoded using the precoding matrix in
5 Equation 40.
[0 1 821
For the symbols at time $1, precoding hops between the above precoding
matrices, but in the time domain, symbols are cyclically shifted. Therefore,
precoding hops between precoding matrices for the symbol groups 2701,2702,2703,
10 and 2704 as follows.
[0 1 831
In the symbol group 2701 in the time domain, symbol #O is precoded using
the precoding matrix in Equation 37, symbol #9 is precoded using the precoding
matrix in Equation 38, symbol #18 is precoded using the precoding matrix in
15 Equation 39, and symbol #27 is precoded using the precoding matrix in Equation 40.
[O 1 841
In the symbol group 2702 in the time domain, symbol #28 is precoded using
the precoding matrix in Equation 37, symbol #1 is precoded using the precoding
matrix in Equation 38, symbol #10 is precoded using the precoding matrix in
20 Equation 39, and symbol # 19 is precoded using the precoding matrix in Equation 40.
[0185]
In the symbol group 2703 in the time domain, symbol #20 is precoded using
the precoding matrix in Equation 37, symbol #29 is precoded using the precoding
matrix in Equation 38, symbol #2 is precoded using the precoding matrix in
25 Equation 39, and symbol #l 1 is precoded using the precoding matrix in Equation 40.
[0 1 861
In the symbol group 2704 in the time domain, symbol #12 is precoded using
the precoding matrix in Equation 37, symbol #21 is precoded using the precoding
matrix in Equation 38, symbol #30 is precoded using the precoding matrix in
Equation 39, and symbol #3 is precoded using the precoding matrix in Equation 40.
[0 1 871
The characteristic of Fig. 27 is that, for example focusing on symbol #11,
5 the symbols on either side in the fi-equency domain at the same time (symbols #10
and #12) are both precoded with a different precoding matrix than symbol #11, and
the symbols on either side in the time domain in the same carrier (symbols #2 and
#20) are both precoded with a different precoding matrix than symbol #11. This is
true not only for symbol #11. Any symbol having symbols on either side in the
10 fi-equency domain and the time domain is characterized in the same way as symbol
#I 1. As a result, precoding matrices are effectively hopped between, and since the
influence on stable conditions of direct waves is reduced, the possibility of improved
reception quality of data increases.
[0188]
15 In Fig. 27, the case of n = 1 has been described, but n is not limited in this
way. The present invention may be similarly embodied with n = 3. Furthermore, in
Fig. 27, when symbols are arranged in the fi-equency domain and time progresses in
the time domain, the above characteristic is achieved by cyclically shifting the
number of the arranged symbol, but the above characteristic may also be achieved
20 by randomly (or regularly) arranging the symbols.
[0 1 891
(Embodiment 2)
In Embodiment 1, regular hopping of the precoding weights as shown in Fig.
6 has been described. In the present embodiment, a method for designing specific
25 precoding weights that differ fiom the precoding weights in Fig. 6 is described.
[0 1901
In Fig. 6, the method for hopping between the precoding weights in
Equations 37-40 has been described. By generalizing this method, the precoding
weights may be changed as follows. (The hopping period (cycle) for the precoding
weights has four slots, and Equations are listed similarly to Equations 37-40.)
For symbol number 4i (where i is an integer greater than or equal to zero):
[0191]
5 Math 42
Equation 42
[0 1 921
10 Here, j is an imaginary unit.
For symbol number 4i + 1 :
[0 1931
Math 43
Equation 43
[0 1 941
For symbol number 4i + 2:
[0 1 951
20 Math 44
Equation 44
[0 1 961
For symbol number 4i + 3:
[0 1971
5 Math45
Equation 45
, [0198]
10 From Equations 36 and 41, the received vector R(t) = (rl(t), r2(t)lT can be
represented as follows.
For symbol number 4i:
[0 1 991
Math 46
15 Equation 46
[0200]
For symbol number 4i + 1 :
20 [0201]
Math 47
Equation 47
[0202]
For symbol number 4i + 2:
5 [0203]
Math 48
Equation 48
10 [0204]
For symbol number 4i + 3:
[0205]
Math 49
Equation 49
[0206]
In this case, it is assumed that only components of direct waves exist in the
channel elements hll(t), hI2(t), h21(t), and hu(t), that the amplitude components of
20 the direct waves are all equal, and that fluctuations do not occur over time. With
these assumptions, Equations 46-49 can be represented as follows.
For symbol number 4i:

CLAIMS
1. A preceding method for generating, from a plurality of baseband signals, a
plurality of precoded signals to be transmitted over the same frequency bandwidth at
the same time, comprising the steps of:
5 selecting a matrix F[i] from among N matrices while switching between the
N matrices, the N matrices defining preceding performed on the plurality of
baseband signals, i being an integer from 0 to N - 1, and N being an integer at least
two; and
generating a first precoded signal zl and a second precoded signal z2 by
10 preceding, in accordance with the selected matrix F[i], a first baseband signal si
generated from a first plurality of bits and a second baseband signal s2 generated
from a second plurality of bits, a first encoded block and a second encoded block
being generated respectively as the first plurality of bits and the second plurality of
bits using a predetermined error correction block encoding method, the first
15 baseband signal si and the second baseband signal s2 being generated respectively
from the first encoded block and the second encoded block to have M symbols each,
the first precoded signal zl and the second precoded signal z2 being generated to
have M slots each by preceding a combination of the first baseband signal si and the
second baseband signal s2, M being an integer at least two,
20 the first preceded signal zl and the second precoded signal z2 satisfying the
equation (zl, z2f = F[i](sl, s2)'^.
2. A preceding apparatus for generating, from a plurality of baseband signals, a
plurality of precoded signals to be fransmitted ever the same frequency bandwidth at
25 the same time, comprising:
a weighting information generation imit configured to select a mafrix F[i]
from among N matrices while switching between the N mafrices, the N matrices
298
defining preceding performed on the plurality of baseband signals, i being an integer
firom 0 to N - 1, and N being an integer at least two;
a weighting unit configured to generate a first precoded signal zl and a
second precoded signal z2 by precoding, in accordance with the selected matrix F[i],
5 a first baseband signal si generated fi-om a first plurality of bits an<^ a second
baseband signal s2 generated fi-om a second plurality of bits;
an error correction coding unit configured to generate a first encoded block
as the first plurality of bits and a second encoded block as the second plurality of
bits using a predetermined error correction block encoding method; and
10 a mapper configured to generate a baseband signal with M symbols fi-om
the first encoded block and a baseband signal with M symbols fi-om the second
encoded block, M being an integer at least two,
the first precoded signal zl and the second precoded signal z2 satisfying the
equation (z 1, Tlf = F [i] (s 1, s2)^, and
15 the weighting unit generating precoded signals with M slots by precoding a
combination of the baseband signal generated fi-om the first encoded block and the
baseband signal generated fi-om the second encoded block.