Watermark Generator, Watermark Decoder, Method for Providing a Watermark
Signal in Dependence on Binary Message Data, Method for Providing Binary
Message Data in Dependence on a Watermarked Signal and Computer Program
Using a Differential Encoding
Description
Technical Field
Embodiments according to the invention are related to a watermark generator for providing
a watermark signal in dependence on binary message data. Further embodiments according
to the invention relate to a watermark decoder for providing binary message data in
dependence on a watermarked signal. Further embodiments according to the invention are
related to a method for providing a watermark signal in dependence on binary message
data. Further embodiments according to the invention are related to a method for providing
binary message data in dependence on a watermarked signal. Further embodiments are
related to corresponding computer programs.
Some embodiments according to the invention are related to a robust low complexity audio
watermarking system.
Background of the Invention
In many technical applications, it is desired to include an extra information into an
information or signal representing useful data or "main data" like, for example, an audio
signal, a video signal, graphics, a measurement quantity and so on. In many cases, it is
desired to include the extra information such that the extra information is bound to the
main data (for example, audio data, video data, still image data, measurement data, text
data, and so on) in a way that it is not perceivable by a user of said data. Also, in some
cases it is desirable to include the extra data such that the extra data are not easily
removable from the main data (e.g. audio data, video data, still image data, measurement
data, and so on).
This is particularly true in applications in which it is desirable to implement a digital rights
management. However, it is sometimes simply desired to add substantially unperceivable
side information to the useful data. For example, in some cases it is desirable to add side
information to audio data, such that the side information provides an information about the
source of the audio data, the content of the audio data, rights related to the audio data and
so on.
For embedding extra data into useful data or "main data", a concept called "watermarking"
may be used. Watermarking concepts have been discussed in the literature for many
different kinds of useful data, like audio data, still image data, video data, text data, and so
on.
In the following, some references will be given in which watermarking concepts are
discussed. However, the reader's attention is also drawn to the wide field of textbook
literature and publications related to the watermarking for further details.
DE 196 40 814 C2 describes a coding method for introducing a non-audible data signal
into an audio signal and a method for decoding a data signal, which is included in an audio
signal in a non-audible form. The coding method for introducing a non-audible data signal
into an audio signal comprises converting the audio signal into the spectral domain. The
coding method also comprises determining the masking threshold of the audio signal and
the provision of a pseudo noise signal. The coding method also comprises providing the
data signal and multiplying the pseudo noise signal with the data signal, in order to obtain
a frequency-spread data signal. The coding method also comprises weighting the spread
data signal with the masking threshold and overlapping the audio signal and the weighted
data signal.
In addition, WO 93/07689 describes a method and apparatus for automatically identifying
a program broadcast by a radio station or by a television channel, or recorded on a
medium, by adding an inaudible encoded message to the sound signal of the program, the
message identifying the broadcasting channel or station, the program and/or the exact date.
In an embodiment discussed in said document, the sound signal is transmitted via an
analog-to-digital converter to a data processor enabling frequency components to be split
up, and enabling the energy in some of the frequency components to be altered in a
predetermined manner to form an encoded identification message. The output from the
data processor is connected by a digital-to-analog converter to an audio output for
broadcasting or recording the sound signal. In another embodiment discussed in said
document, an analog bandpass is employed to separate a band of frequencies from the
sound signal so that energy in the separated band may be thus altered to encode the sound
signal.
US 5, 450,490 describes apparatus and methods for including a code having at least one
code frequency component in an audio signal. The abilities of various frequency
components in the audio signal to mask the code frequency component to human hearing
are evaluated and based on these evaluations an amplitude is assigned to the code
frequency component. Methods and apparatus for detecting a code in an encoded audio
signal are also described. A code frequency component in the encoded audio signal is
detected based on an expected code amplitude or on a noise amplitude within a range of
audio frequencies including the frequency of the code component.
WO 94/1 1989 describes a method and apparatus for encoding/decoding broadcast or
recorded segments and monitoring audience exposure thereto. Methods and apparatus for
encoding and decoding information in broadcasts or recorded segment signals are
described. In an embodiment described in the document, an audience monitoring system
encodes identification information in the audio signal portion of a broadcast or a recorded
segment using spread spectrum encoding. The monitoring device receives an acoustically
reproduced version of the broadcast or recorded signal via a microphone, decodes the
identification information from the audio signal portion despite significant ambient noise
and stores this information, automatically providing a diary for the audience member,
which is later uploaded to a centralized facility. A separate monitoring device decodes
additional information from the broadcast signal, which is matched with the audience diary
information at the central facility. This monitor may simultaneously send data to the
centralized facility using a dial-up telephone line, and receives data from the centralized
facility through a signal encoded using a spread spectrum technique and modulated with a
broadcast signal from a third party.
WO 95/27349 describes apparatus and methods for including codes in audio signals and
decoding. An apparatus and methods for including a code having at least one code
frequency component in an audio signal are described. The abilities of various frequency
components in the audio signal to mask the code frequency component to human hearing
are evaluated, and based on these evaluations, an amplitude is assigned to the code
frequency components. Methods and apparatus for detecting a code in an encoded audio
signal are also described. A code frequency component in the encoded audio signal is
detected based on an expected code amplitude or on a noise amplitude within a range of
audio frequencies including the frequency of the code component.
However, in the known watermarking systems, reliability issues arise if the watermarked
signal is affected by a Doppler shift, which may occur, for example, because of a
movement of an apparatus receiving the watermarked signal, or in the case of a mismatch
of local oscillators at the watermark generator side and the watermark decoder side.
In view of this situation, it is an object of the present invention to create a watermarking
concept and a watermark detection concept, which allows for an improved reliability in the
case that a Doppler frequency shift affects the watermark signal or in the case that there is
a frequency deviation between the local oscillators of the watermark generator and the
watermark decoder.
Summary of the Invention
This object is achieved by a watermark generator according to claim 1, a watermark
decoder according to claims 7 or 8, a method for providing a watermark signal in
dependence on binary message data according to claim 13, a method for providing binary
message data in dependence on a watermarked signal according to claim 14 and a
computer program according to claim 15.
An embodiment according to the invention creates a watermark generator for providing a
watermark signal in dependence on binary message data. The watermark generator
comprises an information processor configured to provide, in dependence on information
units (e.g. bits) of the binary message data, a first time-frequency-domain representation,
values of which represent the binary message data. The watermark generator also
comprises a differential encoder configured to derive a second time-frequency-domain
representation from the first time-frequency-domain representation, such that the second
time-frequency-domain representation comprises a plurality of values, wherein a difference
between two values of the second time-frequency-domain representation represents a
corresponding value of the first time-frequency-domain representation, in order to obtain a
difference encoding of the values of the first time-frequency-domain representation. The
watermark generator also comprises a watermark signal provider configured to provide the
watermark signal on the basis of the second time-frequency-domain representation.
It is the idea of the present invention that a watermark signal is particularly robust with
respect to a degradation, for example, by the Doppler effect, if adjacent time-frequencydomain
values (for example associated with adjacent frequency bands or bit intervals) are
encoded such that a difference between the characteristics of such adjacent signal portions,
which characteristics are represented by the values of the second time-frequency-domain
representation, allows to uniquely conclude to the corresponding value of the first timefrequency-
domain representation. In other words, the differential encoding in the timefrequency-
domain allows for the generation of a robust watermarked signal, for example,
by providing a time-frequency-domain audio signal, the watermark audio content of which
is determined by the second time-frequency-domain representation.
Thus, improved robustness against movement and frequency mismatch of the local
oscillators is achieved by the differential modulation. In fact, the Doppler effect, which is
caused, for example, by a movement of a signal transducer providing the watermarked
audio signal to a watermark decoder, and frequency mismatches lead to a rotation of a
modulation constellation, for example a binary phase-shift keying (BPSK) constellation.
The detrimental effects of this Doppler shift or frequency mismatch can be reduced or
entirely eliminated by the differential encoding. Thus, the differential encoding has the
effect that the watermarked signal, which is provided on the basis of the second timefrequency-
domain representation, is insensitive with respect to a rotation of the bits in a
complex plane.
In a preferred embodiment, the information processor is configured to provide the first
time-frequency-domain representation such that the values of the first time-frequencydomain
representation represent the binary message data in the form of a binary pattern. In
this case, the differential encoder is configured to derive the second time-frequencydomain
representation such that there is a phase change between two subsequent values of
the second time-frequency-domain representation if a corresponding value of the first timefrequency-
domain representation takes a first value, and such that there is no phase change
between subsequent values of the second time-frequency-domain representation if a
corresponding value of the first time-frequency-domain representation takes a second
value, which is different from a first value.
In a preferred embodiment, the watermark signal provider is configured to provide an
audio signal on the basis of the second time-frequency-domain representation, such that a
watermark frequency component of the watermark signal comprises a step-wise or a
smooth phase change in response to a first value of the first time-frequency-domain
representation, and such that a watermark frequency component of the watermark signal
comprises a temporally constant phase in response to a second value of the first timefrequency-
domain representation, which is different from the first value.
In a preferred embodiment, the watermark signal provider is configured to provide a first
bit-shaping waveform in response to a first value of the second time-frequency-domain
representation, and to provide a second bit-shaping waveform in response to a second
value of the second time-frequency-domain representation. The watermark signal provider
is configured to include into the watermark signal a weighted or non-weighted
superposition of time-shifted versions of the same bit-shaping waveform in response to the
presence of a first value in the first time-frequency-domain representation, and to include
into the watermarked signal a weighted or a non-weighed superposition of time-shifted
versions of a first bit-shaping waveform and of a second bit-shaping waveform in response
to the presence of a second value, which is different from the first value, in the first timefrequency-
domain representation. This embodiment brings along the advantage that the
sum (or superposition) of time-shifted versions of the same bit-shaping waveform can be
distinguished easily from a sum (or superposition) of a first bit-shaping waveform and a
second bit-shaping waveform, if the bit-shaping waveforms are sufficiently different. As
subsequent bit-shaping waveforms are affected by a channel, via which the watermarked
signal is transmitted, in the same or at least approximately the same manner, it is simple to
conclude to the value of the first time-frequency-domain representation, because the
reception of two identical (or approximately identical) bit-shaping waveforms allows the
conclusion that the value of the first time-frequency-domain representation was in first
state (e.g. +1). Similarly, the reception of any two significantly different bit-shaping
waveforms allows the conclusion that the value of the first time-frequency-domain
representation was in the second state (e.g. -1).
In a preferred embodiment, the second bit-shaping waveform is an inverse version of the
first bit-shaping waveform. This allows to easily conclude to the value of the first timefrequency-
domain representation with a minimum filtering effort and/or correlation effort.
A preferred embodiment of the invention creates a watermarked decoder for providing
binary message data in dependence on a watermarked signal. The watermark decoder
comprises a time-frequency-domain representation provider configured to provide a first
time-frequency-domain representation of the watermarked signal. The watermark decoder
also comprises a differential decoder configured to derive a second time-frequency-domain
representation from the first time-frequency-domain representation, such that values of the
second time-frequency-domain representation are dependent on phase differences between
two corresponding (and preferably adjacent) values of the first time-frequency-domain
representation. The watermark decoder also comprises a synchronization determinator
configured to obtain a synchronization information on the basis of the second timefrequency-
domain representation. The watermark decoder also comprises a watermark
extractor configured to extract the binary message data from the first time-frequencydomain
representation of the watermarked signal or from the second time-frequencydomain
representation of the watermarked signal using the synchronization information.
Another embodiment according to the invention creates a watermark decoder for providing
binary message data in dependence on a watermarked signal. The watermark decoder
comprises a time-frequency-domain representation provider configured to provide a first
time-frequency-domain representation of the watermarked signal and a differential
decoder. The differential decoder is configured to derive a second time-frequency-domain
representation from the first time-frequency-domain representation, such that values of the
second time-frequency-domain representation are dependent on phase differences between
two corresponding values of the first time-frequency-domain representation. The
watermark decoder also comprises a watermark extractor configured to extract the binary
message data from the second time-frequency-domain representation.
These embodiments according to the invention are based on the finding that the reliability
of a watermark decoding can be improved by evaluating phase differences between
adjacent values of a first time-frequency-domain representation, which represents, for
example, amplitudes or energies and phases of a watermarked signal in different frequency
bands for a plurality of time intervals. It has been found that differences between adjacent
(e.g. temporally adjacent or frequency-adjacent) values of the first time-frequency-domain
representation, which for example can be derived from the watermarked audio signal using
a filter bank or using a Fourier transform or a MDCT transform, are typically robust with
respect to many typical channel distortions, like sufficiently slow changes of the channel, a
Doppler frequency shift, and so on. Accordingly, the second time-frequency-domain
representation can be obtained in a reliable manner, and the second time-frequency-domain
representation is therefore insensitive with respect to chances of the channel, via which the
watermarked signal is transmitted. Accordingly, the above-described watermark decoder
provides for a very high degree of reliability.
I a preferred embodiment, the time-frequency-domain provider is configured to provide,
for a plurality of frequency bands and for a plurality of time intervals, soft bit coefficients
describing an amplitude and a phase of the watermarked signal in the respective frequency
bands and time intervals. The differential decoder is configured to determine a value of the
second time-frequency-domain representation associated with a given frequency band and
a given time interval on the basis of two corresponding values of the first time-frequencydomain
representation, or a pre-processed version thereof. Using two values of the first
time-frequency-domain representation in order to obtain one value of the second timefrequency-
domain representation, it is possible to evaluate the phase differences between
the two values of the first time-frequency-domain representation. The processing may be
done on the basis of real values and/or complex values. Accordingly, any slow changes of
the channel, which do not have a strongly different impact onto adjacent values of the first
time-frequency-domain representation, can be approximately compensated by using two
values of the first time-frequency-domain representation in order to obtain values of the
second time-frequency-domain representation.
In a preferred embodiment, the watermark decoder comprises an analysis filterbank
configured to convolve the watermarked signal, or a downmixed version thereof, with a bit
forming function. In this case, the watermark decoder is configured to time-sample a result
of the convolution, in order to obtain time-discrete values of the first time-frequencydomain
representation. The watermark decoder is configured to adjust a timing used for a
sampling of the result of the convolution at a sub-bit-interval resolution in dependence on a
synchronization information, in order to maximize the signal-to-noise ratio and to
minimize a symbol interference ratio. It has been found that the output of such an analysis
filterbank is well-suited to serve as the first time-frequency-domain representation for the
differential decoding. Also, it has been found that the differential decoding provides
reasonable results for the first time-frequency-domain representation, even if there is a
slight misalignment of the timing used for sampling the result of the convolution.
In a preferred embodiment, the differential decoder is configured to derive the second
time-frequency-domain representation independently for different frequency bands, such
that different phase rotations of the watermarked signal in different frequency bands are
compensated independently. The synchronization determinator or the watermark decoder is
configured to jointly process a set of values of the second time-frequency-domain
representation associated with a given time portion and different frequency bands, to
obtain a synchronization information or a bit of the binary message data. It has been found
that differential decoding allows for a reliable joint processing of values of the second
time-frequency-domain representation even without using a channel corrector, and even
without knowledge about a channel state. Accordingly, the inventive concept allows for a
particularly efficient implementation.
An embodiment according to the invention creates a portable watermark evaluation device.
The watermark evaluation device comprises a microphone configured to provide an
electrical microphone signal and a watermark decoder, as discussed above. The watermark
decoder is configured to receive the microphone signal as the watermarked signal. It has
been found that the inventive watermark decoder can be applied with particular advantage
in such a portable watermark evaluation device evaluating an audio signal received by a
microphone, because the watermark decoder is particularly insensitive to typical channel
distortions, like, for example, Doppler shifts, transfer function nulls, and so on.
Further embodiments according to the invention create a method for providing a
watermark signal in dependence on binary message data and a method for providing binary
message data in dependence on a watermarked signal. Some further embodiments create
computer programs for performing said methods. The methods and computer programs are
based on the same findings as the above described apparatus.
Brief Description of the Figures
Embodiments according to the invention will subsequently be described taking reference to
the enclosed figures, in which:
Fig. 1 shows a block schematic diagram of a watermark inserter according to an
embodiment of the invention;
Fig. 2 shows a block-schematic diagram of a watermark decoder, according to an
embodiment of the invention;
Fig. 3 shows a detailed block-schematic diagram of a watermark generator,
according to an embodiment of the invention;
Fig. 4 shows a detailed block-schematic diagram of a modulator, for use in an
embodiment of the invention;
Fig. 5 shows a detailed block-schematic diagram of a psychoacoustical processing
module, for use in an embodiment of the invention;
Fig. 6 shows a block-schematic diagram of a psychoacoustical model processor,
for use in an embodiment of the invention;
Fig. 7 shows a graphical representation of a power spectrum of an audio signal
output by block 801 over frequency;
Fig. 8 shows a graphical representation of a power spectrum of an audio signal
output by block 802 over frequency;
shows a block-schematic diagram of an amplitude calculation;
shows a block schematic diagram of a modulator;
shows a graphical representation of the location of coefficients on the timefrequency
claim;
and 1l b show a block-schematic diagrams of implementation alternatives of
the synchronization module;
shows a graphical representation of the problem of finding the temporal
alignment of a watermark;
shows a graphical representation of the problem of identifying the message
start;
shows a graphical representation of a temporal alignment of synchronization
sequences in a full message synchronization mode;
shows a graphical representation of the temporal alignment of the
synchronization sequences in a partial message synchronization mode;
shows a graphical representation of input data of the synchronization
module;
shows a graphical representation of a concept of identifying a
synchronization hit;
shows a block-schematic diagram of a synchronization signature correlator;
shows a graphical representation of an example for a temporal despreading;
shows a graphical representation of an example for an element-wise
multiplication between bits and spreading sequences;
shows a graphical representation of an output of the synchronization
signature correlator after temporal averaging;
shows a graphical representation of an output of the synchronization
signature correlator filtered with the auto-correlation function of the
synchronization signature;
shows a block-schematic diagram of a watermark extractor, according to an
embodiment of the invention;
shows a schematic representation of a selection of a part of the timefrequency-
domain representation as a candidate message;
shows a block-schematic diagram of an analysis module;
shows a graphical representation of an output of a synchronization
correlator;
shows a graphical representation of decoded messages;
shows a graphical representation of a synchronization position, which is
extracted from a watermarked signal;
shows a graphical representation of a payload, a payload with a Viterbi
termination sequence, a Viterbi-encoded payload and a repetition-coded
version of the Viterbi-coded payload;
shows a graphical representation of subcarriers used for embedding a
watermarked signal;
shows a graphical representation of an uncoded message, a coded message,
a synchronization message and a watermark signal, in which the
synchronization sequence is applied to the messages;
shows a schematic representation of a first step of a so-called "ABC
synchronization" concept;
shows a graphical representation of a second step of the so-called "ABC
synchronization" concept;
Fig. 22 shows a graphical representation of a third step of the so-called "ABC
synchronization" concept;
Fig. 23 shows a graphical representation of a message comprising a payload and a
CRC portion;
Fig. 24 shows a block-schematic diagram of a watermark generator, according to an
embodiment of the invention;
Fig. 25 shows a block-schematic diagram of a watermark decoder, according to an
embodiment of the invention;
Fig. 26 shows a block-schematic diagram of a watermark decoder, according to an
embodiment of the invention;
Fig. 27 shows a block-schematic diagram of a portable watermark evaluation device,
according to an embodiment of the invention;
Fig. 28 shows a flowchart of a method for providing a watermarked signal in
dependence on binary message data; and
Fig. 29 shows a flowchart of a method for providing binary message data in
dependence on a watermarked signal.
Detailed Description of the Embodiments
1. Watermark generation
1.1 Watermark generator according to Fig. 24
In the following, a watermark generator 2400 will be described taking reference to 24,
which shows the block-schematic diagram of such watermark generator. The watermark
generator 2400 is configured to receive binary message data 2410 and to provide, on the
basis thereof, a watermarked signal 2420. The watermark generator 2400 comprises an
information processor 2430, which is configured to provide, in dependence on the
information units (e.g. bits) of the binary message data 2410, a first time-frequencydomain
representation 2432, values of which represent the binary message data 2410. The
watermark generator 2400 also comprises a differential encoder 2440, which is configured
to derive a second time-frequency-domain representation 2442 from the first timefrequency-
domain representation 2432, such that the second time-frequency-domain
representation 2442 comprises a plurality of values, wherein a difference between two
values of the second time-frequency-domain representation 2442 represents a
corresponding value of the first time-frequency-domain representation 2432, in order to
obtain a differential encoding of the values of the first time-frequency-domain
representation 2432. The watermark generator 2400 also comprises watermarked signal
provider 2450, which is configured to provide the watermarked signal 2420 on the basis of
the second time-frequency-domain representation 2442.
The watermark generator 2400 may be supplemented by any of the features and
functionalities which are discussed in more detail in section 3 below.
1.2. Method for providing a watermarked signal in dependence on binary message data
according to Fig. 28.
In the following, a method for providing a watermarked signal in dependence on binary
message data will be explained taking reference to Fig. 28, which shows a flowchart of
such method. The method 2800 of Fig. 28 comprises a step 2810 of providing, in
dependence on the information units of the binary message data, a first time-frequencydomain
representation, values of which represent the binary message data. The method
2800 also comprises a step 2820 of deriving a second time-frequency-domain
representation from the first time-frequency-domain representation, such that the second
time-frequency-domain representation comprises a plurality of values, wherein a difference
between two values of the second time-frequency-domain representation represents a
corresponding value of the first time-frequency-domain representation, in order to obtain a
differential encoding of the values of the first time-frequency-domain representation. The
method 2800 also comprises a step 2830 of providing the watermarked signal on the basis
of the second time-frequency-domain representation.
Naturally, the method 2800 can be supplemented by any of the features and functionalities
discussed herein, also with respect to the inventive apparatus.
2. Watermark decoding
2.1. Watermark decoder according to Fig. 2
In the following, a watermark decoder 2500 will be described taking reference to Fig. 25,
which shows a block-schematic diagram of such a watermark decoder.
The watermark decoder 2500 is configured to provide binary message data 2520 in
dependence on a watermarked signal 2510. The watermark decoder 2500 comprises a
time-frequency-domain representation provider 2530, which is configured to provide a first
time-frequency-domain representation 2532 of the watermarked signal 2510. The
watermark decoder 2500 also comprises a differential decoder 2540, which is configured
to derive a second time-frequency-domain representation 2542 from the first timefrequency-
domain representation 2532, such that values of the second time-frequencydomain
representation 2542 are dependent on phase differences between two
corresponding (and preferably adjacent) values of the first time-frequency-domain
representation 2532. The watermark decoder 2500 also comprises a synchronization
determinator 2550, which is configured to obtain a synchronization information 2552 on
the basis of the second time-frequency-domain representation 2542. The watermark
decoder 2500 also comprises a watermark extractor 2560, configured to extract the binary
message data 2520 from the first time-frequency-domain representation 2532 of the
watermarked signal 2510 or from the second time-frequency-domain representation 2542
of the watermarked signal 25 10 using the synchronization information 2552.
Naturally, the watermark decoder 2500 may be supplemented by any of the features and
functionalities discussed here with respect to the watermark decoding.
2.2. Watermark decoder according to Fig. 26
In the following, a watermark decoder 2600 will be described taking reference to Fig. 26,
which shows a block-schematic diagram of such a watermark decoder. The watermark
decoder 2600 is configured to receive a watermarked signal 2610 and to provide on the
basis thereof binary message data 2620. The watermark decoder 2600 comprises a timefrequency-
domain representation provider 2630 configured to provide a first timefrequency-
domain representation 2632 of the watermarked signal 2610. The watermark
decoder 2600 also comprises a differential decoder 2640 configured to derive a second
time-frequency-domain representation 2642 from the first time-frequency-domain
representation 2632, such that values of the second time-frequency-domain representation
are dependent on phase differences between two corresponding (and preferably temporally
adjacent or frequency-adjacent) values of the first time-frequency-domain representation
2632. The watermark decoder 2600 also comprises a watermark extractor 2650, which is
configured to extract the binary message data 2620 from the second time-frequencydomain
representation 2642.
Naturally, the watermark decoder 2600 may be supplemented by any of the means and
functionalities discussed herein with respect to watermark decoding.
2.3. Watermark evaluation device according to Fig. 27
In the following, a portable watermark evaluation device will be described, taking
reference to Fig. 27, which shows a block-schematic diagram of such a device 2700.
The portable watermark evaluation device 2700 comprises a microphone 2710 configured
to provide an electrical microphone signal 2712. The portable watermark evaluation device
2700 also comprises a watermark decoder 2720, which may be identical to the watermark
decoders described herein. The watermark decoder 2720 is configured to receive the
microphone signal 2712 as a watermarked signal, to provide binary message data 2722 on
the basis thereof.
Naturally, the watermark decoder 2720 may be supplemented by any of the means and
functionalities described herein with respect to the watermark decoding.
2.4, Method for providing binary message data in dependence on a watermarked signal
according to Fig. 29.
In the following, a method 2900 for providing binary message data in dependence on a
watermarked signal will be described taking reference to Fig. 29, which shows a flowchart
of such a method.
The method 2900 comprises a step 2910 of providing a first time-frequency-domain
representation of the watermarked signal. The method 2900 also comprises a step 2920 of
deriving a second time-frequency-domain representation from the first time-frequencydomain
representation, such that values of the second time-frequency-domain
representation are dependent on phase differences between two corresponding (and
preferably adjacent) values of the first time-frequency-domain representation.
The method 2900 also comprises a step 2930 of using the second time-frequency-domain
representation to determine a synchronization information, which is used for providing the
binary message data or extracting the binary message data from the watermarked signal.
The method 2900 can be supplemented by any of the features and functionalities described
here with respect to watermark decoding.
3. System Description
In the following, a system for a watermark transmission will be described, which
comprises a watermark inserter and a watermark decoder. Naturally, the watermark
inserter and the watermark decoder can be used independent from each other.
For the description of the system a top-down approach is chosen here. First, it is
distinguished between encoder and decoder. Then, in sections 3.1 to 3.5 each processing
block is described in detail.
The basic structure of the system can be seen in Figures 1 and 2, which depict the encoder
and decoder side, respectively. Fig 1 shows a block schematic diagram of a watermark
inserter 100. At the encoder side, the watermark signal 101b is generated in the processing
block 101 (also designated as watermark generator) from binary data 101a and on the basis
of information 104, 105 exchanged with the psychoacoustical processing module 102. The
information provided from block 102 typically guarantees that the watermark is inaudible.
The watermark generated by the watermark generatorlOl is then added to the audio signal
106. The watermarked signal 107 can then be transmitted, stored, or further processed. In
case of a multimedia file, e.g., an audio-video file, a proper delay needs to be added to the
video stream not to lose audio-video synchronicity. In case of a multichannel audio signal,
each channel is processed separately as explained in this document. The processing blocks
101 (watermark generator) and 102 (psychoacoustical processing module) are explained in
detail in Sections 3.1 and 3.2, respectively.
The decoder side is depicted in Figure 2, which shows a block schematic diagram of a
watermark detector 200. A watermarked audio signal 200a, e.g., recorded by a
microphone, is made available to the system 200. A first block 203, which is also
designated as an analysis module, demodulates and transforms the data (e.g., the
watermarked audio signal) in time/frequency domain (thereby obtaining a time-frequencydomain
representation 204 of the watermarked audio signal 200a) passing it to the
synchronization module 201, which analyzes the input signal 204 and carries out a
temporal synchronization, namely, determines the temporal alignment of the encoded data
(e.g. of the encoded watermark data relative to the time-frequency-domain representation).
This information (e.g., the resulting synchronization information 205) is given to the
watermark extractor 202, which decodes the data (and consequently provides the binary
data 202a, which represent the data content of the watermarked audio signal 200a).
3.1 The Watermark Generator 101
The watermark generator 101 is depicted detail in Figure 3. Binary data (expressed as ±1)
to be hidden in the audio signal 106 is given to the watermark generator 101. The block
301 organizes the data 101a in packets of equal length M . Overhead bits are added (e.g.
appended) for signaling purposes to each packet. Let Ms denote their number. Their use
will be explained in detail in Section 3.5. Note that in the following each packet of payload
bits together with the signaling overhead bits is denoted message.
Each message 301a, of length Nm = Ms + Mp, is handed over to the processing block 302,
the channel encoder, which is responsible of coding the bits for protection against errors. A
possible embodiment of this module consists of a convolutional encoder together with an
interleaver. The ratio of the convolutional encoder influences greatly the overall degree of
protection against errors of the watermarking system. The interleaver, on the other hand,
brings protection against noise bursts. The range of operation of the interleaver can be
limited to one message but it could also be extended to more messages. Let R denote the
code ratio, e.g., 1/4. The number of coded bits for each message is Nm R . The channel
encoder provides, for example, an encoded binary message 302a.
The next processing block, 303, carries out a spreading in frequency domain. In order to
achieve sufficient signal to noise ratio, the information (e.g. the information of the binary
message 302a) is spread and transmitted in Nf carefully chosen subbands. Their exact
position in frequency is decided a priori and is known to both the encoder and the decoder.
Details on the choice of this important system parameter is given in Section 3.2.2. The
spreading in frequency is determined by the spreading sequence Cf of size N XI. The
output 303a of the block 303 consists of Nf bit streams, one for each subband. The i-th bit
stream is obtained by multiplying the input bit with the i-th component of spreading
sequence Cf. The simplest spreading consists of copying the bit stream to each output
stream, namely use a spreading sequence of all ones.
Block 304, which is also designated as a synchronization scheme inserter, adds a
synchronization signal to the bit stream. A robust synchronization is important as the
decoder does not know the temporal alignment of neither bits nor the data structure, i.e.,
when each message starts. The synchronization signal consists of Ns sequences of Nf bits
each. The sequences are multiplied element wise and periodically to the bit stream (or bit
streams 303a). For instance, let a, b, and c, be the Ns = 3 synchronization sequences (also
designated as synchronization spreading sequences). Block 304 multiplies a to the first
spread bit, b to the second spread bit, and c to the third spread bit. For the following bits
the process is periodically iterated, namely, a to the fourth bit, b for the fifth bit and so on.
Accordingly, a combined information-synchronization information 304a is obtained. The
synchronization sequences (also designated as synchronization spread sequences) are
carefully chosen to minimize the risk of a false synchronization. More details are given in
Section 3.4. Also, it should be noted that a sequence a, b, c,... may be considered as a
sequence of synchronization spread sequences.
Block 305 carries out a spreading in time domain. Each spread bit at the input, namely a
vector of length Nf , is repeated in time domain Nt times. Similarly to the spreading in
frequency, we define a spreading sequence ct of size NtXl. The i-th temporal repetition is
multiplied with the i-th component of ct.
The operations of blocks 302 to 305 can be put in mathematical terms as follows. Let mof
size 1 Nm= be a coded message, output of 302. The output 303a (which may be
considered as a spread information representation R) of block 303 i' s
C · of size Nf x Nm / R
(1)
the output 304a of block 304, which may be considered as a combined informationsynchronization
representation C, is
S o (cf · ) of size Nf x N / Rc
(2)
where ° denotes the Schur element-wise product and
= [ . . . a b c . . . a b . . . ] of size N f x N / Rc .
(3)
The output 305a of 305 is
{S o (cf · )) «CtT of size N x N · N / Rc
(4)
where and denote the Kronecker product and transpose, respectively. Please recall that
binary data is expressed as ±1.
Block 306 performs a differential encoding of the bits. This step gives the system
additional robustness against phase shifts due to movement or local oscillator mismatches.
More details on this matter are given in Section 3.3. If b(i; j ) is the bit for the i-th
frequency band and j-th time block at the input of block 306, the output bit bdiff (i; j ) is
fediff ( ) = is i.j - 1) · {i j).
(5)
At the beginning of the stream, that is for j = 0, b if (ij - 1) is set to 1.
Block 307 carries out the actual modulation, i.e., the generation of the watermark signal
waveform depending on the binary information 306a given at its input. A more detailed
schematics is given in Figure 4. Nf parallel inputs, 401 to 40Nf contain the bit streams for
the different subbands. Each bit of each subband stream is processed by a bit shaping block
(41 1 to 41N ) . The output of the bit shaping blocks are waveforms in time domain. The
waveform generated for the j-th time block and i-th subband, denoted by Sjy (t), on the basis
of the input bit bdif (i, j )
is computed as follows
Sij{t) = b i ( j ) {i ) 9i(t - j Tb ) ,
(6)
where g ( ; j ) is a weighting factor provided by the psychoacoustical processing unit 102, Tb
is the bit time interval, and gj(t) is the bit forming function for the i-th subband. The bit
forming function is obtained from a baseband function · *) modulated in frequency
with a cosine
9i t ) = g t ) - oa 2 fit)
(7)
where f is the center frequency of the i-th subband and the superscript T stands for
transmitter. The baseband functions can be different for each subband. If chosen identical,
a more efficient implementation at the decoder is possible. See Section 3.3 for more
details.
The bit shaping for each bit is repeated in an iterative process controlled by the
psychoacoustical processing module (102). Iterations are necessary to fine tune the weights
g( , j ) to assign as much energy as possible to the watermark while keeping it inaudible.
More details are given in Section 3.2.
The complete waveform at the output of the i-th bit shaping fillter 4 1i is
= å ¾(*)
j
(8)
The bit forming baseband function ( *) is normally non zero for a time interval much
larger than T , although the main energy is concentrated within the bit interval. An
example can be seen if Figure 12a where the same bit forming baseband function is plotted
for two adjacent bits. In the figure we have T = 40 ms. The choice of T as well as the
shape of the function affect the system considerably. In fact, longer symbols provide
narrower frequency responses. This is particularly beneficial in reverberant environments.
In fact, in such scenarios the watermarked signal reaches the microphone via several
propagation paths, each characterized by a different propagation time. The resulting
channel exhibits strong frequency selectivity. Interpreted in time domain, longer symbols
are beneficial as echoes with a delay comparable to the bit interval yield constructive
interference, meaning that they increase the received signal energy. Notwithstanding,
longer symbols bring also a few drawbacks; larger overlaps might lead to intersymbol
interference (ISI) and are for sure more difficult to hide in the audio signal, so that the
psychoacoustical processing module would allow less energy than for shorter symbols.
The watermark signal is obtained by summing all outputs of the bit shaping filters
i
(9)
3.2 The Psychoacoustical Processing Module 102
As depicted in Figure 5, the psychoacoustical processing module 102 consists of 3 parts.
The first step is an analysis module 501 which transforms the time audio signal into the
time/frequency domain. This analysis module may carry out parallel analyses in different
time/frequency resolutions. After the analysis module, the time/frequency data is
transferred to the psychoacoustic model (PAM) 502, in which masking thresholds for the
watermark signal are calculated according to psychoacoustical considerations (see E.
Zwicker H.Fastl, "Psychoacoustics Facts and models"). The masking thresholds indicate
the amount of energy which can be hidden in the audio signal for each subband and time
block. The last block in the psychoacoustical processing module 102 depicts the amplitude
calculation module 503. This module determines the amplitude gains to be used in the
generation of the watermark signal so that the masking thresholds are satisfied, i.e., the
embedded energy is less or equal to the energy defined by the masking thresholds.
3.2.1 The Time/Frequency Analysis 501
Block 501 carries out the time/frequency transformation of the audio signal by means of a
lapped transform. The best audio quality can be achieved when multiple time/frequency
resolutions are performed. One efficient embodiment of a lapped transform is the short
time Fourier transform (STFT), which is based on fast Fourier transforms (FFT) of
windowed time blocks. The length of the window determines the time/frequency
resolution, so that longer windows yield lower time and higher frequency resolutions,
while shorter windows vice versa. The shape of the window, on the other hand, among
other things, determines the frequency leakage.
For the proposed system, we achieve an inaudible watermark by analyzing the data with
two different resolutions. A first filter bank is characterized by a hop size of Tb, i.e., the bit
length. The hop size is the time interval between two adjacent time blocks. The window
length is approximately Tb. Please note that the window shape does not have to be the
same as the one used for the bit shaping, and in general should model the human hearing
system. Numerous publications study this problem.
The second filter bank applies a shorter window. The higher temporal resolution achieved
is particularly important when embedding a watermark in speech, as its temporal structure
is in general finer than Tb.
The sampling rate of the input audio signal is not important, as long as it is large enough to
describe the watermark signal without aliasing. For instance, if the largest frequency
component contained in the watermark signal is 6 kHz, then the sampling rate of the time
signals must be at least 12 kHz.
3.2.2 The Psychoacoustical Model 502
The psychoacoustical model 502 has the task to determine the masking thresholds, i.e., the
amount of energy which can be hidden in the audio signal for each subband and time block
keeping the watermarked audio signal indistinguishable from the original.
The i-th subband is defined between two limits, namely / » m and / i x The subbands are
determined by defining N center frequencies f and letting / - = / im ,:)i for i - 2, 3, ... ,
N . An appropriate choice for the center frequencies is given by the Bark scale proposed
by Zwicker in 1961. The subbands become larger for higher center frequencies. A possible
implementation of the system uses 9 subbands ranging from 1.5 to 6 kHz arranged in an
appropriate way.
The following processing steps are carried out separately for each time/frequency
resolution for each subband and each time block. The processing step 801 carries out a
spectral smoothing. In fact, tonal elements, as well as notches in the power spectrum need
to be smoothed. This can be carried out in several ways. A tonality measure may be
computed and then used to drive an adaptive smoothing filter. Alternatively, in a simpler
implementation of this block, a median-like filter can be used. The median filter considers
a vector of values and outputs their median value. In a median-like filter the value
corresponding to a different quantile than 50% can be chosen. The filter width is defined in
Hz and is applied as a non-linear moving average which starts at the lower frequencies and
ends up at the highest possible frequency. The operation of 801 is illustrated in Figure 7.
The red curve is the output of the smoothing.
Once the smoothing has been carried out, the thresholds are computed by block 802
considering only frequency masking. Also in this case there are different possibilities. One
way is to use the minimum for each subband to compute the masking energy Ej. This is the
equivalent energy of the signal which effectively operates a masking. From this value we
can simply multiply a certain scaling factor to obtain the masked energy Jj. These factors
are different for each subband and time/frequency resolution and are obtained via empirical
psychoacoustical experiments. These steps are illustrated in Figure 8.
In block 805, temporal masking is considered. In this case, different time blocks for the
same subband are analyzed. The masked energies J are modified according to an
empirically derived postmasking profile. Let us consider two adjacent time blocks, namely
k-1 and k. The corresponding masked energies are Ji(k-l) and J (k). The postmasking
profile defines that, e.g., the masking energy Ej can mask an energy J at time k and j at
time k+1. In this case, block 805 compares J (k) (the energy masked by the current time
block) and a-J,(k+l) (the energy masked by the previous time block) and chooses the
maximum. Postmasking profiles are available in the literature and have been obtained via
empirical psychoacoustical experiments. Note that for large Tb, i.e., > 20 ms, postmasking
is applied only to the time/frequency resolution with shorter time windows.
Summarizing, at the output of block 805 we have the masking thresholds per each subband
and time block obtained for two different time/frequency resolutions. The thresholds have
been obtained by considering both frequency and time masking phenomena. In block 806,
the thresholds for the different time/frequency resolutions are merged. For instance, a
possible implementation is that 806 considers all thresholds corresponding to the time and
frequency intervals in which a bit is allocated, and chooses the minimum.
3.2.3 The Amplitude Calculation Block 503
Please refer to Figure 9. The input of 503 are the thresholds 505 from the psychoacoustical
model 502 where all psychoacoustics motivated calculations are carried out. In the
amplitude calculator 503 additional computations with the thresholds are performed. First,
an amplitude mapping 901 takes place. This block merely converts the masking thresholds
(normally expressed as energies) into amplitudes which can be used to scale the bit shaping
function defined in Section 3.1. Afterwards, the amplitude adaptation block 902 is run.
This block iteratively adapts the amplitudes g ( , j) which are used to multiply the bit
shaping functions in the watermark generator 101 so that the masking thresholds are
indeed fulfilled. In fact, as already discussed, the bit shaping function normally extends for
a time interval larger than Tb. Therefore, multiplying the correct amplitude g ( , j ) which
fulfills the masking threshold at point i, j does not necessarily fulfill the requirements at
point i, j-1. This is particularly crucial at strong onsets, as a preecho becomes audible.
Another situation which needs to be avoided is the unfortunate superposition of the tails of
different bits which might lead to an audible watermark. Therefore, block 902 analyzes the
signal generated by the watermark generator to check whether the thresholds have been
fulfilled. If not, it modifies the amplitudes y(i, j ) accordingly.
This concludes the encoder side. The following sections deal with the processing steps
carried out at the receiver (also designated as watermark decoder).
3.3 The Analysis Module 203
The analysis module 203 is the first step (or block) of the watermark extraction process. Its
purpose is to transform the watermarked audio signal 200a back into Nf bit streams b j )
(also designated with 204), one for each spectral subband i. These are further processed by
the synchronization module 201 and the watermark extractor 202, as discussed in Sections
3.4 and 3.5, respectively. Note that the b are soft bit streams, i.e., they can take, for
example, any real value and no hard decision on the bit is made yet.
The analysis module consists of three parts which are depicted in Figure 16: The analysis
filter bank 1600, the amplitude normalization block 1604 and the differential decoding
1608.
3.3.1 Analysis filter bank 1600
The watermarked audio signal is transformed into the time-frequency domain by the
analysis filter bank 1600 which is shown in detail in Figure 10a. The input of the filter
bank is the received watermarked audio signal r(t). Its output are the complex coefficients
b {j) for the i-th branch or subband at time instant j . These values contain information
about the amplitude and the phase of the signal at center frequency f j and time j-Tb.
The filter bank 1600 consists of Nf branches, one for each spectral subband i. Each branch
splits up into an upper subbranch for the in-phase component and a lower subbranch for
the quadrature component of the subband i. Although the modulation at the watermark
generator and thus the watermarked audio signal are purely real-valued, the complexvalued
analysis of the signal at the receiver is needed because rotations of the modulation
constellation introduced by the channel and by synchronization misalignments are not
known at the receiver. In the following we consider the i-th branch of the filter bank. By
combining the in-phase and the quadrature subbranch, we can define the complex- valued
baseband signal as
b (t) = r t) e - j fit *g t)
(10)
where * indicates convolution and ) is the impulse response of the receiver lowpass
filter of subband i. Usually fl (*)i (t) is equal to the baseband bit forming function
subband i in the modulator 307 in order to fulfill the matched filter condition, but other
impulse responses are possible as well.
In order to obtain the coefficients & J ) with rate l=T b, the continuous output i' ( )
must be sampled. If the correct timing of the bits was known by the receiver, sampling
with rate l=T would be sufficient. However, as the bit synchronization is not known yet,
sampling is carried out with rate Nos/Tb where Nos is the analysis filter bank oversampling
factor. By choosing Nos sufficiently large (e.g. Nos = 4), we can assure that at least one
sampling cycle is close enough to the ideal bit synchronization. The decision on the best
oversampling layer is made during the synchronization process, so all the oversampled
data is kept until then. This process is described in detail in Section 3.4.
At the output of the i-th branch we have the coefficients b F - k where j indicates the bit
number or time instant and k indicates the oversampling position within this single bit,
where k = 1; 2; . ..., Nos .
Figure 10b gives an exemplary overview of the location of the coefficients on the timefrequency
plane. The oversampling factor is Nos = 2. The height and the width of the
rectangles indicate respectively the bandwidth and the time interval of the part of the signal
that is represented by the corresponding coefficient B
If the subband frequencies f are chosen as multiples of a certain interval Af the analysis
filter bank can be efficiently implemented using the Fast Fourier Transform (FFT).
3.3.2 Amplitude normalization 1604
Without loss of generality and to simplify the description, we assume that the bit
synchronization is known and that Nos = 1 in the following. That is, we have complex
coeffcients ( . )at the input of the normalization block 1604. As no channel state
information is available at the receiver (i.e., the propagation channel in unknown), an equal
gain combining (EGC) scheme is used. Due to the time and frequency dispersive channel,
the energy of the sent bit bi(j) is not only found around the center frequency f, and time
instant j , but also at adjacent frequencies and time instants. Therefore, for a more precise
weighting, additional coefficients at frequencies f, ±n Af are calculated and used for
normalization of coefficient -If « = 1we have, for example,
( )
The normalization for n > 1 is a straightforward extension of the formula above. In the
same fashion we can also choose to normalize the soft bits by considering more than one
time instant. The normalization is carried out for each subband i and each time instant j .
The actual combining of the EGC is done at later steps of the extraction process.
Differential decoding 1608
At the input of the differential decoding block 1608 we have amplitude normalized
complex coefficients " ;' )which contain information about the phase of the signal
components at frequency f and time instant j . As the bits are differentially encoded at the
transmitter, the inverse operation must be performed here. The soft bits )are obtained
by first calculating the difference in phase of two consecutive coefficients and then taking
the real part:
bi = e{ ) · m ( - 1)}
= Re{|6 ) l · m - 1)1 · -
(13)
This has to be carried out separately for each subband because the channel normally
introduces different phase rotations in each subband.
The Synchronization Module 201
The synchronization module's task is to find the temporal alignment of the watermark. The
problem of synchronizing the decoder to the encoded data is twofold. In a first step, the
analysis filterbank must be aligned with the encoded data, namely the bit shaping functions
used in the synthesis in the modulator must be aligned with the filters - ) used
for the analysis. This problem is illustrated in Figure 12a, where the analysis filters are
identical to the synthesis ones. At the top, three bits are visible. For simplicity, the
waveforms for all three bits are not scaled. The temporal offset between different bits is T ,.
The bottom part illustrates the synchronization issue at the decoder: the filter can be
applied at different time instants, however, only the position marked in red (curve 1299a)
is correct and allows to extract the first bit with the best signal to noise ratio SNR and
signal to interference ratio SIR. In fact, an incorrect alignment would lead to a degradation
of both SNR and SIR. We refer to this first alignment issue as "bit synchronization". Once
the bit synchronization has been achieved, bits can be extracted optimally. However, to
correctly decode a message, it is necessary to know at which bit a new message starts. This
issue is illustrated in Figure 12b and is referred to as message synchronization. In the
stream of decoded bits only the starting position marked in red (position 1299b) is correct
and allows to decode the k-th message.
We first address the message synchronization only. The synchronization signature, as
explained in Section 3.1, is composed of Ns sequences in a predetermined order which are
embedded continuously and periodically in the watermark. The synchronization module is
capable of retrieving the temporal alignment of the synchronization sequences. Depending
on the size Ns we can distinguish between two modes of operation, which are depicted in
Figure 12c and 12d, respectively.
In the full message synchronization mode (Fig. 12c) we have Ns= Nm/Rc. For simplicity in
the figure we assume Ns = Nm/Rc = 6 and no time spreading, i.e., Nt = 1. The
synchronization signature used, for illustration purposes, is shown beneath the messages.
In reality, they are modulated depending on the coded bits and frequency spreading
sequences, as explained in Section 3.1. In this mode, the periodicity of the synchronization
signature is identical to the one of the messages. The synchronization module therefore can
identify the beginning of each message by finding the temporal alignment of the
synchronization signature. We refer to the temporal positions at which a new
synchronization signature starts as synchronization hits. The synchronization hits are then
passed to the watermark extractor 202.
The second possible mode, the partial message synchronization mode (Fig. 12d), is
depicted in Figure 12d. In this case we have Ns < In the figure we have taken Ns =
3, so that the three synchronization sequences are repeated twice for each message. Please
note that the periodicity of the messages does not have to be multiple of the periodicity of
the synchronization signature. In this mode of operation, not all synchronization hits
correspond to the beginning of a message. The synchronization module has no means of
distinguishing between hits and this task is given to the watermark extractor 202.
The processing blocks of the synchronization module are depicted in Figures 1l a and 1lb.
The synchronization module carries out the bit synchronization and the message
synchronization (either full or partial) at once by analyzing the output of the
synchronization signature correlator 1201. The data in time/frequency domain 204 is
provided by the analysis module. As the bit synchronization is not yet available, block 203
oversamples the data with factor Nos, as described in Section 3.3. An illustration of the
input data is given in Figure 12e. For this example we have taken Nos = 4, Nt = 2, and Ns =
3. In other words, the synchronization signature consists of 3 sequences (denoted with a, b,
and c). The time spreading, in this case with spreading sequence ct = [ 1 1] , simply repeats
each bit twice in time domain. The exact synchronization hits are denoted with arrows and
correspond to the beginning of each synchronization signature. The period of the
synchronization signature is Nt · Nos · Ns = NSbi which is 2 · 4 · 3 = 24, for example. Due to
the periodicity of the synchronization signature, the synchronization signature correlator
(1201) arbitrarily divides the time axis in blocks, called search blocks, of size NSbi, whose
subscript stands for search block length. Every search block must contain (or typically
contains) one synchronization hit as depicted in Figure 12f. Each of the NSb i bits is a
candidate synchronization hit. Block 1201's task is to compute a likelihood measure for
each of candidate bit of each block. This information is then passed to block 1204 which
computes the synchronization hits.
3.4.1 The synchronization signature correlator 1201
For each of the Nsbi candidate synchronization positions the synchronization signature
correlator computes a likelihood measure, the latter is larger the more probable it is that the
temporal alignment (both bit and partial or full message synchronization) has been found.
The processing steps are depicted in Figure 12g.
Accordingly, a sequence 1201aof likelihood values, associated with different positional
choices, may be obtained.
Block 1301 carries out the temporal despreading, i.e., multiplies every Nt bits with the
temporal spreading sequence ¾ and then sums them. This is carried out for each of the Nf
frequency subbands. Figure 13a shows an example. We take the same parameters as
described in the previous section, namely Nos = 4, Nt = 2, and Ns = 3. The candidate
synchronization position is marked. From that bit, with Nos offset, Nt · Ns are taken by
block 1301 and time despread with sequence ct, so that Ns bits are left.
In block 1302 the bits are multiplied element-wise with the Ns spreading sequences (see
Figure 13b).
In block 1303 the frequency despreading is carried out, namely, each bit is multiplied with
the spreading sequence Cf and then summed along frequency.
At this point, if the synchronization position were correct, we would have Ns decoded bits.
As the bits are not known to the receiver, block 1304 computes the likelihood measure by
taking the absolute values of the Ns values and sums.
The output of block 1304 is in principle a non coherent correlator which looks for the
synchronization signature. In fact, when choosing a small Ns, namely the partial message
synchronization mode, it is possible to use synchronization sequences (e.g. a, b, c) which
are mutually orthogonal. In doing so, when the correlator is not correctly aligned with the
signature, its output will be very small, ideally zero. When using the full message
synchronization mode it is advised to use as many orthogonal synchronization sequences
as possible, and then create a signature by carefully choosing the order in which they are
used. In this case, the same theory can be applied as when looking for spreading sequences
with good auto correlation functions. When the correlator is only slightly misaligned, then
the output of the correlator will not be zero even in the ideal case, but anyway will be
smaller compared to the perfect alignment, as the analysis filters cannot capture the signal
energy optimally.
3.4.2 Synchronization hits computation 1204
This block analyzes the output of the synchronization signature correlator to decide where
the synchronization positions are. Since the system is fairly robust against misalignments
of up to Tb 4 and the Tb is normally taken around 40 ms, it is possible to integrate the
output of 1201 over time to achieve a more stable synchronization. A possible
implementation of this is given by an IIR filter applied along time with a exponentially
decaying impulse response. Alternatively, a traditional FIR moving average filter can be
applied. Once the averaging has been carried out, a second correlation along different Nt-Ns
is carried out ("different positional choice"). In fact, we want to exploit the information
that the autocorrelation function of the synchronization function is known. This
corresponds to a Maximum Likelihood estimator. The idea is shown in Figure 13c. The
curve shows the output of block 1201 after temporal integration. One possibility to
determine the synchronization hit is simply to find the maximum of this function. In Figure
13d we see the same function (in black) filtered with the autocorrelation function of the
synchronization signature. The resulting function is plotted in red. In this case the
maximum is more pronounced and gives us the position of the synchronization hit. The
two methods are fairly similar for high SNR but the second method performs much better
in lower SNR regimes. Once the synchronization hits have been found, they are passed to
the watermark extractor 202 which decodes the data.
In some embodiments, in order to obtain a robust synchronization signal, synchronization
is performed in partial message synchronization mode with short synchronization
signatures. For this reason many decodings have to be done, increasing the risk of false
positive message detections. To prevent this, in some embodiments signaling sequences
may be inserted into the messages with a lower bit rate as a consequence.
This approach is a solution to the problem arising from a sync signature shorter than the
message, which is already addressed in the above discussion of the enhanced
synchronization. In this case, the decoder doesn't know where a new message starts and
attempts to decode at several synchronization points. To distinguish between legitimate
messages and false positives, in some embodiments a signaling word is used (i.e. payload
is sacrified to embed a known control sequence). In some embodiments, a plausibility
check is used (alternatively or in addition) to distinguish between legitimate messages and
false positives.
3.5 The Watermark Extractor 202
The parts constituting the watermark extractor 202 are depicted in Figure 14. This has two
inputs, namely 204 and 205 from blocks 203 and 201, respectively. The synchronization
module 201 (see Section 3.4) provides synchronization timestamps, i.e., the positions in
time domain at which a candidate message starts. More details on this matter are given in
Section 3.4. The analysis filterbank block 203, on the other hand, provides the data in
time/frequency domain ready to be decoded.
The first processing step, the data selection block 1501, selects from the input 204 the part
identified as a candidate message to be decoded. Figure 15b shows this procedure
graphically. The input 204 consists of Nf streams of real values. Since the time alignment is
not known to the decoder a priori, the analysis block 203 carries out a frequency analysis
with a rate higher than 1/Tb Hz (oversampling). In Figure 15b we have used an
oversampling factor of 4, namely, 4 vectors of size N 1 are output every T seconds.
When the synchronization block 201 identifies a candidate message, it delivers a
timestamp 205 indicating the starting point of a candidate message. The selection block
1501 selects the information required for the decoding, namely a matrix of size Nf xNm R .
This matrix 1501a is given to block 1502 for further processing.
Blocks 1502, 1503, and 1504 carry out the same operations of blocks 1301, 1302, and
1303 explained in Section 3.4.
An alternative embodiment of the invention consists in avoiding the computations done in
1502-1504 by letting the synchronization module deliver also the data to be decoded.
Conceptually it is a detail. From the implementation point of view, it is just a matter of
how the buffers are realized. In general, redoing the computations allows us to have
smaller buffers.
The channel decoder 1505 carries out the inverse operation of block 302. If channel
encoder, in a possible embodiment of this module, consisted of a convolutional encoder
together with an interleaver, then the channel decoder would perform the deinterleaving
and the convolutional decoding, e.g., with the well known Viterbi algorithm. At the output
of this block we have Nm bits, i.e., a candidate message.
Block 1506, the signaling and plausibility block, decides whether the input candidate
message is indeed a message or not. To do so, different strategies are possible.
The basic idea is to use a signaling word (like a CRC sequence) to distinguish between true
and false messages. This however reduces the number of bits available as payload.
Alternatively we can use plausibility checks. If the messages for instance contain a
timestamp, consecutive messages must have consecutive timestamps. If a decoded message
possesses a timestamp which is not the correct order, we can discard it.
When a message has been correctly detected the system may choose to apply the look
ahead and/or look back mechanisms. We assume that both bit and message
synchronization have been achieved. Assuming that the user is not zapping, the system
"looks back" in time and attempts to decode the past messages (if not decoded already)
using the same synchronization point (look back approach). This is particularly useful
when the system starts. Moreover, in bad conditions, it might take 2 messages to achieve
synchronization. In this case, the first message has no chance. With the look back option
we can save "good" messages which have not been received only due to back
synchronization. The look ahead is the same but works in the future. If we have a message
now we know where the next message should be, and we can attempt to decode it anyhow.
3.6. Synchronization Details
For the encoding of a payload, for example, a Viterbi algorithm may be used. Fig. 18a
shows a graphical representation of a payload 1810, a Viterbi termination sequence 1820, a
Viterbi encoded payload 1830 and a repetition-coded version 1840 of the Viterbi-coded
payload. For example, the payload length may be 34 bits and the Viterbi termination
sequence may comprise 6 bits. If, for example a Viterbi code rate of 1/7 may be used the
Viterbi-coded payload may comprise (34+6)*7=280 bits. Further, by using a repetition
coding of 1/2, the repetition coded version 1840 of the Viterbi-encoded payload 1830 may
comprise 280*2=560 bits. In this example, considering a bit time interval of 42.66 ms, the
message length would be 23.9 s. The signal may be embedded with, for example, 9
subcarriers (e.g. placed according to the critical bands) from 1.5 to 6 kHz as indicated by
the frequency spectrum shown in Fig. 18b. Alternatively, also another number of
subcarriers (e.g. 4, 6, 12, 15 or a number between 2 and 20) within a frequency range
between 0 and 20 kHz maybe used.
Fig. 19 shows a schematic illustration of the basic concept 1900 for the synchronization,
also called ABC synch. It shows a schematic illustration of an uncoded messages 1910, a
coded message 1920 and a synchronization sequence (synch sequence) 1930 as well as the
application of the synch to several messages 1920 following each other.
The synchronization sequence or synch sequence mentioned in connection with the
explanation of this synchronization concept (shown in Fig. 19 - 23) may be equal to the
synchronization signature mentioned before.
Further, Fig. 20 shows a schematic illustration of the synchronization found by correlating
with the synch sequence. If the synchronization sequence 1930 is shorter than the message,
more than one synchronization point 1940 (or alignment time block) may be found within
a single message. In the example shown in Fig. 20, 4 synchronization points are found
within each message. Therefore, for each synchronization found, a Viterbi decoder (a
Viterbi decoding sequence) may be started. In this way, for each synchronization point
1940 a message 2 11 may be obtained, as indicated in Fig. 21.
Based on these messages the true messages 2210 may be identified by means of a CRC
sequence (cyclic redundancy check sequence) and/or a plausibility check, as shown in Fig.
22.
The CRC detection (cyclic redundancy check detection) may use a known sequence to
identify true messages from false positive. Fig. 23 shows an example for a CRC sequence
added to the end of a payload.
The probability of false positive (a message generated based on a wrong synchronization
point) may depend on the length of the CRC sequence and the number of Viterbi decoders
(number of synchronization points within a single message) started. To increase the length
of the payload without increasing the probability of false positive a plausibility may be
exploited (plausibility test) or the length of the synchronization sequence (synchronization
signature) may be increased.
4. Concepts and Advantages
In the following, some aspects of the above discussed system will be described, which are
considered as being innovative. Also, the relation of those aspects to the state-of-the-art
technologies will be discussed.
4.1. Continuous synchronization
Some embodiments allow for a continuous synchronization. The synchronization signal,
which we denote as synchronization signature, is embedded continuously and parallel to
the data via multiplication with sequences (also designated as synchronization spread
sequences) known to both transmit and receive side.
Some conventional systems use special symbols (other than the ones used for the data),
while some embodiments according to the invention do not use such special symbols.
Other classical methods consist of embedding a known sequence of bits (preamble) timemultiplexed
with the data, or embedding a signal frequency-multiplexed with the data.
However, it has been found that using dedicated sub-bands for synchronization is
undesired, as the channel might have notches at those frequencies, making the
synchronization unreliable. Compared to the other methods, in which a preamble or a
special symbol is time-multiplexed with the data, the method described herein is more
advantageous as the method described herein allows to track changes in the
synchronization (due e.g. to movement) continuously.
Furthermore, the energy of the watermark signal is unchanged (e.g. by the multiplicative
introduction of the watermark into the spread information representation), and the
synchronization can be designed independent from the psychoacoustical model and data
rate. The length in time of the synchronization signature, which determines the robustness
of the synchronization, can be designed at will completely independent of the data rate.
Another classical method consists of embedding a synchronization sequence codemultiplexed
with the data. When compared to this classical method, the advantage of the
method described herein is that the energy of the data does not represent an interfering
factor in the computation of the correlation, bringing more robustness. Furthermore, when
using code-multiplexing, the number of orthogonal sequences available for the
synchronization is reduced as some are necessary for the data.
To summarize, the continuous synchronization approach described herein brings along a
large number of advantages over the conventional concepts.
However, in some embodiments according to the invention, a different synchronization
concept may be applied.
4.2. 2D spreading
Some embodiments of the proposed system carry out spreading in both time and frequency
domain, i.e. a 2-dimensional spreading (briefly designated as 2D-spreading). It has been
found that this is advantageous with respect to ID systems as the bit error rate can be
further reduced by adding redundance in e.g. time domain.
However, in some embodiments according to the invention, a different spreading concept
may be applied.
4.3. Differential encoding and Differential decoding
In some embodiments according to the invention, an increased robustness against
movement and frequency mismatch of the local oscillators (when compared to
conventional systems) is brought by the differential modulation. It has been found that in
fact, the Doppler effect (movement) and frequency mismatches lead to a rotation of the
BPSK constellation (in other words, a rotation on the complex plane of the bits). In some
embodiments, the detrimental effects of such a rotation of the BPSK constellation (or any
other appropriate modulation constellation) are avoided by using a differential encoding or
differential decoding.
However, in some embodiments according to the invention, a different encoding concept
or decoding concept may be applied. Also, in some cases, the differential encoding may be
omitted.
4.4. Bit shaping
In some embodiments according to the invention, bit shaping brings along a significant
improvement of the system performance, because the reliability of the detection can be
increased using a filter adapted to the bit shaping.
In accordance with some embodiments, the usage of bit shaping with respect to
watermarking brings along improved reliability of the watermarking process. It has been
found that particularly good results can be obtained if the bit shaping function is longer
than the bit interval.
However, in some embodiments according to the invention, a different bit shaping concept
may be applied. Also, in some cases, the bit shaping may be omitted.
4.5. Interactive between Psychoacoustic Model (PAM) and Filter Bank (FB) synthesis
In some embodiments, the psychoacoustical model interacts with the modulator to fine
tune the amplitudes which multiply the bits.
However, in some other embodiments, this interaction may be omitted.
4.6. Look ahead and look back features
In some embodiments, so called "Look back" and "look ahead" approaches are applied.
In the following, these concepts will be briefly summarized. When a message is correctly
decoded, it is assumed that synchronization has been achieved. Assuming that the user is
not zapping, in some embodiments a look back in time is performed and it is tried to
decode the past messages (if not decoded already) using the same synchronization point
(look back approach). This is particularly useful when the system starts.
In bad conditions, it might take 2 messages to achieve synchronization. In this case, the
first message has no chance in conventional systems. With the look back option, which is
used in some embodiments of the invention, it is possible to save (or decode) "good"
messages which have not been received only due to back synchronization.
The look ahead is the same but works in the future. If I have a message now I know where
my next message should be, and I can try to decode it anyhow. Accordingly, overlapping
messages can be decoded.
However, in some embodiments according to the invention, the look ahead feature and/or
the look back feature may be omitted.
4.7. Increased synchronization robustness
In some embodiments, in order to obtain a robust synchronization signal, synchronization
is performed in partial message synchronization mode with short synchronization
signatures. For this reason many decodings have to be done, increasing the risk of false
positive message detections. To prevent this, in some embodiments signaling sequences
may be inserted into the messages with a lower bit rate as a consequence.
However, in some embodiments according to the invention, a different concept for
improving the synchronization robustness may be applied. Also, in some cases, the usage
of any concepts for increasing the synchronization robustness may be omitted.
4.8. Other enhancements
In the following, some other general enhancements of the above described system with
respect to background art will be put forward and discussed:
. lower computational complexity
2. better audio quality due to the better psychoacoustical model
3. more robustness in reverberant environments due to the narrowband multicarrier
signals
4. an SNR estimation is avoided in some embodiments. This allows for better
robustness, especially in low SNR regimes.
Some embodiments according to the invention are better than conventional systems, which
use very narrow bandwidths of, for example, 8Hz for the following reasons:
1. 8 Hz bandwidths (or a similar very narrow bandwidth) requires very long time
symbols because the psychoacoustical model allows very little energy to make it inaudible;
2. 8 Hz (or a similar very narrow bandwidth) makes it sensitive against time varying
Doppler spectra. Accordingly, such a narrow band system is typically not good enough if
implemented, e.g., in a watch.
Some embodiments according to the invention are better than other technologies for the
following reasons:
1. Techniques which input an echo fail completely in reverberant rooms. In contrast,
in some embodiments of the invention, the introduction of an echo is avoided.
2. Techniques which use only time spreading have longer message duration in
comparison embodiments of the above described system in which a two-dimensional
spreading, for example both in time and in frequency, is used.
Some embodiments according to the invention are better than the system described in DE
196 40 814, because one of more of the following disadvantages of the system according to
said document are overcome:
• the complexity in the decoder according to DE 196 40 814 is very high, a filter of
length 2N with N = 128 is used
• the system according to DE 196 40 814 comprises a long message duration
• in the system according to DE 196 40 814 spreading only in time domain with
relatively high spreading gain (e.g. 128)
• in the system according to DE 196 40 814 the signal is generated in time domain,
transformed to spectral domain, weighted, transformed back to time domain, and
superposed to audio, which makes the system very complex
5. Applications
The invention comprises a method to modify an audio signal in order to hide digital data
and a corresponding decoder capable of retrieving this information while the perceived
quality of the modified audio signal remains indistinguishable to the one of the original.
Examples of possible applications of the invention are given in the following:
1. Broadcast monitoring: a watermark containing information on e.g. the station and
time is hidden in the audio signal of radio or television programs. Decoders, incorporated
in small devices worn by test subjects, are capable to retrieve the watermark, and thus
collect valuable information for advertisements agencies, namely who watched which
program and when.
2. Auditing: a watermark can be hidden in, e.g., advertisements. By automatically
monitoring the transmissions of a certain station it is then possible to know when exactly
the ad was broadcast. In a similar fashion it is possible to retrieve statistical information
about the programming schedules of different radios, for instance, how often a certain
music piece is played, etc.
3. Metadata embedding: the proposed method can be used to hide digital information
about the music piece or program, for instance the name and author of the piece or the
duration of the program etc.
6. Implementation Alternatives
Although some aspects have been described in the context of an apparatus, it is clear that
these aspects also represent a description of the corresponding method, where a block or
device corresponds to a method step or a feature of a method step. Analogously, aspects
described in the context of a method step also represent a description of a corresponding
block or item or feature of a corresponding apparatus. Some or all of the method steps may
be executed by (or using) a hardware apparatus, like for example, a microprocessor, a
programmable computer or an electronic circuit. In some embodiments, some one or more
of the most important method steps may be executed by such an apparatus.
The inventive encoded watermark signal, or an audio signal into which the watermark
signal is embedded, can be stored on a digital storage medium or can be transmitted on a
transmission medium such as a wireless transmission medium or a wired transmission
medium such as the Internet.
Depending on certain implementation requirements, embodiments of the invention can be
implemented in hardware or in software. The implementation can be performed using a
digital storage medium, for example a floppy disk, a DVD, a Blue-Ray, a CD, a ROM, a
PROM, an EPROM, an EEPROM or a FLASH memory, having electronically readable
control signals stored thereon, which cooperate (or are capable of cooperating) with a
programmable computer system such that the respective method is performed. Therefore,
the digital storage medium may be computer readable.
Some embodiments according to the invention comprise a data carrier having
electronically readable control signals, which are capable of cooperating with a
programmable computer system, such that one of the methods described herein is
performed.
Generally, embodiments of the present invention can be implemented as a computer
program product with a program code, the program code being operative for performing
one of the methods when the computer program product runs on a computer. The program
code may for example be stored on a machine readable carrier.
Other embodiments comprise the computer program for performing one of the methods
described herein, stored on a machine readable carrier.
In other words, an embodiment of the inventive method is, therefore, a computer program
having a program code for performing one of the methods described herein, when the
computer program runs on a computer.
A further embodiment of the inventive methods is, therefore, a data carrier (or a digital
storage medium, or a computer-readable medium) comprising, recorded thereon, the
computer program for performing one of the methods described herein.
A further embodiment of the inventive method is, therefore, a data stream or a sequence of
signals representing the computer program for performing one of the methods described
herein. The data stream or the sequence of signals may for example be configured to be
transferred via a data communication connection, for example via the Internet.
A further embodiment comprises a processing means, for example a computer, or a
programmable logic device, configured to or adapted to perform one of the methods
described herein.
A further embodiment comprises a computer having installed thereon the computer
program for performing one of the methods described herein.
In some embodiments, a programmable logic device (for example a field programmable
gate array) may be used to perform some or all of the functionalities of the methods
described herein. In some embodiments, a field programmable gate array may cooperate
with a microprocessor in order to perform one of the methods described herein. Generally,
the methods are preferably performed by any hardware apparatus.
The above described embodiments are merely illustrative for the principles of the present
invention. It is understood that modifications and variations of the arrangements and the
details described herein will be apparent to others skilled in the art. It is the intent,
therefore, to be limited only by the scope of the impending patent claims and not by the
specific details presented by way of description and explanation of the embodiments
herein.
Claims
A watermark generator (101; 2400) for providing a watermark signal (101b; 2420)
in dependence on binary message data (101a, m ; 2410), the watermark generator
comprising:
an information processor (303,304,305; 2430) configured to provide, in dependence
on information units of the binary message data, a first time-frequency domain
representation (b(i,j); 2432), values of which represent the binary message data; and
a differential encoder (306; 2440) configured to derive a second time-frequency
domain representation (306a, bdiff(i,j); 2442) from the first time-frequency-domain
representation, such that the second time-frequency-domain representation
comprises a plurality of values, wherein a difference between two values (bd,ff(i,j),
di (ij - )) of the second time-frequency-domain representation represents a
corresponding value of the first time-frequency-domain representation, in order to
obtain a differential encoding of the values of the first time-frequency-domain
representation; and
a watermark signal provider (307; 2450) configured to provide the watermark
signal on the basis of the second time-frequency-domain representation.
The watermark generator according to claim 1, wherein the information processor
is configured to provide the first time-frequency-domain representation such that
the values of the first time-frequency-domain representation represent the binary
message data in the form of a spread binary pattern; and
wherein the differential encoder is configured to derive the second time-frequency
domain representation such that a phase change between two subsequent values of
the second time-frequency domain representation is introduced, if a corresponding
value of the first time-frequency-domain representation takes a first value, and such
that there is no phase change between two subsequent values of the second timefrequency-
domain representation, if a corresponding value of the first timefrequency-
domain representation takes a second value, which is different from the
first value.
The watermark generator according to claim 2, wherein the information processor
is configured to provide a bit value b iff (i,j), associated with the i-th frequency band
and the j-th time block of the second time- frequency-domain representation such
that
bdiff(ij) = d f (ij-l)-b (i,j),
wherein bdi f (i,j-l) designates a bit value associated with the i-th frequency band
and the j-l-th time block of the second time-frequency-domain representation;
wherein b (i,j) designates a bit value associated with the i-th frequency band and the
j-th time block of the first time-frequency-domain representation; and
wherein binary states of the first time-frequency-domain representation are
represented by the values + 1 and - 1.
4. The watermark generator according to one of claims 1 to 3, wherein the watermark
signal provider is configured to provide a combined audio signal (107) on the basis
of the second time-frequency domain representation, such that a watermark
component of the watermark signal comprises a step-wise or smooth phase change
in response to a first value of the first time-frequency-domain representation, and
such that the watermark frequency component of the watermark signal comprises a
temporally constant phase in response to a second value of the first time-frequency
domain representation, which is different from the first value.
5. The watermark generator according to one of claims 1 to 4, wherein the watermark
signal provider is configured to provide a first bit-shaping waveform (gi (t)) in
response to a first value of the second time-frequency-domain representation, and to
provide a second bit-shaping waveform (-gi (t)) in response to a second value of the
second time-frequency-domain representation, and
wherein the watermark signal provider is configured to include into the watermark
signal a weighted or non-weighted superposition of time-shifted versions of the
same bit-shaping waveform in response to the presence of a first value in the first
time-frequency-domain representation, and to include into the watermark signal a
weighted or non-weighted superposition of time-shifted versions of the first bitshaping
waveform and of the second bit-shaping waveform in response to the
presence of a second value, which is different from the first value, in the first timefrequency
domain representation.
The watermark generator according to claim 5, wherein the second bit-shaping
waveform is an inverse of the first bit-shaping waveform.
A watermark decoder (200; 2500) for providing binary message data (202a; 2520)
in dependence on a watermarked signal (200a; 2510), the watermark decoder
comprising:
a time-frequency-domain representation provider (1600; 2530) configured to
provide a first time-frequency-domain representation (b o (j); 2532) of the
watermarked signal;
a differential decoder (1608; 2540) configured to derive a second time-frequencydomain
representation ( ) ; 2542) from the first time-frequency-domain
representation, such that values of the second time-frequency-domain
representation are dependent on phase differences between two corresponding
values (b no (j), b o (j-l) of the first time-frequency domain representation; and
a synchronization determinator (201; 2550) configured to obtain a synchronization
information (205; 2552) on the basis of the second time-frequency-domain
representation; and
a watermark extractor (202; 2560) configured to extract the binary message data
from the first time-frequency-domain representation of the watermarked signal or
from the second time-frequency-domain representation of the watermarked signal
using the synchronization information.
A watermark decoder (200; 2600)for providing binary message data (202a; 2620)
in dependence on a watermarked signal(200a; 2610), the watermark decoder
comprising:
a time-frequency-domain representation provider (1600; 2630) configured to
provide a first time-frequency-domain representation (bj or (j); 2632) of the
watermarked signal;
a differential decoder (1608; 2640) configured to derive a second time-frequencydomain
representation ( ,·( ) ; 2542) from the first time-frequency-domain
representation, such that values of the second time-frequency-domain
representation are dependent on phase differences between two corresponding
values (b o (j), b or (j-l) of the first time-frequency-domain representation; and
a watermark extractor (202; 2650) configured to extract the binary message data
from the second time-frequency-domain representation.
9. The watermark decoder according to claim 7 or claim 8, wherein the timefrequency-
domain representation provider is configured to provide, for a plurality
of frequency bands and for a plurality of time intervals, soft-bit coefficients
(b B(i), b o (j)) describing an amplitude and a phase of the watermarked signal in
the respective frequency bands and time intervals; and
wherein the differential decoder is configured to determine a value ( b ( ) ) of the
second time-frequency-domain representation associated with a given frequency
band (f) and a given time interval ( ) on the basis of two corresponding values
(b o (j) b o (j-l) of the first time-frequency-domain representation.
10. The watermark decoder according to one of claims 7 to 9, wherein the watermark
decoder comprises an analysis filter (1600) configured to convolve the
watermarked signal or a downmixed version thereof with a first bit-forming
function (gi (t)); and
wherein the watermark decoder is configured to time-sample a result of the
convolution in order to obtain time-discrete values of the first time-frequencydomain
representation; and
when the watermark decoder is configured to adjust a timing used for sampling the
result of the convolution at a sub-bit-interval resolution in dependence on a
synchronization information, in order to maximize a signal-to-noise ratio and to
minimize a symbol interference ratio.
11. The watermark decoder according to one of claims 7 to 10, wherein the differential
decoder is configured to derive the second time-frequency-domain representation
independently for different frequency bands (fj) , such that different phase rotations
of the watermarked signal in different frequency bands are compensated
independently by the differential decoder; and
wherein the synchronization determinator or the watermark detector is configured
to jointly process a set of values of the second time-frequency-domain
representation associated with a given time portion and different frequency bands,
to obtain a synchronization information or a bit of the binary message data.
1 . A portable watermark evaluation device (2700), comprising:
a microphone (2710) configured to provide an electrical microphone signal (2712);
and
a watermark decoder (2720) according to one of the claims 7 to 11, wherein the
watermark decoder is configured to receive a microphone signal as the
watermarked signal.
13. A method (2800) for providing a watermarked signal in dependence on binary
message data, the method comprising:
Providing (2810), in dependence on information units of the binary message data, a
first time-frequency-domain representation, values of which represent the binary
message data;
deriving (2820) a second time-frequency-domain representation from the first timefrequency
domain representation, such that the second time-frequency-domain
representation comprises a plurality of values, wherein a difference between two
values of the second time-frequency-domain representation represents a
corresponding value of the first time-frequency-domain representation, in order to
obtain a differential encoding of the values of the first time-frequency-domain
representation; and
providing (2830) the watermark signal on the basis of the second time-frequencydomain
representation.
14. A method (2900) for providing binary message data in dependence on a
watermarked signal, the method comprising:
providing (2910) a first time-frequency-domain representation of the watermarked
signal;
deriving (2920) a second time-frequency-domain representation from the first timefrequency-
domain representation, such that values of the second time-frequencydomain
representation are dependent on phase differences between two
corresponding values of the first time-frequency-domain representation; and
using (2930) the second time-frequency-domain representation to determine a
synchronization information, which is used for providing the binary message data,
or for extracting the binary message data from the watermarked signal.
A computer program for performing the method according to claim 13 or claim 14,
when the computer program runs on a computer.
A watermark generator (101; 2400) for providing a watermark signal (101b; 2420)
in dependence on binary message data (101a, m ; 2410), the watermark generator
comprising:
an information processor (303,304,305; 2430) configured to spread the binary
message data to a plurality of frequency bands or frequency subbands and to
provide, in dependence on information units of the binary message data, a first
time-frequency domain representation (b(i,j); 2432), values of which represent the
binary message data for a plurality of frequency bands or frequency subbands and
time blocks; and
a differential encoder (306; 2440) configured to derive a second time-frequency
domain representation (306a, bdi (i,j); 2442) from the first time-frequency-domain
representation, such that the second time-frequency-domain representation
comprises a plurality of values, wherein a value bdif (i,j) of the second timefrequency-
domain representation is a function of a value bdi (ij-l) of the second
time-frequency-domain representation and of a value b(i,j) of the first timefrequency-
domain representation and wherein a difference between two values
( di (ij), iff (i,j-l)) of the second time-frequency-domain representation represents
a corresponding value of the first time-frequency-domain representation, in order to
obtain a differential encoding of the values of the first time-frequency-domain
representation; and
a watermark signal provider (307; 2450) configured to provide the watermark
signal on the basis of the second time-frequency-domain representation.
A watermark decoder (200; 2500) for providing binary message data (202a; 2520)
in dependence on a watermarked signal (200a; 2510), the watermark decoder
comprising:
a time-frequency-domain representation provider (1600; 2530) configured to
provide a first time-frequency-domain representation (b norm(j); 2532) of the
watermarked signal, wherein values b o m(j) of the first time-frequency-domain
representation comprise information about the phase of signal components at
frequency and time instant j ;
a differential decoder (1608; 2540) configured to derive a second time-frequencydomain
representation ( , ( ) ; 2542) from the first time-frequency-domain
representation, such that values of the second time-frequency-domain
representation are dependent on phase differences between two corresponding
values (bi r (j), b o (j-l) of the first time-frequency domain representation; and
a synchronization determinator (201; 2550) configured to obtain a synchronization
information (205; 2552) on the basis of the second time-frequency-domain
representation; and
a watermark extractor (202; 2560) configured to extract the binary message data
from the first time-frequency-domain representation of the watermarked signal or
from the second time-frequency-domain representation of the watermarked signal
using the synchronization information.
A watermark decoder (200; 2600)for providing binary message data (202a; 2620)
in dependence on a watermarked signal(200a; 2610), the watermark decoder
comprising:
a time-frequency-domain representation provider (1600; 2630) configured to
provide a first time-frequency-domain representation (b, o 1(j); 2632) of the
watermarked signal, wherein values bj o (j) ° e fi s time-frequency-domain
representation comprise information about the phase of signal components at
frequency f and time instant j ;
a differential decoder (1608; 2640) configured to derive a second time-frequencydomain
representation b j 2542) from the first time-frequency-domain
representation, such that values of the second time-frequency-domain
representation are dependent on phase differences between two corresponding
values (bi or (j), b nomi(J-l) of the first time-frequency-domain representation; and
a watermark extractor (202; 2650) configured to extract the binary message data
from the second time-frequency-domain representation.
19. A method (2800) for providing a watermarked signal in dependence on binary
message data, the method comprising:
spreading the binary message data to a plurality of frequency bands or frequency
subbands, to provide (2810), in dependence on information units of the binary
message data, a first time-frequency-domain representation, values of which
represent the binary message data for a plurality of frequency bands or frequency
subbands and time blocks;
deriving (2820) a second time-frequency-domain representation from the first timefrequency
domain representation, such that the second time-frequency-domain
representation comprises a plurality of values, wherein a value bdi i, ) of the
second time-frequency-domain representation is a function of a value bdif (i,j-l) of
the second time-frequency-domain representation and of a value b(i,j) of the first
time-frequency-domain representation and wherein a difference between two values
of the second time-frequency-domain representation represents a corresponding
value of the first time-frequency-domain representation, in order to obtain a
differential encoding of the values of the first time-frequency-domain
representation; and
providing (2830) the watermark signal on the basis of the second time-frequencydomain
representation.
20. A method (2900) for providing binary message data in dependence on a
watermarked signal, the method comprising:
providing (2910) a first time-frequency-domain representation of the watermarked
signal, wherein values b o ( ) of the first time-frequency-domain representation
comprise information about the phase of signal components at frequency f and time
instant j ;
deriving (2920) a second time-frequency-domain representation from the first timefrequency-
domain representation, such that values of the second time-frequencydomain
representation are dependent on phase differences between two
corresponding values of the first time-frequency-domain representation; and
using (2930) the second time-frequency-domain representation to determine a
synchronization information, which is used for providing the binary message data,
or for extracting the binary message data from the watermarked signal.