Wideband transmitter/receiver arrangement for
multifunctional radar and communication
TECHNICAL FIELD
The present invention relates to a wideband multifunctional transmitter and
receiver arrangement, preferably for transmitting and receiving at VHF, UHF
or in the microwave bands. Such an arrangement can simultaneously and in
the same frequency band serve as radar, surveillance, and communication
system.
BACKGROUND ART
In its simplest form, prior art continuous wave CW radar emits a single
frequency continuous wave. By measuring the Doppler shift of the return
signal, the CW radar can determine the velocity of detected objects. By linear
frequency modulation (sawtooth) of the continuous wave in a CW radar, a
frequency-modulated continuous wave FMCW radar system is provided. The
frequency modulation of the CW has the advantage that also the distance to
a detected object can be determined. Such a FMCW radar system is known
from US 3,789,398. The FMCW radar is mainly motivated by the fact that it
minimizes the peak-to-average power ratio for the radar transmit signal. Its
applications are thus as whispering radar, used to defeat detection of the
radar signal.
The FMCW radar is however limited to said radar function and lacks any
multifunctional capability.
Moreover, the FMCW radar exhibits poor signal cohabitation between its
radar function and environmental signals.
There is thus a need for an improved transmitter/receiver arrangement
removing the above mentioned disadvantages.
SUMMARY
The object of the present invention is to provide an inventive wideband
transmitter/receiver arrangement for transmitting and receiving
electromagnetic waves, and a method for transmitting and receiving
wideband electromagnetic waves, where the previously mentioned problems
are partly avoided. This object is achieved by the features of the
characterising portion of claim , wherein said transmitter/receiver
arrangement comprises a digital arbitrary waveform generator AWG
connected to a transmitter, and wherein said waveform generator is
configured to generate an arbitrary waveform within a given bandwidth. Said
transmitter/receiver arrangement further comprises an antenna arrangement
configured to emit a transmitter signal and to receive an incident signal, and
a receiver configured to receive a receiver signal. Said transmitter/receiver
arrangement further comprises an analogue isolator connected to said
antenna arrangement, said transmitter, and said receiver. Said analogue
isolator is adapted to route said transmitter signal from said transmitter to
said antenna arrangement, and said incident signal from said antenna
arrangement to said receiver, and to isolate said transmitter signal from said
receiver signal. Said receiver is adapted to cancel any residual transmitter
signal in said receiver signal by means of at least one digital model of at least
said isolator, said antenna arrangement, and said transmitter.
Said object is further achieved by the characterizing portion of claim 7,
wherein said method comprises the steps of generating an arbitrary
waveform within a given bandwidth by means of a digital arbitrary waveform
generator AWG connected to a transmitter, routing a transmitter signal from
said transmitter to an antenna arrangement and an incident signal on said
antenna arrangement to a receiver, and isolating said transmitter signal from
said receiver signal, by means of an analogue isolator connected to said
antenna arrangement, said transmitter, and said receiver, cancelling any
residual transmitter signal in said receiver signal by means of at least one
digital model of at least said isolator, said antenna arrangement and said
transmitter.
Further advantages are achieved by implementing one or several of the
features of the dependent claims.
The inventive wideband transmitter/receiver arrangement serves as core for
a multifunctional radar, surveillance, and communication system. The system
is, due to its excellent leakage cancellation, particularly suitable when
compact equipment is required, for example on small or medium sized
Unmanned Airborne Vehicle UAV applications. The inventive
transmitter/receiver arrangement also leads to reduced weight and volume
compared to pulsed radar equipment, and in that transmitter peak power is
reduced using CW radar.
The present invention intends to combine the radar function with other uses
of electromagnetic signal reception and transmission. A fundamental
requirement is that the radar is not pulsed as ordinary radar but operates on
a continuous waveform principle, i.e. as the FMCW radar. However, in
contrast to the FMCW radar the very idea of the invention is that the
waveform must be allowed to be arbitrary and e.g. modulated to serve as a
communication signal, whilst still fulfilling its role as radar transmit signal. In
this respect known FMCW radar designs are of little use. Indeed the main
challenge of continuous wave radar is to achieve isolation between transmit
and receive and the methods to do so in the present invention are very
different from those of the FMCW radar.
The multifunctional capability of the inventive wideband transmitter/receiver
arrangement according to the invention includes providing:
1. Radar functions providing microwave and/or low frequency Synthetic
Aperture Radar SAR.
2. Passive surveillance of the signal environment to establish ongoing
transmissions.
3. Communication capacity to downlink radar and/or passive surveillance
data acquired by the transmitter/receiver arrangement.
4 . Communication relay service between different locations/objects, for
example between ground units, for extending the range of ground
communication links.
Wideband in this concept implies that the system covers bandwidths of at
least octave order with a centre frequency anywhere from VHF to well up in
the microwave region, i.e. 50 MHz - 5 GHz.
Below, aspects of the inventive transmitter/receiver arrangement with respect
to radar operation and signal cohabitation will be described. Aspects of the
transmitter/receiver arrangement with respect to passive surveillance and
communication will also be described but not at the same level of technical
detail. Indeed it is obvious to the skilled person in the area of radar
technology and RF communication how to adopt the possibility of arbitrary
waveforms for the purpose of letting these contain modulated communication
and radar data downlink signals. Moreover that the ability of continuous wide
band signal reception enables passive signal surveillance. Finally the
combination of continuous operation and arbitrary waveform generation
enables a variety of communication relay services between different locations
in which the radar mimics and repeats the particular modulation the
communication channel exploits.
The inventive transmitter/receiver arrangement provides an improved level of
signal cohabitation between the radar function and environmental signals.
Improvements are possible both in that the radar will not cause a strong
interference to communication, and in that the radar will be robust with
respect to the interference caused by communication signals.
The issue of cohabitation concerns ways to:
. Mitigate the interference caused by the radar to these other services,
and
2 . Mitigate the interference of these services on the radar operation.
Cohabitation issues tend to become important for a!i radar frequencies as
competing uses of the spectrum spreads upwards in frequency to established
microwave radar bands. However the issue of cohabitation is particularly
crucial for frequencies below about 1 GHz. A large fraction of all
communication services are situated in this band. SAR operating below 1
GHz and with bandwidths of the order 00 MHz has the ability of penetrating
ground and vegetation, and thus detect objects and structures hidden from
ordinary sensors. There is an evident clash in that communication and this
type of radar share the same frequencies. One application of the present
invention is to moderate this clash.
Consider cohabitation interference mitigation of type 1. This could potentially
be reduced if the radar is given an ability to receive and immediately repeat
incident signals. This will have a similar effect in a communication channel as
multipath effects to which communication modulation schemes must be
significantly robust. Modern digital modulation techniques such as OFDM are
particularly robust.
Consider cohabitation interference mitigation of type 2. As for robustness of
the radar with respect to environmental signal interference, there are several
schemes. Narrow band interference can be removed by creating a notch
corresponding to the communication band in the range (or "fast time")
spectrum. For wide band inference, e.g. digitally modulated television
broadcast, this method results in a seriously large degradation of radar data
and SAR image quality. Better rejection schemes are to use either
cancellation or 2-dimensional notching. The latter method is directly
applicable when the radar signal adapts to incident signals as described. This
is in fact the essence of the patent application WO 2004/097451 , in which the
radar mimics an analogue television signal, such that the received television
signal will have nearly zero Doppler (or "slow time") bandwidth in the SAR
signal processing. It can thus be 2-dimensionally notched, i.e. notched in
both the range and the slow time spectrum.
Cancellation, as a method of interference mitigation of type 2, is well suited to
digitally modulated signals with built in error correction. Cancellation can be
applied as and when the radar ground response signal is so weak that
decoding and error correction will fully retrieve the interfering signal. This can
then again be encoded, subtracted and thereby mitigated in the overall
received signal.
The situation of the radar response signal being weak compared to the
interfering signal is what is commonly experienced. It is also the condition for
the scheme of transmitter signal adaption to the incoming signal, since this
will re-transmit the radar response signal, which must not impair the decoding
of the re-transmitted interference signal. A suitable scheme would thus be to
adapt transmitter power levels so that the radar response remains a small
part of any environmental signal for which one wishes to avoid causing
interference.
To achieve above mentioned robustness capabilities, the radar according to
the invention should fully capture the interfering signals and retransmit these
without interruption. Thus the radar should operate on a continuous transmit
waveform principle, and since the radar is required to have an ability to re¬
transmit any incident signals, it must apart from continuous operation have
the added capability of being able to transmit waveforms of arbitrary shape.
Such radar, which is the core of the invention, will be called Arbitrary
Waveform Continuous Wave AWCW radar. An important candidate for
application is as a compact and multifunctional sensor onboard UAVs. The
smallness of the UAVs, for which the invention is consider in particular, is in
fact an important enabling factor for the realization of the AWCW radar. The
radar will operate at short ranges, i.e. with a low transmitter power.
The concept of transmitting an arbitrary wave is not new. Typically arbitrary
waveforms are used in the sense pseudo-random noise signal, repeated in a
fixed form from pulse to pulse. The present invention however launches the
new concept that the waveform can be entirely non-repetitive for any length
of time and in fact be able to continuously copy incident signals, which are re¬
transmitted and used as a part of a total transmit waveform. Moreover this
invention launches the concept that this non-repetitive waveform can be
continuously transmitted without interrupts for receiving as in ordinary pulsed
radar. The arbitrary waveform is suitably generated by an arbitrary waveform
generator, which synthesize the waveforms using digital signal processing
techniques.
Evidently an AWCW radar device has apart from its applications as radar
also the ability to act as a communication link, which is an immediate
consequence of its capability to continuously copy incident signals in the
transmitted waveform. Moreover, since the transmit waveform is arbitrary it
can be modulated to convey sensor data information and thus used to
downlink these data. Since the AWCW radar operates with continuous
reception it can also fulfil a role of performing signal surveillance within the
frequency band for which it is designed. This is done in parallel without any
requirement on time sharing or otherwise splitting the capability of the radar
device between different applications.
The theory behind the inventive transmitter/receiver arrangement in a radar
application is explained below. Assume that a continuous and arbitrary
transmitter signal is (t) and assume that the received signal is r(t). The
signal a(t) is assumed to have bandwidth B and is required to have two
basic properties: For some pre-se!ected period of time T , it is required that:
1. the signal a(t) spans B with equal power density for any time interval
of length T.
2 . the signal a{t) does not repeat itself in the sense that the cross
correlation between is uniformly low for all
times .
Reconstruction of range reflectivity , where is radar slow time can
under these conditions be obtained by correlation
Because of said basic property 1, formula (I) provides bandwidth limited
range resolution with a given side lobe level t is seen that if would have
been periodic with period T there would have been range ambiguities with a
period Hence the absence of such ambiguities is due to said
basic property 2. The suppression depth of range ambiguities is limited by
the time-bandwidth product . Indeed, for optimal waveforms, the
suppression depth will equal BT for all ranges and times. A suitable
suppression depth of 40 decibels dB and B= 1 GHz would correspond to T~
10 m s .
By (I), f(R,t) is low pass filtered to a highest slow time frequency \/T . The
required slow time bandwidth of f (R, ) is set by the SAR Doppler bandwidth.
When the transmitter/receiver arrangement is used in an UAV based SAR
radar, this is typically of the order 100 Hz and thus significantly smaller than
1/ T. In the subsequent SAR processing f(R,t) will be further low pass
filtered to a bandwidth corresponding to the reciprocai of the SAR integration
time of possibly several seconds. As for interference, the degree of
suppression will depend on whether the transmit signal adapts to the
interference or not, as discussed above.
If the transmitter signal does not adapt and thus is uncorreiated to an
interference signal, then the interference will be suppressed by (I) just as any
noise component added to r{t). In effect the suppression ratio is set partly by
the averaging occurring in (1) and partly by the further time averaging f{R,t)
set by the Doppler bandwidth. Thus for a bandwidth B= 1 GHz and Doppler
bandwidth 100 Hz, the suppression rate will be 80 dB for interference as well
as noise. One difference in this suppression ability between a AWCW radar
and e.g. a stepped frequency radar is that the receiver in an AWCW radar
receives the full bandwidth at all times, contrary to a stepped frequency
radar, which receives only a subband, i.e. step, of the total frequency of the
radar bandwidth.
If the transmitter signal does adopt, i.e. mimics the interference within the
interference bands, then the interference will be substantially stationary in
slow radar time. The method of interference mitigation is in that case to
reduce the bandwidth of the interference within the Doppler spectrum, and
notch the interference 2-dimensionally - both in the range and Doppler
spectrum. The interference is thus effectively removed - not merely
suppressed. The impact on SAR image quality is small since the amount of
data removed by 2-dimensional filtering is small. As for cancellation, the level
of suppression which can be achieved by this methods will add to the
cancellation depth obtainable when the transmit signal does not adapt, as
described above.
The overriding concern of AWCW radar is to obtain sufficient isolation
between the transmitter and receiver signals. The achievable isolation puts a
limit on allowable transmitter power level, since the isolation should reduce
this level to the receive noise level determined by environmental noise or the
receive channel noise. The larger the bandwidth of the radar, the larger the
receive channel noise level. On the other hand the possible isolation
diminishes with increased bandwidth, due to e.g. the reduced dynamic range
of analogue to digital converters ADCs and digital to analogue converters
DACs operating at high sampling rates.
Narrow band isolation is achievable by analogue cancellation techniques, as
is illustrated in FMCW radars. For wideband isolation however, as required
by the inventive transmitter/receiver arrangement, fully developed means for
digital cancellation is required. A wideband isolation approach comprising a
combined analogue and digital cancellation scheme providing isolation in
three logical steps is disclosed:
1. An initial analogue cancellation in an isolator of the
transmitter/receiver arrangement, which analogue cancellation is
required to be highly linear and time stable and thus accurately
characterizable by a digital model.
2 . A first analogue and digital cancellation step, which removes any
analogue cancellation error with respect to a digital transmitter signal,
by making use of that this is accurately known and also that the
analogue cancellation system characteristics are accurately known.
3. A second digital cancellation step, which removes the analogue
cancellation error for the transmitter noise by making use of that this
noise is of relatively low intensity and thus can be accurately AD
converted.
No accurate transmitter model is required, since step 3 takes care of such
model errors. Note however that the achievable isolation depends on two
conflicting requirements, i.e. A) the ability of the analogue system to achieve
a high degree of isolation by itself, and ) the characterizabiiity of the
analogue system required for obtaining a high isolation in steps 2 and 3.
Characterizability here means that an analogue system may possibly be
designed to yield a high degree of isolation but only in such a way that
leakage residuals are less linear or less stable in time. Hence the degree of
isolation achievable in the digital suppression stages will be limited for such a
system. A suitable trade off between analogue and digital cancellation must
be found.
Another limiting factor is the accuracy of the required DACs. There are thus
three main parameters which will limit isolation i.e. the analogue isolation
level, the characterizability of the analogue cancellation and the DAC
resolution. ADC resolution is a lesser concern since signals to be ADconverters
are all low level.
BRIEF DESCRIPTION OF THE DRAWINGS
The present invention will now be described in detail with reference to the
figures, wherein:
Figure 1 shows the system building blocks of AWCW radar according to
the invention;
Figure 2 shows the leakage and noise reduction stages in the receiver
according to the invention;
Figure 3 shows an isolator design according to the invention;
Figure 4 shows an antenna arrangement of the isolator of figure 3
provided on an UAV;
Figure 5A shows a calibration arrangement for determining a digital model
of a transmitter according to the invention;
Figure 5B shows a calibration arrangement for determining the
characteristics of a subtraction unit according to the invention;
Figure 5C shows a calibration arrangement for determining a digital model
of an isolator and antenna arrangement according to the
invention;
Figure 5D shows a calibration arrangement for determining the
characteristics of a subtraction unit according to the invention.
DETAILED DESCRIPTION
In the following only one embodiment of the invention is shown and
described, simply by way of illustration of one mode of carrying out the
invention.
Figure 1 shows the system building blocks of AWCW radar according to the
invention. A digital arbitrary waveform generator AWG 1 feeds a transmitter
19 comprising a first digital to analogue converter DAC 2 , and a power
amplifier 3. The signal generated by the AWG 1 is thus converted to an
analogue signal, which is amplified in the power amplifier 3. The transmitter
9 subsequently feeds a transmitter signal S 1 into a RF isolator 4, which
serves to control the direction of signal flow. The isolator 4 is further
connected to an antenna arrangement 5 , and to a receiver, such that signals
5 1 from the transmitter 19 are routed by means of the isolator 4 to the
antenna arrangement 5 and isolated from the receiver, and incident signals
52 on the antenna are routed solely to the receiver. The term isolator (4) is
here considered to encompass any type of device capable of routing said
signals S 1, S2 and isolating the transmitter signal S 1 from entering the
receiver, such as isolator, circulator, and power splitter/combiner networks
etc.
A receiver signal S3 from the isolator is fed to a first subtraction unit 10 via a
first attenuator 9. The receiver signal S3 has at least three terms, viz. (a) any
exterior signal S2 incident on the antenna arrangement 5 , (b) antenna
reflections and (c) isolator leakage.
A first cancellation signal S4 is generated by feeding the generated waveform
signal from the AWG 1 into a digital model 6 of the transmitter 19, then to a
first digital model 7 of the isolator 4, the antenna arrangement 5, and a first
subtraction unit 10, and then to a second DAC 8. Said first cancellation signal
S4 is subsequently fed from the second DAC 8 to the first subtraction unit 10,
which is an analogue subtraction unit 10. The signal terms (b) and (c)
mentioned above are removed in the first subtraction unit 0 , insofar the
digital model 6 of the transmitter 9 and said first digital model 7 correctly
describes the physical behaviour of the transmitter 19 , isolator 4, antenna
arrangement 5, and first subtraction unit 10, i.e. how accurate they depict the
transfer characteristics of said devices 19, 4, 5, 10. A discrepancy forming at
least transmitter noise is most likely to remain in the signal output of the first
subtraction unit . The intensity of the signal output from the first subtraction
unit 10 is significantly reduced compared to the levels of the receiver signal
S3 and the first cancellation signal S4 fed to the first subtraction unit 10. The
low level signal output from the first cancellation unit 10 is fed to a first
analogue to digital converter ADC and subsequently to a second
subtraction unit 12, which is a digital subtraction unit 12.
A second cancellation signal S5 is generated by first feeding the output of the
digital model 6 of the transmitter 19 to a third DAC 13 whose output signal is
fed to a third subtraction unit 15 . Another input signal to the third subtraction
unit 15 is supplied from the output of the transmitter 9, having passed
through a second attenuator 14. The output signal from the third subtraction
unit 15 corresponds therefore to the difference between the digital model 6 of
the transmitter 9 and the actual transmitter output, i.e. any transmitter noise
not subtracted in the first subtraction unit 10 due to an inaccurate digital
model 6 of the transmitter , amongst others. The signal output of the third
subtraction unit 15 will be analogue to digital converted in a second ADC 16
and subsequently fed to a second digital model 17 of the isolator 4, the
antenna arrangement 5, and the third subtraction unit 15. This signal will thus
match the transmitter noise term fed into the second subtraction unit 2 from
the first ADC 1 . Hence the output of the second subtraction unit 12 will
correspond to any signal S2 incident on the antenna arrangement 5, apart
from the above mentioned errors related to the characterizability limitations
discussed above and resolution of the first, second and third digital to
analogue converters 2, 8 , 3, which also limits the possible level of
cancellation in the first and third subtraction units 10, 15.
Consequently, the route of the first cancellation signal S4 accurately handles
the large amplitude part of the cancellation by omitting analogue to digital
conversion. This omission is possible because the large amplitude part of the
signal is known. The route of the second cancellation signal S5 handles
residuals, which are unknown but have small amplitude and thus allow being
analogue to digital converted without critical accuracy being lost.
Signals S 1 generated by the transmitter 9 and coupled to the second
attenuator 14 and emitted by the antenna arrangement 5, as well as signals
S2 received by the antenna arrangement 5 and coupled to the first attenuator
9 are high intensity signals, whereas the signals at the output of the second
subtraction unit 12 are just above ADC quantization level, i.e. ADC interior
noise level. All other signals in the block diagram of figure 1 are signals with
reduced levels not to saturate the first and second ADCs 11, 16.
This reasoning can be illustrated using numerical examples:
Assuming a bandwidth of the system of 1 GHz, and 3000 K (degrees Kelvin)
internal noise temperature and 30000 K external noise, wherein the elevated
external noise is caused by an abundance of interference. The noise level
"kTB " is calculated according to:
wherein "kTB " is the noise temperature in watts, kB is the Boltzmann
constant ( 1 .381 c 10~23 J/K, joules per Kelvin), Ts is the noise temperature (K),
and B is the noise bandwidth (Hz). The internal noise level "kTB " is thus -
73 dBm and the external noise level -63 dBm. Consider -70 dBm to be the
LSB (Least Significant Bit) power for the first ADC 11. If this first ADC 1 has
9 bits, it provides a maximum allowable input power level of about -20 dBm.
The 50 dB ADC dynamic range is also assumed to be the accuracy of the
radar system digital models.
For state of the art digital to analogue converters DACs noise levels of -70
dBm are assumed achievable at 0 dBm output level. Hence, this means that
DAC errors are -70 dB with respect to full scale. The transmitter amplifier 3
will be assumed to output 1 dBW, requiring 30 dB gain. A digital model of a
power amplifier is not expected to be highly accurate. Assuming 10 dBm
transmitter noise allowing for transmitter digital modelling errors. This
corresponds to a transmitter noise factor of 63 dB.
Figure 2 illustrates the leakage and noise reduction stages in the receiver,
wherein the Y-axis represents the signal power in dBm and the X-axis
illustrates sequential parts of the receiver, e.g. "9" refers to the first attenuator
9, and " 0" refers to the first subtraction unit 0 etc. Assume isolator leakage
to be -20 dB. This puts the transmitter signal leakage 20 at the isolator output
to 10 dBm, the transmitter noise leakage 2 1 to -10 dBm, and the external
noise level 22 to -63 dBm.
With a 10 dB value for the first attenuator 9 , the transmitter signal leakage 20
becomes in parity to the output level of the second DAC 8, whereupon they
can cancel in the first subtraction unit 10. The transmitter noise leakage 2 1 at
the output of the first attenuator 9 is -20 dBm, which is full scale for the first
ADC 1. A strong interference signal may also yield - 10 dBm at the isolator
output and correspond to full scale for the first ADC 1.
With a -70 dB accuracy for the DAC resolution and a -50 dB accuracy for the
said first digital model 7, the transmitter signal leakage 20 is reduced in the
first subtraction unit 10 to -50 dBm. The transmitter noise leakage 2 1 is not
reduced from its full scale value, and the external noise 22 is elevated to -70
dBm due to added noise from the second DAC 8.
The second attenuator 1 must have a value 30 dB to reduce transmitter
signal levels to 0 dBm. The transmitter signal S 1 from the output of the
transmitter 9 and the output of the digital model 6 of the transmitter 9 will
thus cancel in the third subtraction unit 15, while transmitter noise remains
unmitigated and at its value after the second attenuator 14. With the 30 dB
attenuation in the second attenuator 14 this value is -20 dBm, which also will
be full scale for the second ADC 16. In the second subtraction unit 12 the
transmitter noise will cancel. Cancellation depth will be given by the
quantization noise of the first and second ADCs 1 , 16 as well as the
accuracy of said second digital model 17. Based on the assumed
performance of the first and second ADCs 11, 16, this value is -70 dBm.
In the range reconstruction part the transmitter signal leakage 20 and
transmitter noise leakage 2 1 are further reduced in fact, the transmitter
signal leakage 20 is a replica of the transmit signal delayed by its
transmission through the system, as well as distorted by uncontrolled
leakage, dispersion and other linear effects, which to a large part are taken
care of by the digital corrections imposed. The residual of the transmitter
signal leakage 20 imposed on the signal in the range-time reflectivity
reconstruction in the signal extraction unit 18 will however remain
concentrated to very short delays after transmission and thus mainly affect
the signal for very small radar ranges, at which relevant radar data are not
expected. Being temporally stable it will also be narrow band in the Doppler
spectrum. In ail the residual transmitter signal leakage 20 will have a small
effect indeed on the reconstructed range reflectivity data f{R,t).
The residual transmitter noise leakage 2 1 becomes suppressed in proportion
to the time bandwidth product of the range reconstruction in the same way as
any other noise component added to the received signal.
As described in this example the AWCW radar principle is best suited to
moderate and low transmit power and in particular in application such radars
operating below 1 GHz, where the external noise levels are elevated
compared to the purely thermal environmental noise.
Obvious to the skilled person is that in order to achieve the required
cohabitation and communication capabilities, the signal extraction unit 8
must contain means to extract communication subbands and to transfer and
insert these signals in the AWG 1, as a part of the transmitted waveform. An
ability to demodulate the extracted signals is also required in some of the
cohabitation and communication modes. Thus there is a digital part of the
system which however is schematically obvious and therefore not shown in
Figure 3. This part connects the signal extraction unit 18 and the AWG 1 and
furthermore contains suitable digital encoding and decoding devices, as well
as a system controller guiding the signal extraction in the signal extraction
unit 18, the waveform generation in the AWG 1, and encoding/decoding
processes, as required.
The isolator 4 is a critical part of the AWCW radar design. For one it is
difficult to achieve analogue wide band isolation of any significant depth by
such a device. Presently isolation value requirements of at least 20 dB are
dictated by the assumed -70 dB DAC accuracy.
The other issue regarding isolation is characterizability of the analogue
components, implicitly meaning their linearity and time stability. As for
linearity, there are both issues concerning intermodu!ation and (since octave
operation is required) harmonic distortion. The accuracy in these aspects
must be of the order -70 dB, i.e. the order of the assumed DAC accuracy of
50 dB in the example above. A very high degree of characterizability typically
requires passive components and preferably linear materials. Hence both
semiconductor and ferrite materials may imply linearity limitations and must
be used with particular caution in the present application.
Linear wide band isolator systems can be based on signal cancellation.
Such a cancellation system consists of two channels which in independent
ways transmit and receive but which cancel each other as regards transmitter
leakage through either channel. As for obtaining a large degree of
cancellation in a wide band system, the challenge is the many degrees of
freedom across which the leakage transfer function in the two channels must
be similar. This number of degrees of freedom is basically set by the delay
occurring in the system times the signal bandwidth. Hence a circuitry of small
extension is normally characterized by short delays and a simple transfer
function also in the wide band case. For AWCW radar operating at traditional
microwave frequencies, the extension of the system itself is typically many
wavelengths. Below 1 GHz, the extension of the system may be less than
wavelength order. However in this case the extension of the radar platform
becomes decisive, since low frequency antennas are never highly directive
and will interact with the platform structure to non-negligible extent. It follows
that irrespective of frequency the system will have delays of several periods
of the frequencies at which the system operates. An octave bandwidth
system will thus correspond to a transfer function of non-trivial shape.
There is one particular way in which such a wide band cancellation scheme
for aircraft or UAV operation may be carried out, and which is well suited for
frequencies below 1 GHz. The principle relies on the right left symmetry of
the airframe and adopts two cancellation channels, which are symmetric in
this sense. The device operates simultaneously to both right and left. The
transmitter signal leakages through either of the two channels will be the
same and the two leakage signals can thus be brought to cancel each other.
Figure 3 illustrates the principle.
In figure 3, a preferred realization of a wide band isolator system is
described, wherein the cancellation principle is based on first and second
circulators 32, 36, preferably 90-degree hybrids, and a power combiner 39,
preferably a 180-degree hybrid. The transmitter signal S 1 from the
transmitter is fed via a power splitter 3 1 to input ports P 11 of the first and
second 90-degree hybrids 32, 36. Said first 90-degree hybrid 32 in turn feeds
a first antenna element 34 from a coupled port 2, and a second antenna
element 33 from a transmitted port P22. Said second 90-degree hybrid 36 in
turn feeds a third antenna element 38 from a coupled port P 2, and a fourth
antenna element 37 from a transmitted port P22.
The first and second antenna element 34, 33 form a right antenna 42, and
the third and fourth antenna element 38, 37 form a left antenna 43. The
separation between said antenna elements 33, 34, 37, 38 in either antenna
42, 43 is a quarter of a wavelength at the centre frequency. Hence each
antenna 42, 43 will propagate the transmitter signal S 1 along the separation
axis between the antenna elements 33, 34, 37, 38, and in the direction of a
phase delay generated by the fact that a signal fed to input port P 1 of a 90-
degree hybrid 32, 36 will be supplied to transmitted port P22 without phase
shift, whereas the same signal will be phase shifted 0 - 90° on its way from
input port P to the coupled port P 2. Consequently, the direction of the
phase delay goes from the coupled port P 2 to the transmitted port P22, and
the transmitter signal S1 will thus propagate in the direction of the block
arrows illustrated in Figure 3 .
Figure 4 shows the antenna arrangement 5 on an airborne vehicle 44, such
as an UAV. Each antenna 42, 43 is oriented so that the propagation main
beam is orthogonal to the airframe centre plane, with the right antenna 42
propagating to the right and the left antenna 43 propagating to the left of the
vehicle 44. The right and left antennas 42, 43 are located in a right/left
symmetric fashion with respect to the airframe centre plane.
Return signals from the left antenna 43 are collected from the isolator port
P21 of the second 90-degree hybrid 36 isolated from transmission and fed to
a first input port P 1 of the 180-degree hybrid 39. Similarly, return signals
from the right antenna 42 are collected from the isolator port P21 of the first
90-degree hybrid 32 and fed to a second input port P21 of a 180-degree
hybrid 39. The 180-degree hybrid 39 further comprises a delta port P12 and
a sum port P22. Due to the inherent function of the 180-degree hybrid 39, the
output signal at sum port P22 is proportional to the sum of the signals at the
first and second input ports P 11, P21 , whereas the output signal at the delta
port P12 is proportional to the difference between the signals at the first and
second input ports P 11, P21. Hence the collected signals from the right and
left antennas 42, 43 are provided added and subtracted with an overall phase
modification in said delta and sum ports P12 and P22 of the 180-degree
hybrid 39. The first switch 35 is either set to transfer the transmitter signal S 1
unaffected in which case the second switch 40 is set to feed the difference
signal to the to the radar receiver, or the first switch 35 is set to a 180-degre
phase shift in which case the second switch 40 is set to feed the sum signal
to the radar receiver. In either case, the second switch 40 feeds the
remaining signal out from the 80-degree hybrid 39 into a resistive load 4 1.
If no reflections occur at the coupled and transmitted ports P12, P22 in the
90-degree hybrids 32, 36, an output at the isolator port P21 less than -20 dB
with respect to the input signal is achievable, also in broad band applications.
However, as a consequence of the 90° phase shifts occurring in both the
forward and reflected direction in the first and second 90-degree hybrids 32,
36, reflections due to inferior Voltage Standing Wave Ratio VSWR of the
antenna elements 33, 34, 37, 38 in figure 3 will combine coherently at the
isolator ports P21 of the first and second 90-degree hybrids 32, 36 and thus
cause leakage residues. In contrast any coupling between antenna elements
33 - 34 and 34 - 33 cancel each other, as do coupling between antenna
elements 37 - 38 and 38 -» 37 at the isolator ports P21 of the first and
second 90-degree hybrids 32, 36. The couplings going from the antenna
elements in one antenna to the other will not cancel. Just as the antenna
reflections these will thus contribute to the signal residues at the isolator
ports P21 of the 90-degree hybrids 32, 36.
However given the right left symmetry of antenna arrangement 5 and
airframe, the leakage residues at the isolator ports P21 of the 90-degree
hybrids 32, 36 are equal in magnitude and equal or opposite in phase,
depending on the setting of the first switch 35 as explained. They will thus
always cancel n the output of the 180-degree hybrid 39 feeding the radar
receiver via the switch 40. This conclusion is valid both for the direct
reflections and couplings occurring in or between the antenna elements 33,
34, 37, 38, and also when these effects are extended to include all the
electromagnetic interactions with airframe.
In applications below 1 GHz, the antenna elements 33, 34, 37, 38 are
typically broad band monopoles or dipoles. They can be vertically or
horizontally polarized in figure 3 they are vertically polarized and mounted
underneath the UAV fuselage. A mechanism making the antenna elements
foldable or retractable might be required for their protection during starting
and landing.
The omnidirectional property of radiation, created by this type of isolator 4 , is
acceptable for the applications 2-4 mentioned. However, in particular in the
case of SAR (application 1) one requires one-sidedness for non-ambiguous
SAR image formation. Such one-sidedness can be achieved over time. The
first and second switch 35, 40 provide this function. While reversing the
polarity of the transmitter signal S going to one side of the antenna
arrangement 5, the polarity of the return signal is reversed for the same side.
By the correlation of equation (i) in which the same un-shifted transmitter
signal replica is used for both sides either the sum or difference
of the right and left radar reflectivity functions
is obtained. Switching at a period sufficiently larger
than the Doppler bandwidth (of the order 100 Hz) the resulting reflectivities
can be linearly combined to extract individually.
For SAR operation switching must occur sufficiently fast with respect to slow
time bandwidth of the ground reflectivity. A typical switching frequency might
be 1 kHz. Switching could for instance be made synchronous with the
orthogonal frequency-division multiplexing OFDM symbol rate in cohabitation
schemes with television OFDM modulation.
The first subtraction unit 10 is suitably based on a 180-degree hybrid. In
contrast to the isolator 4 , the first subtraction unit 10 is required to achieve a
very high degree of cancellation. The required levels of cancellation are
obtained by a pre-distortion of the transmitter signal S 1, carried out in the
digital mode! 6 of the transmitter 19 and said first digital model 7 . Apart from
the transfer characteristics of the transmitter 19, the isolator 4, and antenna
arrangement 5 , the pre-distortion must account for the non-perfectness of the
180-degree hybrid forming the first subtraction unit 10. The third subtraction
unit 15 is suitably designed in the same manner of a 180-degree hybrid, and
is subjected to similar requirements as the first subtraction unit 10.
The cancellation step in the second subtraction unit 12 is performed entirely
digital. Said second digital mode! 7 follows exactly the same principles as
the digital model 6 of the transmitter 19, and said first digital model 7, but is
acting on transmitter noise. All digital models 6 , 7, 17 included in the
receiver/transmitter arrangement are preferably implemented as digital finite
impulse response FIR filter. The characteristics of such filters and the
techniques for modelling them are generally known. Those skilled in the art
who have the benefit of this description will be able to develop the necessary
software to achieve the digital filter required to meet the needs of their
particular situation.
Knowing the transfer characteristics of the system, the pre-distortion is
determined by the equation
Here y{t) is the output signal of the first subtraction unit 1 for any particular
frequency . The 180-degree hybrid forming the first subtraction unit 10 is
characterised by the real, non-negative but frequency dependent
transmission and coupling constants and the frequency
dependent phase shifts . For a 180-degree hybrid
whereas . Moreover a is the real, non-negative but
frequency dependent lumped transmission constant for the isolator 4 and
amplifier 3, and the corresponding frequency dependent phase shift. In
order to achieve perfect cancellation the required distortion of the transmitter
signal S 1 in said first digital mode! 7, the following transfer characteristics
must be imposed
The resulting cancellation will be perfect up to the accuracy of the
assumption (II) as well as the accuracy of the second DAC 8.
To achieve sufficient accuracy of the cancellation filters they should be
calibrated by launching test signals from the AWG 1 through the system. In
principle, the test signals can be series of impulses, each consisting of a
single non-zero sample transmitted by the AWG 1. The impulse will spread
time by the expected delays in the analogue parts of the system. The
separation of each impulse in the test signal should be sufficient not to cause
ambiguities due to this spreading. The length of the calibration series should
be chosen so that when responses are coherently added the obtained
signals levels are well above the system internal noise.
The calibration measurement will directly determine the FIR filter coefficients
of the digital models 6 , 7, 17 representing the analogue system undergoing
calibration. By this method said first and second digital models 7 , 17, as well
as the digital model 6 of the transmitter 19 can all be determined.
For a complete calibration four calibration configurations are advocated,
these four calibration arrangements being illustrated in figures 5A - 5D. in
figure 5A, a test signal is generated by the AWG 1 and transmitted to the
signal extraction unit 18, which can derive the digital model 6 of the
transmitter 19 directly. During this calibration step, no transformation of the
signal is performed in the second digital model 17. The accuracy is limited by
the resolution of the second ADC 16, and will be relatively coarse. However
the transmitter 19 needs not to be determined very accurately due to the
second subtraction unit 2, which cancels transmitter noise.
In figure 5B the characteristics of the 180-degree hybrid forming the third
subtraction unit 15, and the pre-distortion coefficients required for its role as
subtraction unit, are determined. The test signal generated by the AWG 1
and applied to the digital model 6 of the transmitter 19 is fed to both input
ports, one by one, of the third subtraction unit 15, and subsequently to the
signal extraction unit 18, where the characteristics of the 180-degree hybrid
forming the third subtraction unit 1 is derived. During this calibration step, no
transformation of the signal is performed in the second digital model 17.
In figure 5C, the characteristics of the isolator 4 and antenna arrangement 5
is determined. Since both the digital model 6 of the transmitter 19 and the
second digital model 17 of the 180-degree hybrid forming the third
subtraction unit 15 are known, a transmitter noise cancellation loop can be
set up. Isolator 4 characteristics have to be determined to the accuracy level
of the actual radar returns, which thus has to be separated from the isolator
responses. This is done by the signal extraction unit where the system
response all will be at essentially zero range, as already described in
connection with figure 2 . Since signals out of the isolator 4 are weak,
calibration requires the power amplifier 3. This is possible with the transmitter
noise cancelation loop active.
Finally in figure 5D the 180-degree hybrid of the first subtraction unit 10 is
characterized in a way similar to the 180-degree hybrid of the third
subtraction unit 15, wherein the first digital model 7 of the isolator 4 and
antenna arrangement 5 previously determined is used.
The different calibration configuration can all be realized by suitable switches
inserted to redirect signals as required. Moreover, calibrations are suitable
conducted with the AWCW radar system in situ in the radar platform, and
with the radar platform located in an environment corresponding to its
predetermined field of use, for example airborne in case of a SAR
application. Thus effects of antenna interaction with the platform structure will
be fully accounted for.
As will be realised, the invention is capable of modification in various obvious
respects, all without departing from the scope of the appended claims. For
example, the wideband cancellation scheme might also be realized without
said third digital cancellation step where appropriate, if reduced isolation can
be tolerated, whereby the digital mode! 6 of the transmitter 19 and said first
digital model 7 can be integrated into a single digital model. Accordingly, the
drawings and the description thereto are to be regarded as illustrative in
nature, and not restrictive.
CLAIMS
1. A wideband transmitter/receiver arrangement for transmitting and
receiving electromagnetic waves, characterized in that said
transmitter/receiver arrangement comprises:
a digital arbitrary waveform generator AWG (1) connected to a
transmitter ( 9), wherein said waveform generator (1) is configured to
generate an arbitrary waveform within a given bandwidth,
an antenna arrangement (5) configured to emit a transmitter
signal (S1), and to receive an incident signal (S2),
a receiver configured to receive a receiver signal (S3),
an analogue isolator (4) connected to said antenna arrangement
(5), said transmitter ( 19), and said receiver, wherein said analogue
isolator (4) is adapted to route said transmitter signal (S1) from said
transmitter ( 9) to said antenna arrangement (5), and said incident
signal (S2) from said antenna arrangement (5) to said receiver, and to
isolate said transmitter signal (S1) from said receiver signal (S3),
wherein said receiver is adapted to cancel any residual
transmitter signal in said receiver signal (S3) by means of at least one
digital model (6, 7, 17) of at least said isolator (4), said antenna
arrangement (5), and said transmitter (19).
2. The wideband transmitter/receiver arrangement according to claim 1,
characterized in that said transmitter/receiver arrangement further
comprises a first subtraction unit ( 0), which is configured to subtract a
first cancellation signa! (S4) from said receiver signal (S3) received from
said isolator (4), said first cancellation signal (S4) received from said at
east one digital model (6, 7) is configured to correspond to a predicted
effect of said waveform on at least said transmitter (19), said isolator (4)
and said antenna arrangement (5).
The wideband transmitter/receiver arrangement according to claim 2 ,
characterized in that said transmitter/receiver arrangement further
comprises a second subtraction unit (12), which is configured to
subtract a second cancellation signal (S5) from a signal received from
an output of said first subtraction unit 0) , said second cancellation
signal (S5) received from said at least one digital model ( 7) is
configured to correspond to at least a predicted effect of a transmitter
noise generated in said transmitter ( 19) on at least said isolator (4) and
said antenna arrangement (5).
The wideband transmitter/receiver arrangement according to any of
claims 2 or 3, characterized in that said at least one digital model
comprises a first digital model (7) of at least said isolator (4) and said
antenna arrangement (5), and a digital model (6) of said transmitter
( 9), wherein said first cancellation signal (S4) corresponds to an output
signal of said AWG (1) applied to said digital model (6) of said
transmitter ( 19) and to said first digital model (7).
The wideband transmitter/receiver arrangement according to any of
claims 3 or 4 , characterized in that said at least one digital model
comprises a second digital model ( 1 ) of at least said isolator (4) and
said antenna arrangement (5), wherein said second cancellation signal
(S5) corresponds to a digitized representation of said transmitter noise
applied to said second digital model (17).
The wideband transmitter/receiver arrangement according to claim 5,
characterized in that said digitized representation of said transmitter
noise is arranged to be generated by a third subtraction unit (15), which
is coupled to said transmitter (19), and to said digital model (6) of said
transmitter ( 9).
7. The wideband transmitter/receiver arrangement according to claim 6 ,
characterized in that said first and third subtraction units (10, 15) are
analogue subtraction units, in that said first digital model (7) is a digital
model of at least said isolator (4), said antenna arrangement (5) and
said first analogue subtraction unit (10), and in that said second digital
model ( 7) is a digital model of at least said isolator (4), said antenna
arrangement (5), and said third analogue subtraction unit ( 5).
8. The wideband transmitter/receiver arrangement according to any
previous claim, characterized in that said transmitter (19) comprises a
first digital to analogue converter DAC (2) coupled to a power amplifier
(3), wherein said first DAC (2) is connected to said AWG (1), and said
power amplifier (3) is connected to said isolator (4).
9. The wideband transmitter/receiver arrangement according to any of
claims 5 to 8, characterized in that
said first subtraction unit (10) is an analogue subtraction unit ( 0)
connected to an output of said isolator (4) via a first attenuator (9), and
to said first digital model (7) via a second DAC (8),
said second subtraction unit (1 ) is a digital subtraction unit (12)
connected to an output of said first subtraction unit ( 10) via a first
analogue to digital converter ADC ( 11), and to said second digital model
( ), and in that
said second digital model ( 7) is connected to said second
subtraction unit (15) via a second ADC (16), wherein said second
subtraction unit (15) is coupled to said power amplifier (3) via a second
attenuator (14), and to said digital model (6) of said transmitter (19) via
a third DAC (13).
10. The wideband transmitter/receiver arrangement according to any
previous claim, characterized in that said isolator (4) comprises a
power splitter (31), a first circulator (32) connected to a right antenna
(42) of said antenna arrangement (5), a second circulator (36)
connected to a left antenna (43) of said antenna arrangement (5), and a
power combiner (39), wherein said power splitter (31) is configured to
receive a transmitter signal (S1) from said transmitter ( 19), and to feed
said first and second circulators (32, 36), which are configured to route
any incident signals on said right and left antennas (42, 43) to said
power combiner, wherein said power splitter (31) and power combiner
(39) are configured to supply at an output of said power combiner (39) a
difference signal corresponding to a difference between said right and
left antenna incident signals, when said right and left antennas (42, 43)
are fed in phase, or a sum signal corresponding to a sum of said right
and left antenna incident signals when said right and left antennas (42,
43) are fed in opposite phase.
11. The wideband transmitter/receiver arrangement according to claim 10,
characterized in that said isolator (4) further comprises:
a first switch (35) configured to switch the polarity of a signal
received from said power splitter (31) and fed to one of said right and
eft antennas (42, 43), such that said right and left antennas (42, 43)
can be fed in phase, and in opposite phase, and
a second switch (40) connecting said power combiner (39) with
said output of said isolator (4), and configured to switch simultaneous
with said first switch (35), such that to route said difference signal to
said output of said isolator (4) when said right and left antennas (42, 43)
are fed in phase, and to route said sum signal to said output of said
isolator (4) when said right and left antennas (42, 43) are fed in
opposite phase.
12. The wideband transmitter/receiver arrangement according to claim 11,
characterized in that said first circulator (32) is formed of a first 90-
degree hybrid connected by an input port (P1 1) to said power splitter
(31), in that said second circulator (36) is formed of a second 90-degree
hybrid connected by an input port (P1 1) to said power splitter (31),
wherein one of said first and second 90-degree hybrids (32, 36)
is connected to said power splitter (31) via said first switch (35),
wherein said right antenna (42) comprises a first and a second
antenna element (34, 33), which are connected to a coupled and
transmitted port (P12, P22) of said first 90-degree hybrid (32)
respectively,
wherein said left antenna (43) comprises a third and a fourth
antenna element (38, 37), which are connected to a coupled and
transmitted port (P 2, P22) of said second 90-degree hybrid (36)
respectively,
wherein said power combiner (39) is formed of a 80-degree
hybrid, and wherein a first input port (P1 1) of said 180-degree hybrid
(39) is connected to an isolator port (P21) of said second 90-degree
hybrid (36), a second input port (P21) of said 80-degree hybrid (39) is
connected to an isolator port (P21) of said first 90-degree hybrid (32),
and delta and sum ports (P1 , P22) of said 80-degree hybrid (39) are
connected to said second switch (40), and
wherein said second switch (40) connects said delta and sum
ports (P12, P22) with said resistive load (41), and said output of said
isolator (4).
13. Airborne vehicle (44) comprising an airframe, which is mirror symmetric
with respect to a central plane of said vehicle dividing said vehicle in a
right and left part as seen from the rear of said vehicle, characterized
in that said airborne vehicle (44) comprises said wideband
transmitter/receiver arrangement according to any previous claim, and
in that said right antenna (42) is arranged on said right part of said
vehicle (44), and in that said left antenna (43) is arranged on said left
part of said vehicle (44), and in that said right and left antennas (42, 43)
are arranged symmetrically with respect to said central plane.
14. An arbitrary waveform continuous wave AWCW radar system,
especially suitable for providing high resolution microwave SAR, and
iow frequency SAR, characterized in that said radar system comprises
said wideband transmitter/receiver arrangement according to any of the
claims 1 to 12, wherein an output of said second subtraction unit (12) is
connected to a signal extraction unit ( 18) for reconstruction of range
reflectivity.
15. A wideband RF communication unit for simultaneously receiving any
incident RF communication signals and re-transmitting said signals,
characterized in that said communication unit comprises said
wideband transmitter/receiver arrangement according to any of claims
1-12, wherein said output of said second subtraction unit (12), which is
configured to correspond to incident communication signals on said
antenna arrangement (5), feeds said digital arbitrary waveform
generator (1) for substantially immediate re-transmittance of said
incident communication signals.
16. Combined radar, surveillance and communication system,
characterized in that said system comprises said AWCW radar system
according to claims 14 and said RF communication unit of claim 15.
17. A method for transmitting and receiving wideband electromagnetic
waves, characterized in that said method comprises the steps of:
generating an arbitrary waveform within a given bandwidth by
means of a digital arbitrary waveform generator AWG (1) connected to
a transmitter ( 19),
routing a transmitter signal (S1 ) from said transmitter ( 19) to an
antenna arrangement (5) and an incident signal (S2) on said antenna
arrangement (5) to a receiver, and isolating said transmitter signal (S1)
from said receiver signal (S3), by means of an analogue isolator (4)
connected to said antenna arrangement (5), said transmitter ( 19), and
said receiver,
cancelling any residual transmitter signal in said receiver signal
(S3) by means of at least one digital modei (6, 7 , 17) of at least said
isolator (4), said antenna arrangement (5) and said transmitter (19).